WO2012026157A1

WO2012026157A1 - Filter and duplexer

Info

Publication number: WO2012026157A1
Application number: PCT/JP2011/058925
Authority: WO
Inventors: 原基揚; 井上将吾; 岩城匡郁; 堤潤; 中村浩
Original assignee: 太陽誘電株式会社
Priority date: 2010-08-26
Filing date: 2011-04-08
Publication date: 2012-03-01
Also published as: JP2012049758A

Abstract

The present invention is a filter and duplexer that are equipped with: series oscillators (S1-S4); parallel oscillators (P1-P4) that are formed on a piezoelectric substrate (10) and that are connected in parallel to the series oscillators (S1-S4); a inductor (L3) that is connected in series to a parallel oscillator (P4); a capacitor (C1) that is formed on the piezoelectric substrate (10) and connected in parallel to the inductor (L3), and wiring (18) that is formed on the piezoelectric substrate (10) and that connects the parallel oscillator (P4) and the capacitor (C1).

Description

Filters and duplexers

The present invention relates to a filter and a duplexer.

In recent years, with the development of mobile communication systems, mobile phones, portable information terminals, etc. are rapidly spreading. For this reason, the demand for high-frequency filters is expanding, and there is a demand for filters that are particularly small and have sharp characteristics. The filter includes harmonics (second harmonic, third harmonic, etc.), wireless LAN (Local Area Network), and a local area network, which are located on the higher frequency side than the passband in order to suppress cross modulation and interference. High suppression is required in the frequency band used by Bluetooth (registered trademark).

As a high frequency filter, a ladder type filter in which a resonator is connected to a series arm and a parallel arm may be used. Patent Document 1 and Patent Document 2 disclose an invention in which an inductor and a capacitor are connected to a parallel arm of a ladder type filter to increase attenuation on the high frequency side from the pass band.

JP 2008-205947 A JP 2006-333012 A

However, in the conventional technique, there is a possibility that it is difficult to reduce the size of the filter in the filter in which the attenuation pole is formed on the high frequency side from the pass band. In view of the above problems, an object of the present invention is to provide a filter and a duplexer that can form an attenuation pole at a high frequency and can be miniaturized.

The present invention includes a series resonator, a parallel resonator formed on the substrate and connected in parallel to the series resonator, an inductor connected in series to the parallel resonator, and the inductor formed on the substrate. And a capacitor formed on the substrate and connected to the parallel resonator and the capacitor. ADVANTAGE OF THE INVENTION According to this invention, the filter which can form an attenuation pole in high frequency and can be reduced in size can be provided.

In the above configuration, the inductor is connected in series to a plurality of the parallel resonators, and the capacitor is connected in parallel to a first inductor among the plurality of inductors. The capacitor, the first inductor, and the first inductor The attenuation pole formed by the parallel resonator connected in series with the inductor is lower in frequency than the attenuation pole formed by the inductor other than the first inductor and the parallel resonator connected in series with the inductor other than the first inductor. It can be set as the structure located in the side. According to this configuration, the filter can be more effectively downsized.

In the above-described configuration, the attenuation pole formed by the first parallel arm may be positioned on the lower frequency side than the attenuation pole formed by each of the plurality of second parallel arms. According to this configuration, the filter can be downsized and the pass characteristic of the filter can be improved.

In the above configuration, the attenuation pole formed by the first parallel arm may be positioned on a higher frequency side than the pass band of the filter. According to this configuration, the filter can be downsized and the pass characteristic of the filter can be improved.

In the above-described configuration, the attenuation pole formed by the second parallel arm is located in a frequency band corresponding to a second harmonic of the pass band of the filter, and the attenuation pole formed by the first parallel arm is passed through the filter. It can be set as the structure located in the frequency band between a band and the frequency band equivalent to the 2nd harmonic of the said pass band. According to this configuration, the filter can be downsized and the pass characteristic of the filter can be improved.

In the above configuration, the pass band of the filter is a transmission frequency band of a mobile phone, and the attenuation pole formed by the first parallel arm is located in a use frequency band of Bluetooth or a use frequency band of a wireless LAN. can do. According to this configuration, cross modulation and interference can be suppressed.

In the above configuration, the chip includes the series resonator, the parallel resonator, the capacitor, and the wiring on the substrate, and a package substrate on which the chip is mounted, and the inductor is provided on the package substrate. It can be set as the structure currently provided. According to this configuration, the filter can be reduced in size.

In the above configuration, the series resonator and the parallel resonator are acoustic wave resonators having opposing comb-shaped electrodes, and the capacitor is made of the same metal layer as the comb-shaped electrodes included in the acoustic wave resonator, and has electrode fingers. It can be set as the structure provided with the comb-shaped electrode from which pitch differs. According to this configuration, the process can be simplified. In addition, the capacitor can be formed with high accuracy. Furthermore, the capacitance value of the capacitance can be easily adjusted.

In the above configuration, the series resonator and the parallel resonator are acoustic wave resonators having opposing comb-shaped electrodes, and the capacitor is made of the same metal layer as the comb-shaped electrode included in the acoustic wave resonator, and has an acoustic wave It can be set as the structure provided with the comb-shaped electrode from which the propagation direction of this differs. According to this configuration, the process can be simplified. In addition, the capacitor can be formed with high accuracy. Furthermore, the capacitance value of the capacitance can be easily adjusted.

In the above configuration, the wiring may include a region having the same width as that of the comb electrode included in the capacitor or a region having a width larger than the width of the comb electrode. According to this configuration, the inductance component of the wiring can be reduced.

In the above configuration, the series resonator and the parallel resonator are piezoelectric thin film resonators, and the capacitor includes a lower electrode provided in the piezoelectric thin film resonator and an electrode made of the same metal layer as each of the upper electrode; can do. According to this configuration, the process can be simplified. In addition, the capacitor can be formed with high accuracy. Furthermore, the capacitance value of the capacitance can be easily adjusted.

The present invention includes a series resonator formed on a substrate, a parallel resonator formed on the substrate and connected in parallel to the series resonator, an inductor connected in series to the parallel resonator, and the substrate A duplexer including a filter formed on the capacitor and connected in parallel to the inductor, and a wiring formed on the substrate and connecting the parallel resonator and the capacitor. ADVANTAGE OF THE INVENTION According to this invention, the duplexer which can form an attenuation pole in high frequency and can be reduced in size can be provided.

In the above configuration, the filter may be a transmission filter. According to this configuration, the characteristics of the duplexer can be improved. Further, interference between the reception signal and the transmission signal can be suppressed.

According to the present invention, it is possible to provide a filter and duplexer that can form an attenuation pole at a high frequency and can be miniaturized.

1A is a configuration diagram of a series resonator, FIG. 1B is a configuration diagram of a parallel resonator, and FIG. 1C is a diagram illustrating pass characteristics of the series resonator and the series resonator. is there. FIG. 2A is a configuration diagram of a one-stage ladder filter, and FIG. 2B is a diagram illustrating pass characteristics of the one-stage ladder filter. 3A and 3B are configuration diagrams of a multistage ladder filter. FIG. 4A is a configuration diagram of a ladder type filter that forms an attenuation pole on the high frequency side, and FIG. 4B is a diagram showing attenuation characteristics of the ladder type filter that forms an attenuation pole on the high frequency side. FIG. 5A is a configuration diagram of a ladder type filter that forms an attenuation pole in a harmonic, and FIG. 5B is a diagram showing attenuation characteristics of the ladder type filter that forms an attenuation pole in a harmonic. FIG. 6A is a configuration diagram of a ladder type filter to which a capacitor is added, and FIG. 6B is a diagram illustrating attenuation characteristics of the ladder type filter to which a capacitor is added. FIG. 7A is a diagram showing the relationship between the capacitance value and the inductance value, and FIG. 7B is a diagram showing the attenuation characteristic of the ladder filter with the inductance value changed. Fig.8 (a) and FIG.8 (b) are block diagrams of the parallel arm of a ladder type filter. FIGS. 9A and 9B are configuration diagrams of parallel arms of a ladder type filter, and FIG. 9C is a diagram illustrating impedance of the parallel arms. FIG. 10 is a diagram illustrating the impedance of the parallel arms. FIG. 11A is a block diagram illustrating a duplexer, and FIG. 11B is a block diagram illustrating a radio frequency (RF) module of a mobile phone. FIG. 12A is a plan view illustrating the duplexer according to the first embodiment, and FIG. 12B is a cross-sectional view illustrating the duplexer according to the first embodiment. FIG. 13 is a plan view illustrating a transmission filter chip included in the duplexer according to the first embodiment. FIG. 14A to FIG. 14C are plan views illustrating the package substrate included in the duplexer according to the third embodiment. FIG. 15 is a plan view illustrating a transmission filter chip provided in a duplexer according to a modification of the first embodiment. FIG. 16A is a plan view illustrating a piezoelectric thin film resonator, and FIG. 16B is a cross-sectional view illustrating a piezoelectric thin film resonator. FIG. 17A to FIG. 17C are cross-sectional views illustrating a piezoelectric thin film resonator. FIG. 18 is a plan view illustrating a transmission filter chip included in the duplexer according to the second embodiment. FIG. 19 is a plan view illustrating a transmission filter chip included in a duplexer according to a modification of the second embodiment.

First, the ladder filter will be described. 1A is a configuration diagram of a series resonator, FIG. 1B is a configuration diagram of a parallel resonator, and FIG. 1C is a diagram illustrating pass characteristics of the series resonator and the series resonator. is there.

As shown in FIG. 1 (a), when the resonator S21 is a one-terminal-pair resonator, the series resonator has one of its two signal terminals as an input terminal In and the other as an output terminal Out. It is. As shown in FIG. 1B, when the resonator P21 is a one-terminal-pair resonator, the parallel resonator has one of the two signal terminals connected to the ground terminal and the other connected to the input terminal In. This is connected to the short-circuit line of the output terminal Out.

In FIG. 1C, the horizontal axis represents frequency, and the vertical axis represents passage amount. The pass characteristic of the series resonator is indicated by a solid line, and the pass characteristic of the parallel resonator is indicated by a broken line. As shown in FIG. 1 (c), pass characteristics of the series resonator comprises a single resonance point (resonance frequency) f _rs and one antiresonance point and (antiresonance frequency) f _{the as.} Amount passing at the resonance point f _rs is maximized, the amount passed in the antiresonance point f _as is minimized. On the other hand, the pass characteristic of the parallel resonator has one resonance point f _rp and one anti-resonance point f _ap . The passing amount is minimized at the resonance point f _rp , and the passing amount is maximized at the anti-resonance point f _ap .

FIG. 2 (a) is a configuration diagram of a one-stage ladder type filter, and FIG. 2 (b) is a diagram showing pass characteristics of the one-stage ladder type filter.

As shown in FIG. 2A, the series resonator S22 is connected as a series resonator in series with the input terminal In and the output terminal Out, and the parallel resonator P22 is connected as a parallel resonator between the output terminal Out and the ground. The At this time, the resonance point f _rs of the series resonator and the anti-resonance point f _ap of the parallel resonator are designed so as to substantially coincide.

In FIG. 2B, the horizontal axis indicates the frequency, and the vertical axis indicates the passing amount. With the configuration of FIG. 2A, the pass characteristics of the series resonator and the parallel resonator are combined, and the pass characteristic of FIG. 2B is obtained. Amount passing, near the anti-resonance point f _ap with the resonance point f _rs of the series resonators parallel resonator is maximized, the resonance point f _rp of the anti-resonance point f _as and parallel resonators of the series resonator is minimized. The frequency band from the resonance point f _rp of the parallel resonator to the anti-resonance point f _as of the series resonator is a pass band, and the frequency is equal to or lower than the resonance point f _rp of the parallel resonator and the anti-resonance point f _as of the series resonator. The band becomes the attenuation range. Thus, the ladder type filter functions as a bandpass filter.

Next, the multistage ladder type filter will be described. 3A and 3B are configuration diagrams of a multistage ladder filter. In FIG. 3B, the horizontal axis represents frequency and the vertical axis represents attenuation.

As shown in FIG. 3A, the ladder filter F10 includes series resonators S1, S2a, S2b and S3, and parallel resonators P1a, P1b, P2a and P2b. Series resonators S1, S2a, S2b, and S3 are connected in series between the input terminal In and the output terminal Out. The series resonator S1 is connected to the input terminal In. The series resonator S4 is connected to the output terminal Out.

A parallel resonator P1a and a parallel resonator P1b are connected in parallel between the series resonator S1 and the series resonator S2a. A parallel resonator P2a and a parallel resonator P2b are connected in parallel between the series resonator 2b and the series resonator 3, respectively. The parallel resonators P1a, P1b, P2a and P2b are connected to the ground terminal. The multi-stage ladder filter F10 is configured by connecting a plurality of single-stage ladder filters. In addition, in order to suppress the reflection of the signal between each stage, the 1-stage ladder is connected in the form of inverting the filter.

As shown in FIG. 3B, the series resonators S2a and S2b may be used as one series resonator S2 in order to reduce the size of the filter. The parallel resonators P1a and P1b may be one parallel resonator P1, and the parallel resonators P2a and P2b may be one parallel resonator P2. That is, the configuration surrounded by the dotted line in FIG. As will be described later, the series resonator and the parallel resonator function as a capacitor. The capacitance value of the series resonator S2 is equal to the combined capacitance value of the series resonators S2a and S2b when the series resonator S2a and the series resonator S2b are connected in series. The capacitance value of the parallel resonator P2 is equal to the combined capacitance value of the parallel resonators P2a and P2b when the parallel resonator P2a and the parallel resonator P2b are connected in parallel.

Next, a ladder type filter in which an attenuation pole is formed in the harmonic will be described. FIG. 4A is a configuration diagram of a ladder type filter that forms an attenuation pole on the high frequency side, and FIG. 4B is a diagram showing attenuation characteristics of the ladder type filter that forms an attenuation pole on the high frequency side.

As shown in FIG. 4A, in the ladder filter F11, an inductor L1 is connected in series to the parallel resonator P1, and an inductor L2 is connected in series to the parallel resonator P2. One end of the inductor L1 is connected to the parallel resonator P1, and the other end is connected to the ground terminal. One end of the inductor L2 is connected to the parallel resonator P2, and the other end is connected to the ground terminal. The resonator functions as a capacitor outside the passband. For this reason, the parallel resonator P1 and the inductor L1, and the parallel resonator P2 and the inductor L2 each function as an LC resonance circuit. _Let C _{p be} the capacitance value of each of the parallel resonators P1 and P2, and L _{1 be} the inductance value of the inductor L1. In this case, the resonance frequency f ₁ of the LC resonance circuit in parallel with the resonator P1 and the inductor L1 is formed is represented by Equation 1.

The resonance frequency of the LC resonance circuit formed by the parallel resonator P2 and the inductor L2 is also expressed by the same expression as Expression 1.

As shown in FIG. 4B, attenuation poles are formed at the frequencies f ₁ and f ₂ . In the ladder type filter F11, two parallel arms are used. However, an attenuation pole can be added by adding parallel arms and connecting an inductor. Also, the frequency at which the attenuation pole appears can be adjusted by adjusting the capacitance value of the parallel resonator and the inductance value of the inductor. For example, the frequencies f ₁ and f ₂ are included in each of a frequency band (Tx second harmonic) corresponding to the second harmonic of the pass band of the filter and a frequency band (Tx third harmonic) corresponding to the third harmonic. Also good. Thereby, attenuation poles can be formed in harmonics such as Tx second harmonic and Tx third harmonic. Next, a comparative example will be described.

FIG. 5A is a configuration diagram of a ladder type filter that forms an attenuation pole in the harmonic, and FIG. 5B is a diagram showing attenuation characteristics of the ladder type filter that forms an attenuation pole in the harmonic. FIG. 5B shows the result of calculating the attenuation characteristics of the ladder filter F12.

As shown in FIG. 5A, the ladder filter F12 is obtained by adding a series resonator S4, parallel resonators P3 and P4, and an inductor L3 to the ladder filter F11. The parallel resonators P2 and P3 are respectively connected in series to the inductor L2. The parallel resonator P4 is connected in series with the inductor L3. That is, the ladder type filter F12 has three parallel arms, and each parallel arm functions as an LC resonance circuit. Next, simulation of the attenuation characteristics of the ladder filter F12 will be described.

The conditions used for calculating the attenuation characteristics will be described. Consider a case where the ladder type filter F12 is used as a transmission filter of a duplexer. The ladder filter F12 is a W-CDMA (Wideband Code Division Multiple Access) Band 2 transmission filter having the following specifications.
Transmission filter passband: 1850-1910 MHz
Receive filter passband: 1930-1990 MHz
Tx second harmonic frequency band: 3700-3820 MHz
Tx third harmonic frequency band: 5550-5730 MHz
Resonator used: Surface acoustic wave resonator Series resonator S1 capacitance value: 1.87 pF
Capacitance value of each of series resonators S2 to S4: 0.935 pF
Capacitance value of each of parallel resonators P1 to P3: 2.24 pF
Capacitance value of the parallel resonator P4: 1.12 pF
Inductance value L _{1 of} inductor L ₁ : 0.355 nH
Inductance value L _{2 of the} inductor L2: 0.182 nH
Inductance value L _{3 of the} inductor L3: 4.15 nH
The resonance frequencies of the series resonators S1 to S4 are the same, and the resonance frequencies of the parallel resonators P1 to P4 are also the same. The electromechanical coupling constants of the series resonators S1 to S4 and the parallel resonators P1 to P4 are the same.

As shown in FIG. 5B, the ladder type filter F12 has attenuation poles formed in frequency bands corresponding to the Tx second harmonic and the Tx third harmonic. In addition, attenuation poles were formed in 2400 to 2500 MHz, which is a use band of wireless LAN and Bluetooth (hereinafter “BT / LAN”).

Here, the relationship between the inductance value and the filter size will be described. Table 1 summarizes the inductance values L ₁ to L ₃ of the inductors L1 to L3 and the frequency band in which the attenuation pole is formed in the ladder filter F12.

As shown in Table 1, the inductance value _{L 1} of the inductor L1 corresponding to the attenuation pole of Tx3 harmonic is 0.355NH. The inductance value L2 of the inductor L2 corresponding to the attenuation pole of the Tx _second harmonic is 0.182 nH. Inductance value _{L 3} of the inductor L3 which corresponds to the attenuation pole of the BT / LAN is 4.15NH. As shown in the above formula, the resonance frequency of the LC resonance circuit decreases as the inductance value increases. The inductance L3 contributes to the formation of the attenuation pole in the BT / LAN. Note that, although Tx2 is a frequency band lower than Tx3, the inductor L2 has an inductance value smaller than the inductance L1. This is because the parallel resonators P2 and P3 to which the inductor L2 is connected are connected in parallel.

The inductor L3 has a very large inductance value of 4.15 nH. For this reason, an external chip inductor may be used as the inductor L3, and it may be difficult to form the inductor L3 in the same chip or in the same package as other inductors and resonators. As a result, it is difficult to reduce the size and height of the filter. In addition, it is difficult to reduce the size and height of a duplexer that uses a filter as a component.

Next, a ladder filter with a suppressed inductance value will be described. FIG. 6A is a configuration diagram of a ladder type filter to which a capacitor is added, and FIG. 6B is a diagram illustrating attenuation characteristics of the ladder type filter to which a capacitor is added.

As shown in FIG. 6A, the ladder filter F13 includes a capacitor C1. The capacitor C1 is connected in parallel to the inductor L3 (first inductor). As surrounded by a broken line in the figure, a parallel arm including the parallel resonator P1 and the inductor L1 is defined as a parallel arm Pa. A parallel arm including the parallel resonator P2 and the inductor L2 and a parallel arm including the parallel resonator P3 and the inductor L2 are combined to form a parallel arm Pb. A parallel arm including the parallel resonator P4, the inductor L3, and the capacitor C1 is defined as a parallel arm Pc (first parallel arm).

The ladder filter F13 is a filter that forms an attenuation pole in the same band as the ladder filter F12. As described later in FIG. 6 (b), by parallel connection of a capacitor C1, an inductance value L ₃ of the inductor L3 can be greatly reduced.

In FIG. 6B, the attenuation characteristic of the ladder type filter F12 and the attenuation characteristic of the ladder type filter F13 are overlapped. The attenuation characteristic of the ladder type filter F12 is illustrated by a dotted line, and the attenuation characteristic of the ladder type filter F13 is illustrated by a solid line. The calculation parameters of the ladder filter F12 are the same as those used in the calculation of FIG. The calculation parameters of the ladder type filter F13 are the same as the calculation parameters of the ladder type filter F12 except those relating to the next inductor L3 and capacitor C1.
Inductance value L _{3 of} inductor L3: 2.00 nH
Capacitance value C ₁ of capacitor C ₁ : 1.07 pF
This connects the capacitors C1, as FIG. 5 (b) and the attenuation pole in the same frequency band are formed, to adjust the inductance value L ₃ of the inductor L3.

As shown in FIG. 6B, the attenuation characteristics of the ladder type filter F12 and the ladder type filter F13 include attenuation poles in frequency bands corresponding to BT / LAN, Tx second harmonic, and Tx third harmonic, respectively. Been formed. That is, the attenuation pole formed by the parallel arm Pc (first parallel arm) having the capacitor C1, the inductor L3, and the parallel resonator P4 is formed by each of the other parallel arms Pa and Pb (second parallel arm). It is located on the lower frequency side than the attenuation pole. The attenuation pole formed by the parallel arm Pc is located on the higher frequency side than the pass band of the filter. The attenuation pole formed by the parallel arm Pa is located at the Tx third harmonic, and the attenuation pole formed by the parallel arm Pb is located at the Tx second harmonic, as in the case of the ladder filter F12. That is, the attenuation pole formed by the parallel arm Pc is located between the passband of the filter and the Tx second harmonic.

Table 2 summarizes the inductance values L ₁ to L ₃ and the capacitance value C ₁ in the ladder filter F13.

As shown by the dotted line in Table 2, by parallel connection of a capacitor C1, to form an attenuation pole at harmonics and BT / LAN, and the inductance value L ₃ of the inductor L3 can be greatly reduced. Further, the ladder type filter F13 has a larger attenuation amount on the higher frequency side than the frequency band of the Tx second harmonic compared with the ladder type filter F12. Further, the increase in attenuation was more remarkable in the high frequency band.

Further, a case where the capacitance value and the inductance value are changed will be described. In the ladder type filter F13, by changing the inductance value L ₃ of the capacitance value C ₁ and the inductor L3 of the capacitor C1 as the attenuation pole in the same frequency band are formed, was calculated attenuation characteristics. FIG. 7A is a diagram showing the relationship between the capacitance value and the inductance value, and FIG. 7B is a diagram showing the attenuation characteristic of the ladder filter with the inductance value changed.

In FIG. 7A, the horizontal axis represents the capacitance value C ₁ of the capacitor C1, and the vertical axis represents the inductance value L ₃ of the inductor L3. As shown in FIG. 7 (a), with the capacitance value _{C 1} of the capacitor C1 increases, the inductance value _{L 3} of the inductor L3 is reduced.

In FIG. 7 (b), a thick solid line shows the case of _L 3 = _4.0nH. A thick dotted line indicates a case where L ₃ = 3.5 nH. A thick broken line indicates a case where L ₃ = 3.0 nH. A thick alternate long and short dash line indicates a case where L ₃ = 2.5 nH. A thin solid line indicates a case where L ₃ = 2.0 nH. A thin dotted line indicates a case where L ₃ = 1.5 nH. A thin broken line indicates a case where L ₃ = 1.0 nH. A thin alternate long and short dash line indicates a case where L ₃ = 0.5 nH.

As shown in FIG. 7 (b), by adjusting the inductance value _{L 3} of the capacitance value _{C 1} and the inductor L3 of the capacitor C1, the ladder-type filter F13 is the same frequency band as the ladder-type filter F12 (BT / LAN, Tx2 Attenuation poles could be formed at the harmonic and Tx triple). Along with the inductance value L ₃ of the inductor L3 is reduced, due to the capacitance value C ₁ of capacitor C1 is increased in other words, the attenuation of a high frequency band is larger than Tx2 harmonic, characteristics were improved .

Next, the principle that the inductance value can be suppressed will be described. Fig.8 (a) and FIG.8 (b) are block diagrams of the parallel arm of a ladder type filter. FIG. 8A shows the right side parallel arm of the ladder filter F12, and FIG. 8B shows the right side parallel arm Pc of the ladder filter F13. In the equation, the inductor value of the inductor L3 is L ₃ in the case of FIG. 8A and L _{3a in} the case of FIG.

The parallel arm as shown in FIG. 8A functions as an LC resonance circuit as described above. The resonance frequency _{f 3} is expressed by Equation 2.

The parallel arm Pc shown in FIG. 8B also functions as an LC resonance circuit. The resonance frequency f ₄ is expressed by the following equation 3.

Here, since the attenuation circuit is formed in the same frequency band in the resonance circuit of FIG. 8A and the resonance circuit of FIG. 8B, the resonance frequency becomes equal. That is, Formula 2 = Formula 3. The inductance value L _3a is expressed by the following mathematical formula 4.

From Equation 4, by the addition of the capacitor C1, an inductance value _{L 3a} is can be seen that less than _{L 3.}

Next, the impedance of the parallel arm will be explained. First, the parallel arms described in

Patent Documents

1 and 2 will be described. FIGS. 9A and 9B are configuration diagrams of parallel arms of a ladder type filter, and FIG. 9C is a diagram illustrating impedance of the parallel arms. In each of

Patent Documents

1 and 2, the parallel arm shown in each of FIGS. 9A and 9B is described.

The parallel arm shown in FIG. 9 (a) and the parallel arm shown in FIG. 9 (b) are obtained by connecting an inductor L4 in series to the parallel arm Pc shown in FIG. 8 (b). In FIG. 9A, the inductor L4 is connected in series between the parallel resonator P4 and the inductor L3. In FIG. 9B, the inductor L4 and the capacitor C1 are connected in parallel with the inductor L3. The inductor L4 is connected in series with the capacitor C1.

In FIG. 9C, the horizontal axis represents the frequency, and the vertical axis represents the impedance of the parallel arm. The impedance becomes zero at the frequencies f _z1 and f _z2 . The impedance represents the ultimate in frequency f _p. An attenuation pole is formed at each of the frequencies f _z1 and f _z2 at which the impedance becomes zero. That is, a plurality of attenuation poles can be formed by the configurations of FIGS. 9A and 9B. However, as the frequency becomes higher than _fz2 , the impedance increases and has a positive value. As the impedance of the parallel arms increases, it becomes difficult for signals to flow through the parallel arms. In other words, the attenuation characteristics of the filter deteriorate due to an increase in impedance of the parallel arms.

Next, the impedance of the parallel arm Pc shown in FIG. FIG. 10 is a diagram illustrating the impedance of the parallel arms.

As shown in FIG. 10, at a frequency higher than the frequency f _p, the impedance gradually approaches zero. As the impedance approaches zero, signals easily flow through the parallel arms. That is, the suppression of the filter is improved at a high frequency. The change in impedance also appears in the attenuation characteristic shown in FIG. 6B described above, where the improvement in the characteristic becomes more remarkable as the frequency becomes higher.

An embodiment of the present invention based on the above consideration will be described. Example 1 is an example using an acoustic wave resonator. First, the configuration of the duplexer will be described. FIG. 11A is a block diagram illustrating a duplexer, and FIG. 11B is a block diagram illustrating a radio frequency (RF) module of a mobile phone.

As shown in FIG. 11A, the duplexer 100 includes a reception filter 100a and a transmission filter 100b. The reception filter 100a and the transmission filter 100b are connected to a common terminal (antenna terminal) 102 in common. The common terminal 102 is connected to the antenna 104. The reception filter 100a receives a signal from the antenna 104 and outputs the received signal to, for example, an amplifier. The transmission filter 100b outputs a signal input from an amplifier or the like to the antenna 104. The antenna 104 transmits a signal. The configuration of the filter will be described later.

As shown in FIG. 11B, the RF module 110 includes an antenna 104, an antenna switch 112, a duplexer bank 114, and an amplifier module 116. The RF module 110 supports a plurality of communication systems such as a GSM (Global System for Mobile Communication) communication system and a W-CDMA communication system. The GSM system corresponds to the 850 MHz band (GSM850), 900 MHz band (GSM900), 1800 MHz band (GSM1800), and 1900 MHz band (GSM1900). The antenna 104 can transmit and receive both GSM and W-CDMA transmission / reception signals. The duplexer bank 114 includes a plurality of duplexers 114a, 114b, and 114c. Each of the plurality of duplexers is a duplexer corresponding to each of a plurality of communication methods. Note that the duplexer bank 114 may include two duplexers or four or more duplexers. The antenna switch 112 selects a duplexer corresponding to the communication method from a plurality of duplexers provided in the duplexer bank 114 according to the communication method of signals to be transmitted and received, and connects the selected duplexer and the antenna 104. Each duplexer is connected to an amplifier module 116. The amplifier module 116 amplifies the signal received by the reception filter of the duplexer and outputs the amplified signal to the processing unit. The amplifier module 116 amplifies the signal generated by the processing unit and outputs the amplified signal to the duplexer transmission filter.

Next, the configuration of the duplexer according to the first embodiment will be described. FIG. 12A is a plan view illustrating the duplexer according to the first embodiment, and FIG. 12B is a cross-sectional view illustrating the duplexer according to the first embodiment. FIG. 12B is a cross-sectional view along AA1 in FIG.

12A and 12B, the duplexer 100 according to the first embodiment includes a reception filter chip 100c, a transmission filter chip 100d, a package substrate 120, and a sealing unit 122. In FIG. 12A, the sealing portion 122 is seen through. The reception filter chip 100 c and the transmission filter chip 100 d are flip-chip mounted on the package substrate 120 with bumps 124. The package substrate 120 is made of an insulator such as ceramic. The package substrate 120 is a substrate having a two-layer structure including a first layer 120-1 and a second layer 120-2 below the first layer 120-1. As will be described later, wiring layers 120 a, 120 b, and 120 c are formed on the upper surface, intermediate layer, and lower surface of the package substrate 120, respectively, and the wiring layers are connected by via wirings 26. The wiring layer 120b is located between the first layer 120-1 and the second layer 120-2. Part of the wiring provided on the package substrate 120 functions as the inductors L1 to L3. That is, the reception filter chip 100c and the transmission filter chip 100d are mounted on the package substrate 120, thereby realizing a reception filter and a transmission filter. The sealing unit 122 is made of, for example, an insulator such as epoxy resin, or solder, and seals the reception filter chip 100c and the transmission filter chip 100d. The bumps 124 are made of a metal such as gold (Au).

Next, the configuration of the filter provided in the duplexer will be described. FIG. 13 is a plan view illustrating a transmission filter chip included in the duplexer according to the first embodiment. An example filter chip is described as a transmission filter chip 100d.

As shown in FIG. 13, the transmission filter chip 100d includes series resonators S1 to S4, parallel resonators P1 to P4, and a capacitor C1 each including a surface acoustic wave resonator including a comb-shaped electrode.

The configuration of the surface acoustic wave resonator will be described taking the series resonator S1 as an example. The series resonator S <b> 1 is configured by disposing a pair of opposing comb electrodes 12 on the piezoelectric substrate 10, and disposing reflectors 14 on both sides of the comb electrode 12. The comb electrode 12 excites an elastic wave having a frequency corresponding to the pitch of the electrode fingers. The reflector 14 reflects the elastic wave propagating from the comb electrode 12. The series resonators S2 to S4 and the parallel resonators P1 to P4 have the same configuration. The surface acoustic wave propagation directions of the surface acoustic wave resonators are the same. Capacitor C1 includes comb electrodes made of the same metal layer as the comb electrodes included in each of series resonators S1 to S4 and parallel resonators P1 to P4. The electrode finger pitch of the comb-shaped electrodes forming the capacitor C1 is different from the electrode finger pitch of the comb-shaped electrodes included in each of the series resonators S1 to S4 and the parallel resonators P1 to P4. That is, the capacitor C1 is designed so that the resonance frequency of the capacitor C1 is outside the pass band of the transmission filter. For this reason, the surface acoustic wave is hardly excited in the capacitor C1, and the influence on the filter characteristics is suppressed. The capacitor C1 does not need to include a reflector, but preferably includes a reflector in order to improve the Q value. The capacitance value of the capacitor C1 can be adjusted by changing the electrode finger pitch or the electrode finger crossing width.

The comb electrodes constituting the series resonator S1 are connected to the antenna terminal Ant1. The antenna terminal Ant1 is connected to the antenna 104 shown in FIG. The comb electrode included in the series resonator S4 is connected to the transmission terminal Tx1. The transmission terminal Tx1 is connected to the amplifier module 116 shown in FIG. 10B, for example, and receives a transmission signal. The comb electrode provided in the parallel resonator P1 is connected to the terminal 16a. The terminal 16a is connected to the inductor L1. The comb electrode included in the parallel resonator P2 and the comb electrode constituting the parallel resonator P3 are connected to the terminal 16b. The terminal 16b is connected to the inductor L2. The comb electrode provided in the parallel resonator P4 and the comb electrode provided in the capacitor C1 are connected by a wiring 18. The wiring 18 is connected to the terminal 16c. The terminal 16c is connected to the inductor L3. Another wiring 19 connected to the comb electrode included in the capacitor C1 is connected to the terminal 16d. The terminal 16d is connected to a ground terminal provided in the package substrate 120. Bumps 124 for connecting to the package substrate 120 are formed on the white circles in the drawing.

The piezoelectric substrate 10 is made of a piezoelectric material such as lithium tantalate (LiTaO ₃ ) or lithium niobate (LiNbO ₃ ). The comb-shaped electrode 12, the reflector 14, and the wiring connecting the comb-shaped electrode 12 are made of a metal such as aluminum (Al), and are formed of the same metal layer. The series resonators S1 to S4, the parallel resonators P1 to P4, the capacitor C1, the wiring 18, the terminals 16a to 16d, the antenna terminal Ant, and the transmission terminal Tx are formed on the same piezoelectric substrate and are formed from the same metal layer. Become. Although the transmission filter chip 100d has been described with reference to FIG. 13, the reception filter chip 100c may have the same configuration. The reception filter chip 100c includes a reception terminal Rx1 instead of the transmission terminal Tx1.

Next, the package substrate 120 will be described. FIG. 14A to FIG. 14C are plan views illustrating the package substrate included in the duplexer according to the third embodiment. FIG. 14A shows the top surface of the package substrate 120. FIG. 14B shows the top surface of the second layer 120-2 through the first layer 120-1. FIG. 14C shows a perspective view of the first layer 120-1 and the second layer 120-2.

As shown in FIG. 14A, the wiring layer 120a provided on the upper surface of the package substrate 120 includes an antenna terminal Ant2, a transmission terminal Tx2, a reception terminal Rx2, a ground terminal GND1, and

wirings

20a, 22a, and 24a. White circles in the figure represent locations where the bumps 124 in FIG. 13 are connected. Note that the positions of bumps for connection to the reception filter chip 100c are not shown.

The antenna terminal Ant2 is connected to the antenna terminal Ant1 provided in the transmission filter chip 100d. The transmission terminal Tx2 is connected to the transmission terminal Tx1 included in the transmission filter chip 100d. The reception terminal Rx2 is connected to the reception terminal Rx1 included in the reception filter chip 100c. The wiring 20a is connected to a terminal 16a included in the transmission filter chip 100d. The wiring 22a is connected to the terminal 16b. The wiring 24a is connected to the terminal 16c. The ground terminal GND1 is connected to the terminal 16d. Each connection is made via a bump 124. The wiring 20a contributes to the formation of the inductor L1. The wiring 22a contributes to the formation of the inductor L2. The wiring 24a contributes to the formation of the inductor L3. Each configuration of the wiring layer 120 a is connected to each corresponding configuration of the wiring layer 120 b by the via wiring 26.

As shown in FIG. 14B, a wiring layer 120b is formed between the first layer 120-1 and the second layer 120-2 of the package substrate 120. The wiring layer 120b includes an antenna pattern Ant3, an output pattern Tx3, an input pattern Rx3, a ground pattern GND2, and

wirings

20b, 22b, and 24b. White circles in the drawing indicate via wirings 26 for connection to the wiring layer 120a. The antenna pattern Ant3 is connected to the antenna terminal Ant2 provided in the wiring layer 120a. The output pattern Tx3 is connected to the transmission terminal Tx2. The input pattern Rx3 is connected to the receiving terminal Rx2. The ground pattern GND2 is connected to the ground terminal GND1. The wiring 20b is connected to the wiring 20a. The wiring 22b is connected to the wiring 22a. The wiring 24b is connected to the wiring 24c. Each connection is made via the via wiring 26. The wiring 20b contributes to the formation of the inductor L1. The wiring 22b contributes to the formation of the inductor L2. The wiring 24b contributes to the formation of the inductor L3.

As shown in FIG. 14C, the wiring layer 120c provided on the lower surface of the package substrate 120 includes an antenna terminal Ant4, a transmission terminal Tx4, a reception terminal Rx4, and a ground terminal GND3. A white circle in FIG. 14C indicates the via wiring 26 for connecting to the wiring layer 120c. The antenna terminal Ant4 is connected to the antenna pattern Ant3 provided in the wiring layer 120b. The transmission terminal Tx4 is connected to the output pattern Tx3. The reception terminal Rx4 is connected to the input pattern Rx3. The ground terminal GND3 is connected to the wiring 20b, the wiring 22b, the wiring 24b, and the ground pattern GND2 included in the wiring layer 120b. The wiring layer 120c functions as a foot pattern.

As shown in FIGS. 14A to 14C, the inductor L1 is formed by

wirings

20a and 20b. The inductor L2 is formed by the

wirings

22a and 22b. The inductor L3 is formed by

wirings

24a and 24b. That is, the inductors L1 to L3 are provided on the package substrate 120. By mounting the transmission filter chip 100d on the package substrate 120, a transmission filter having the configuration of the ladder filter F13 illustrated in FIG. 6A is formed.

In the first embodiment, parallel resonators P1 to P4 formed on the piezoelectric substrate 10, an inductor L3 connected in series to the parallel resonator P4, and a capacitor C1 formed on the piezoelectric substrate 10 and connected in parallel to the inductor L3. And the transmission filter provided with the wiring 18 which connects the parallel resonator P4 and the capacitor C1 was formed. As a result, as shown in FIG. 6B, an attenuation pole can be formed at a high frequency, and the inductor value of the inductor L3 can be reduced. Since the inductor value is small, the inductor L3 can be downsized. That is, the inductor L3 can be formed on the package substrate 120 as shown in FIGS. 14A to 14C, for example, instead of an external component. Thereby, the filter can be miniaturized. That is, a filter that can be miniaturized can be realized. Further, the duplexer 100 can be downsized by using a downsized filter.

Further, according to the first embodiment, since the parallel resonator P4, the capacitor C1, and the wiring 18 are formed on the same piezoelectric substrate 10, the transmission filter chip 100d can be downsized. For example, when the inductor is formed as a wiring in the filter chip or formed as a wire used for wire bonding, the wiring may be routed. For this reason, the inductor may hinder downsizing of the filter. On the other hand, since the capacitor does not have to be routed, it can occupy less area than the inductor. That is, by forming the capacitor C1 in the transmission filter chip 100d, it is possible to reduce the size of the filter and the duplexer. Inductors L1 to L3 are formed in the package substrate 120. In other words, the inductors L1 to L3 are formed as wirings that run around the package substrate 120. For this reason, the size of the filter and the duplexer can be reduced.

For example, when the wiring 18 has a large inductance value such as a wire, the parallel arm Pc (see FIG. 6A) including the capacitor C1, the inductor L1, and the parallel resonator P4 is the same as FIG. 9A or FIG. The configuration corresponds to the parallel arm as shown in (b). In this case, as described with reference to FIG. 9B, the impedance of the parallel arm increases on the high frequency side, and the attenuation characteristic deteriorates.

On the other hand, the wiring 18 in the first embodiment is a wiring that does not easily function as an inductor, such as a solid pattern. Further, the wiring 18 includes a region wider than the comb-shaped electrode constituting the capacitor C1. That is, as shown in FIG. 13, the width W1 of the region included in the wiring 18 is larger than the width W2 of the comb electrode provided in the capacitor C1. The width direction is the arrangement direction of the electrode fingers of the capacitor C1 in FIG. Since the width W1 of the region included in the wiring 18 is larger than the width of the comb electrode included in the capacitor C1, the inductance component of the wiring 18 can be significantly reduced. Further, even if the width W1 of the region included in the wiring 18 is the same as the width W2 of the comb electrode included in the capacitor C1, the inductance component can be reduced. As shown in FIG. 13, the entire wiring 18 may be wider than the comb-shaped electrode included in the capacitor C1. In this case, the inductance component of the wiring 18 becomes smaller. Further, a part of the wiring 18 may be wider than the comb electrode included in the capacitor C1, and the other area may be narrower than the comb electrode included in the capacitor C1. That is, the wiring 18 includes a region having the same width as that of the comb electrode included in the capacitor C1 or a region having a width larger than the width of the comb electrode.

That is, the inductor value of the wiring 18 is significantly smaller than the inductance value of the inductance L3. For this reason, the circuit configuration of the transmission filter formed by mounting the transmission filter chip of FIG. 13 on the package substrate 120 corresponds to the circuit configuration of the ladder filter F13 shown in FIG. That is, the parallel arm Pc in the first embodiment has a configuration in which L4 is reduced in the parallel arm shown in FIGS. 9A and 9B, and the configuration in FIG. 8B. Therefore, the impedance of the parallel arm Pc approaches zero on the high frequency side as shown in FIG. As a result, the attenuation characteristics of the filter are improved on the high frequency side.

Each resonator is formed of a surface acoustic wave resonator, and the capacitor C1 includes a comb electrode made of the same metal layer as the comb electrode included in each resonator. By patterning the same metal layer, each resonator and capacitor C1 can be formed. Therefore, the process can be simplified, and the cost of the filter and the duplexer can be reduced. Further, the capacitor C1 can be formed with high accuracy by patterning. Furthermore, the capacitance value of the capacitor C1 can be easily adjusted by changing the number of electrode fingers or the electrode finger pitch. By forming the wiring and terminals on the piezoelectric substrate 10 from the same metal layer as the comb electrodes and the reflector, the process can be further simplified.

The capacitor C1 may be connected in parallel with, for example, one of the inductors L1 and L2. For example, when the inductor L1 and the capacitor C1 are connected in parallel, the inductance value of the inductor L1 that contributes to the attenuation pole of the Tx third harmonic can be reduced. However, as shown in Table 1, the inductance value of the inductor L3 that contributes to the attenuation pole of the BT / LAN located on the lowest frequency side in the attenuation station is large. For this reason, the inductor L3 is likely to increase in size among the inductors L1 to L3. Therefore, the filter can be more effectively downsized by connecting the inductor L3 and the capacitor C1 in parallel. Further, as shown in FIG. 6B, the parallel arm Pc includes the capacitor C1, so that the attenuation characteristic on the high frequency side is improved. Therefore, in order to reduce the size of the filter and improve the attenuation characteristics, it is preferable that the attenuation pole formed by the parallel arm Pc is located on the lower frequency side than the attenuation pole formed by each of the parallel arm Pa and the parallel arm Pb.

As described above, in order to suppress cross modulation and interference, it is preferable to form attenuation poles at Tx second harmonic, Tx third harmonic, and BT / LAN. In other words, the attenuation pole formed by the parallel arm Pa is preferably located at the Tx third harmonic, and the attenuation pole formed by the parallel arm Pb is preferably located at the Tx second harmonic. As described above, the attenuation pole formed by the parallel arm Pc is preferably located on the lower frequency side, particularly BT / LAN, than the attenuation pole formed by each of the parallel arm Pa and the parallel arm Pb.

The attenuation characteristics of the transmission filter may deteriorate at higher frequencies than the passband. Therefore, the characteristics of the duplexer can be improved by improving the characteristics on the high frequency side of the transmission filter. In general, the frequency band for reception is located on the higher frequency side than the frequency band for transmission. For this reason, as shown in FIG. 6B, by improving the characteristics on the high frequency side of the transmission filter, it is possible to suppress interference between the reception signal and the transmission signal. That is, it is preferable that the attenuation pole formed by the parallel arm Pc is located on the higher frequency side than the pass band of the transmission filter. Note that a filter as shown in FIG. 6B may be employed for both the transmission filter and the reception filter, or only for the reception filter, but it is particularly preferable to employ the filter for the transmission filter as described above. Furthermore, the filter shown in FIG. 13 can be used as a single filter in addition to the duplexer.

The attenuation poles are formed in bands corresponding to BT / LAN, Tx second harmonic, and Tx third harmonic, respectively, but attenuation poles may be formed in at least one of the three bands. An attenuation pole may be formed in a band other than the above three bands. The band in which the attenuation pole is formed can be changed by adjusting the capacitance value and the inductance value. Incidentally, the capacitance value C ₁ of the capacitor C1 is too large, the attenuation pole becomes a sharply pointed shape in the downward direction of FIG. 6 (b), insufficient suppression in the desired band (BT / LAN etc.), the damping characteristics Can get worse. To improve the attenuation characteristic effectively, the capacitance value C ₁ of the capacitor C1, the desired band, it is preferable to adjust so that the attenuation amount increases.

Each resonator is composed of a surface acoustic wave resonator, but may be composed of another acoustic wave resonator such as a boundary acoustic wave resonator. Although the package substrate 120 has a two-layer structure, it may have a single-layer structure or a multilayer structure of three or more layers. The inductors L1 to L3 may be composed of a plurality of wiring layers, or may be provided in a single wiring layer.

Next, a modification of the first embodiment will be described. FIG. 15 is a plan view illustrating a transmission filter chip provided in a duplexer according to a modification of the first embodiment. The description of the same configuration as that already described is omitted.

As shown in FIG. 15, the propagation direction of the surface acoustic wave of the comb-shaped electrode included in the capacitor C1 differs from the series resonators S1 to S4 and the parallel resonators P1 to P4 by, for example, 90 °. This makes it difficult for the surface acoustic wave to be excited in the capacitor C1. If the propagation direction of the surface acoustic wave is different between the capacitor C1, the series resonators S1 to S4 and the parallel resonators P1 to P4, the excitation of the surface acoustic wave is suppressed. However, in order to effectively suppress the excitation of the surface acoustic wave, the difference in the propagation direction is preferably 90 °.

Example 2 is an example using a piezoelectric thin film resonator. Since the configuration of the duplexer is the same as that already described, description thereof is omitted. Next, the piezoelectric thin film resonator will be described. FIG. 16A is a plan view illustrating a piezoelectric thin film resonator, and FIG. 16B is a cross-sectional view illustrating a piezoelectric thin film resonator. FIG. 16B is a cross-sectional view taken along the line BB1 of FIG.

As shown in FIG. 16A, the piezoelectric thin film resonator 130 includes a substrate 132, a lower electrode 134, an upper electrode 136, and a piezoelectric thin film 138. As shown in FIG. 16B, the lower electrode 134 and the upper electrode 136 sandwich the piezoelectric thin film 138. Elastic waves are excited in a region 140 where the lower electrode 134, the piezoelectric thin film 138, and the upper electrode 136 overlap. A through hole 142 that penetrates the substrate 132 is formed under the region 140. For this reason, excitation of elastic waves is not hindered. The substrate 132 is made of, for example, silicon or glass. The lower electrode 134 and the upper electrode 136 include, for example, aluminum (Al), copper (Cu), molybdenum (Mo), tungsten (W), tantalum (Ta), platinum (Pt), ruthenium (Ru), rhodium (Rh), A film made of iridium (Ir), chromium (Cr), titanium (Ti), or the like or a combination thereof can be used. The piezoelectric thin film 138 is made of, for example, aluminum nitride (AlN), zinc oxide (ZnO), lead zirconate titanate (PZT), lead titanate (PbTiO ₃ ), or the like. In the piezoelectric thin film resonator, the resonance frequency can be adjusted by adjusting the electrode film thickness of each of the lower electrode 134 and the upper electrode 136.

Further, another configuration example of the piezoelectric thin film resonator will be described. FIG. 17A to FIG. 17C are cross-sectional views illustrating a piezoelectric thin film resonator. FIGS. 17A to 17C are cross-sectional views taken along the line BB1 of FIG.

As shown in FIG. 17A, a cavity 144 may be provided in the substrate 132. As shown in FIG. 17B, the lower electrode 134, the upper electrode 136, and the piezoelectric thin film 138 may be raised, and a gap 146 may be formed between the lower electrode 134 and the substrate 132. As shown in FIG. 17C, an acoustic reflection film 148 may be provided between the substrate 132, the lower electrode 134, and the piezoelectric thin film 138. The acoustic reflection film 148 has a structure in which a film having a high acoustic impedance and a film having a low acoustic impedance are alternately stacked with a film thickness of λ / 4 (λ: wavelength of elastic wave). 16 (b), 17 (a) and 17 (b) FBAR (Film Bulk Acoustic Resonator) type or SMR (Solidly Mounted Resonator) type in FIG. 17 (c). A vessel may be used.

FIG. 18 is a plan view illustrating a transmission filter chip included in the duplexer according to the second embodiment. The description of the same configuration as that already described is omitted. As shown in FIG. 18, the series resonators S1 to S4 and the parallel resonators P1 to P4 are formed of piezoelectric thin film resonators. In the drawing, the piezoelectric thin film resonator is shown by shading. As described above, the substrate 132 is a non-piezoelectric substrate made of, for example, silicon or glass. The capacitor C1 includes a lower electrode made of the same metal layer as the lower electrode included in each resonator. The capacitor C1 includes an upper electrode made of the same metal layer as the upper electrode included in each resonator. That is, the capacitor C1 has a configuration in which the piezoelectric thin film is sandwiched between the lower electrode and the upper electrode, like each resonator. The capacitor C1 is designed so that the resonance frequency of the capacitor C1 is outside the passband of the filter. Other configurations are the same as those shown in FIG.

According to the second embodiment, similarly to the first embodiment, the attenuation pole is formed at a high frequency, and the filter can be downsized. Each resonator and the capacitor C1 can form the lower electrode tailband upper electrode from the same metal layer, and can form a piezoelectric thin film from the same piezoelectric film. That is, by patterning the same layer, each resonator and the capacitor C1 can be formed at a time, so that the process can be simplified. In addition, instead of the piezoelectric thin film 138, a dielectric film made of, for example, silicon oxide (SiO ₂ ) or silicon nitride (Si ₃ N ₄ ) may be used for the capacitor C1. Thereby, the capacity | capacitance of the capacitor C1 can be adjusted and the excitation of the elastic wave in the capacitor C1 can be suppressed. Further, the capacitance value of the capacitor C1 can be easily adjusted by changing the size of the lower electrode 134 and the upper electrode 136 or the distance between the electrodes.

Next, a modification of the second embodiment will be described. FIG. 19 is a plan view illustrating a transmission filter chip included in a duplexer according to a modification of the second embodiment. As shown in FIG. 19, the capacitor C1 is composed of opposing comb electrodes. Since the substrate 132 is a non-piezoelectric substrate, the comb electrode does not excite an elastic wave and functions as the capacitor C1. That is, you may use combining an elastic wave resonator and a piezoelectric thin film resonator.

Although the embodiments of the present invention have been described in detail above, the present invention is not limited to such specific embodiments, and various modifications and changes can be made within the scope of the gist of the present invention described in the claims. It can be changed.

Claims

A series resonator;
A parallel resonator formed on a substrate and connected in parallel with the series resonator;
An inductor connected in series to the parallel resonator;
A capacitor formed on the substrate and connected in parallel to the inductor;
A filter comprising: a wiring formed on the substrate and connecting the parallel resonator and the capacitor.
The inductor is connected in series to each of the plurality of parallel resonators,
The capacitor is connected in parallel to a first inductor among the plurality of inductors,
An attenuation pole formed by a first parallel arm having the capacitor, the first inductor, and a parallel resonator connected in series with the first inductor is:
It is located in the low frequency side rather than the attenuation pole which a 2nd parallel arm which has an inductor other than the said 1st inductor and a parallel resonator connected in series with an inductor other than the said 1st inductor forms. Item 10. The filter according to item 1.
3. The filter according to claim 2, wherein the attenuation pole formed by the first parallel arm is located on a lower frequency side than the attenuation pole formed by each of the plurality of second parallel arms.
The filter according to claim 2 or 3, wherein the attenuation pole formed by the first parallel arm is located on a higher frequency side than a pass band of the filter.
The attenuation pole formed by the second parallel arm is located in a frequency band corresponding to a second harmonic of the pass band of the filter,
5. The attenuation pole formed by the first parallel arm is located in a frequency band between a pass band of the filter and a frequency band corresponding to a second harmonic of the pass band. The filter according to any one of the above.
The pass band of the filter is a transmission frequency band of a mobile phone,
The filter according to any one of claims 2 to 5, wherein the attenuation pole formed by the first parallel arm is located in a use frequency band of Bluetooth or a use frequency band of a wireless LAN.
A chip comprising the series resonator, the parallel resonator, the capacitor and the wiring on the substrate;
A package substrate on which the chip is mounted,
The filter according to claim 1, wherein the inductor is provided on the package substrate.
The series resonator and the parallel resonator are acoustic wave resonators provided with opposing comb electrodes,
8. The filter according to claim 1, wherein the capacitor includes a comb electrode made of the same metal layer as the comb electrode included in the acoustic wave resonator and having a different electrode finger pitch.
The series resonator and the parallel resonator are acoustic wave resonators provided with opposing comb electrodes,
8. The filter according to claim 1, wherein the capacitor includes a comb-shaped electrode made of the same metal layer as that of the comb-shaped electrode included in the acoustic wave resonator and having a different propagation direction of the acoustic wave.
10. The filter according to claim 8, wherein the wiring includes a region having the same width as that of the comb electrode included in the capacitor or a region having a width larger than the width of the comb electrode.
The series resonator and the parallel resonator are piezoelectric thin film resonators,
The filter according to any one of claims 1 to 7, wherein the capacitor includes an electrode made of the same metal layer as each of a lower electrode and an upper electrode included in the piezoelectric thin film resonator.
A series resonator formed on a substrate, a parallel resonator formed on the substrate and connected in parallel to the series resonator, an inductor connected in series to the parallel resonator, and formed on the substrate A duplexer comprising: a capacitor connected in parallel to the inductor; and a wiring formed on the substrate and connected to the parallel resonator and the capacitor.
13. The duplexer according to claim 12, wherein the filter is a transmission filter.