WO2009156715A1

WO2009156715A1 - Self oscillating flyback circuit

Info

Publication number: WO2009156715A1
Application number: PCT/GB2009/001555
Authority: WO
Inventors: Mark Wayne Huggins; Martin Wong
Original assignee: Xipower Limited
Priority date: 2008-06-23
Filing date: 2009-06-19
Publication date: 2009-12-30
Also published as: GB2473407A; GB2473407B; GB201100679D0; GB0811483D0

Abstract

A self oscillating flyback circuit (101) suitable for use in switch mode power supplies. The circuit has an input (103), a switching transistor (117), a primary winding (109) for receiving energy from a primary power source, a feedback circuit comprising a feedback winding (111) for providing a feedback signal to the base of the switching transistor and an inductor winding (113) for transferring energy to an output (105). The input has an active component (115) which modulates the input signal and provides a controlled signal to the circuit. The feedback circuit also has a second active component (119) which modulates the signal to the base of the switching transistor. The self oscillating flyback converter can enhance and extend the primary switch device drive current and voltage control resulting in a significant increase in operational input voltage and can provide a wide operational input voltage range to the class of switch mode power converter known in the art as Self oscillating flyback power converters.

Description

SELF OSCILLATING FLYBACK CIRCUIT

Field of the Invention This invention relates generally to improvements in switching circuits known as Self Oscillating Flyback circuits which are typically found in switch mode power supplies.

Description of the Prior Art The class of switch mode power supply known as the Self

Oscillating Flyback converter is well known in the art.

Switching operation is achieved through positive feedback of the power transformer. No forced oscillation is required which results in simple and cost effective power conversion. In addition, the positive feedback of the power transformer means that this class of switch mode power converter can operate to input voltages of less than 0.5V

Although a typical self oscillating converter can operate at low input voltages, designing for a wide range of input voltage is problematic due to efficiency concerns in the start up and switch drive circuits.

In general therefore, the circuit has a positive feedback winding which amplifies the voltage of a primary winding and applies it to the base or gate of a transistor switching device. This drive voltage must be limited to keep the switching device within its safe operating range and the feedback voltage limits the maximum operational input voltage range of the converter. Also, the variable drive current provided by the positive feedback winding makes the control of output voltage over a wide input voltage range very challenging.

Figure 1 shows a known type of self oscillating flyback converter. The circuit 1 comprises an input (Vin) 3, an output (Vout) 5, and ground 7. Three coupled windings provide inductance in the circuit. The primary winding 9 is used to transfer energy from a power source into the inductor core. The inductor winding 13 is used to transfer energy from the inductor core to the output supply voltage. The feedback winding 11 supplies a drive voltage and current to the primary switching transistor 17 and ensures that the inductor core discharges all of the stored energy into a secondary circuit before a new primary switch cycle can begin. Diode D2 29 and capacitor C3 31 provide dc voltage bias for the feedback winding Nfb 11.

In use, Resistor (R4) 15 provides a source of start up bias current for primary switching transistor (Ql) 17. Once Ql 17 commences to turn on, a voltage is generated across winding 11 Ll Nfb which reinforces the base drive to Ql 17 ensuring Ql 17 turns on hard very quickly. The turns ratio of the primary winding 9 (Np) and of feedback winding 11 Nfb is chosen to enable the converter to operate at the required minimum voltage i.e. for a minimum operational voltage of 0.7V a Nfb/Np ratio of 2 would be required. A positive turns ratio Nfb/Np ensures positive feedback and thus continuous oscillation.

Resistor (R2) 21 helps limit Ql 17 base current during turn on and turn off switch cycles. Resistor (Rl) 19 helps limit Ql 17 base current during turn on only. During Ql 17 on period the collector current ramps up until Ql base current can no longer support the collector current, this is determined by Ql hfe.

As Ql base current no longer supports it collector current Np Primary winding 9 voltage starts to collapse forcing the voltage across Nfb feedback winding 11 to collapse which reduces Ql base drive. The positive feedback ensures Ql 17 turns off very quickly as per turn on. Diode Dl 23 ensures the stored base emitter charge of Ql 17 is removed quickly to further complete the fast turn off of Ql 17.

At Ql 17 turn off, the energy stored in the core of the inductor is transferred to the secondary circuit through inductor winding Ns 13, Diode D3 27 and capacitor C4 25. The primary winding Np 9 and the feedback winding Nfb 11 are coupled as transformer windings, whereas the primary winding Np 9 and the inductor winding Ns 13 are coupled as inductors.

The Ql 17 turn on period is determined by base current, Ql hfe and the primary winding Np 9 inductance. Ql switch turn off period is determined by the discharge of the inductor core energy into the secondary load namely the feedback winding Nfb inductance and the secondary load current.

Because the Ql 17 switch period is driven by the base current, the converters switching frequency is very sensitive to input voltage variation, as the input voltage increases so does the operating frequency. Also, as the input voltage increases so does the power dissipated by resistors R1/R2 and R4 these two effects limit the practical operating range of input voltage. A typical control system that would regulate the output voltage would comprise a circuit added to Ql base that would steal base current forcing early turn off of Ql to restrict the energy transferred by Ll Np/Ns. However due to the wide ranging base current with input voltage, as a result of using a limiting resistor, any control system is greatly restricted in control range .

Accordingly, state of art self oscillating flyback converters as described above have a limited operational input voltage range.

It is an object of the present invention to provide an improved self oscillating fly back circuit.

Summary of the Invention

In accordance with a first aspect of the invention there is provided a self oscillating fly back circuit comprising: an input; a switching transistor; a primary winding for receiving energy from a primary power source; a feedback circuit comprising a feedback winding for providing a feedback signal to the base of the switching transistor; and an inductor winding for transferring energy to an output; wherein the input has a first active component which modulates an input signal and provides a controlled signal to the circuit and/or the feedback circuit has a second active component which modulates the signal to the base of the switching transistor. Preferably, the first active component is a first active current source.

Preferably, the second active component is a second active current source.

Preferably, the first active current source comprises a constant current source which provides a substantially constant current to the circuit over a range of input voltages.

Preferably, the second active current source comprises a constant current source which provides a substantially constant current to the circuit over a range of input voltages.

Preferably, the first and/or second active current source comprises a field effect transistor (FET).

Optionally, the FET is a Junction Field Effect Transistor (JFET)

Optionally, the FET is a Metal Oxide Semiconductor Field Effect Transistor (MOSFET) .

Preferably, the circuit is regulated by controlling the base drive current of the switching transistor.

Preferably, the circuit is adapted to lowering the maximum supported collector current by reducing the base current.

Preferably, the circuit further comprises a shunt which carries current away from the base of the switching transistor.

Preferably, the shunt comprises a transistor and a diode. Preferably, the shunt is adapted to operate only when the switching transistor is on.

Preferably, the shunt is driven by a current mirror circuit.

Preferably, the shunt is driven by a shunt regulator.

A self oscillating flyback converter incorporating current sources can enhance and extend the primary switch device drive current and voltage control resulting in a significant increase in operational input voltage.

The current source can be implemented as diode connected/configured JFET and or diode connected/configured depletion mode MOSFET devices. These configured devices act as a constant current source whose current value can be programmed using an additional resistor in series with the diode junction.

In one aspect, the present invention provides a wide operational input voltage range to the class of switch mode power converter known in the art as Self oscillating flyback power converters.

Brief Description of the Drawings

The present invention will now be described by way of example only with reference to the accompanying drawings in which:

Figure 1 is a circuit diagram of a self oscillating fly back circuit of known design; Figure 2 is a circuit diagram of a first embodiment of a self oscillating fly back circuit in accordance with the present invention;

Figure 3 is a circuit diagram of a second embodiment of a self oscillating fly back circuit in accordance with the present invention; and

Figures 4a and 4b are graphs which compare the performance of an active component (a JFET) and a passive component (a resistor) .

Detailed Description of the Preferred Embodiments

Figure 2 shows a first embodiment of the present invention. The circuit 101 comprises an input (Vin) 103, an output (Vout) 105, and ground 107. Three coupled windings provide inductance in the circuit. The primary winding 109 is used to transfer energy from a power source into the inductor core. The inductor winding 113 is used to transfer energy from the inductor core to the output supply voltage. The feedback winding 111 supplies a drive voltage and current to the primary switching transistor 117 and ensures that the inductor core discharges all of the stored energy into a secondary circuit before a new primary switch cycle can begin.

The circuit further comprises active component 115 which modulates a signal from input 103 and active component 119 which modulates a feedback signal from the feedback winding 111. The circuit also comprises transistor and a number of components which operate to shunt current away from the base of transistor 117. Specifically, Transistor QIl 129 and diode D5 131 shunt current away from the base of transistor 117 and transistor Q9 133 ensures that current is only shunted during transistor 117 ON time. This is driven by the current mirror circuit 135. The addition of two current sources Il and 12 enables a significant improvement in operational input voltage range.

In use, an input voltage is applied between the Vin node 103 and Gnd node 107 and a small current flows through current source 115 into capacitor C23 121. Current source 115 is configured as a constant current source ensuring the start up current remains constant over the full input voltage range thus reducing losses. Once capacitor C23 121 has charged to close to a Vbe (base- emitter) level, transistor QlO 117 will begin to turn on. As the collector voltage of QlO 117 falls, the voltage across L2 pins 6,5 (not shown) rises inducing a positive voltage across L2 pins 4,3 this positive feedback ensures that QlO 117 turns fully on very quickly even with a supply voltage of less than 0.4V

Once QlO 117 has turned on, the current will ramp up in the inductor 109 until the QlO 117 base current can no longer support the collector current OR the inductor core saturates. Once the voltage across primary winding 109 starts to fall as a result of insufficient QlO 117 base current or inductor core saturation the voltage across feedback winding 111 will begin to reverse and turn off QlO very quickly.

Feedback winding 111 provides a feedback signal to drive the base of the flyback switching transistor QlO 117 and facilitate oscillation. The purpose of this winding 111 is to ensure that the transistor turns fully on at low line voltage and once the inductor core has been demagnetised. The winding 109 acts to terminate the QlO 117 switching period and ensure complete core demagnetisation theough inductor winding 113 prior to a new switch cycle.

Energy is transferred from the primary winding 109 into the inductor winding 113 during QlO 117 ON period. This stored energy is then released into the output capacitors and load through diode 127 during QlO OFF period. The rate of transfer (switching frequency) and peak QlO 117 collector current determines the power delivered to the load.

The base drive current is supplied to the primary switching transistor QlO 117 through active component 119 configured as a current source. Current source 119 ensures the base current is independent of input supply voltage or output load current. The maximum collector current of QlO 117 is determined by the fixed base current and QlO 117' s hfe The regulator control loop eliminates the variation in QlO hfe affecting collector current.

The circuit of the present invention is regulated by controlling the base drive current of switching transistor QlO 117. By reducing the base current the maximum supported collector current is lowered resulting in a reduction of energy transferred per cycle of operation.

Transistor QIl 129 and diode D5 131 shunts current away from the base of QlO 117, transistor Q9 133 ensures that current is only shunted during QlO 117 ON time. QIl 129 is driven by current mirror Q8 135 and shunt regulator U2 137 generates an error current which drives the diode half of Ql. A detailed description of a second embodiment of a circuit in accordance with the present invention will now be provided with reference to Figure 3.

The circuit 201 comprises an input (Vin) 203, an output (Vout) 205, and ground 207. Three coupled windings provide inductance in the circuit. The primary winding 209 is used to transfer energy from a power source into the inductor core. The inductor winding 213 is used to transfer energy from the inductor core to the output supply voltage. The feedback winding 211 supplies a drive voltage and current to the primary switching transistor 217 and ensures that the inductor core discharges all of the stored energy into a secondary circuit before a new primary switch cycle can begin.

The circuit further comprises active component 215 which modulates a signal form input 203 and active component 219 which modulates a feedback signal from the feedback winding 211. The two current sources are provided by a JFET Q4 215 and depletion mode n channel MOSFET Q6 219.

The circuit also comprises transistor a number of components which operate to shunt current away from the base of transistor 217. Specifically, Transistor Q3 229 and diode D5 231 shunts current away from the base of transistor 217 and transistor Q5 233 ensures that current is only shunted during transistor 117 ON time. This is driven by the current mirror circuit 235.

With reference to figure 3 output voltage regulation is achieved as follows; Q2 217 base drive current is removed by transistor Q3 229 through D3 231. Transistor Q3 229 is controlled by voltage reference source Ul 237 through current mirror device Ql. As output voltage rises Ql 217 current increases supplying Q3 229 with more base current which then starves Q2 217 of further base current. As Q2 217 base current is reduced so does Q2 collector current which results in a lower transferred energy to inductor winding 213 producing a lower output voltage.

Figure 3 shows a typical implementation using a JFET as a source of start-up current, Q4 215, and a depletion mode MOSFET as the turn on base drive current source, Q6 219, of primary switching transistor, Q2 217.

On application of an input voltage between the Vin node 203 and Gnd node 207 a small current will flow through the JFET Q4 215 into capacitor C8 221. It is to be noted that Q4 215 is configured as a constant current source ensuring the start up current remains constant over the full input voltage range thus reducing losses. Once capacitor C8 221 has charged to close to a Vbe transistor Q2 will begin to turn on. As Q2's collector voltage falls the voltage across primary winding 209 rises inducing a positive voltage across feedback winding 211 this positive feedback ensures that Q2 217 turns fully on very quickly even with a supply voltage of less than 0.4V

Once Q2 217 has turned, on current will ramp up in the primary winding until Q2 base current can no longer support the collector current OR the inductor core saturates. Once the voltage across the primary winding 209 starts to fall as a result of insufficient Q2 base current or inductor core saturation, the voltage across feedback winding 211 will begin to reverse and turn off Q2 very quickly. Feedback winding 211 provides a feedback signal to drive the base of the flyback switching transistor Q2 217. The purpose of this winding 211 is to ensure that the transistor turns fully on at low line voltage and once the inductor core has been demagnetised (energy discharged into secondary capacitors ClO, 11 and 12). The winding also acts to terminate Q2 217 switching period and ensure complete core demagnetisation prior to a new switch cycle. Finally the positive feedback winding facilitates self oscillation.

Energy is transferred from the primary winding 109 into the inductor winding 213 during Q2 ON period. This stored energy is then released into the output capacitors and load through diode Dl during Q2 OFF period. The rate of transfer (switching frequency) and peak Q2 collector current determines the power delivered to the load.

Base drive current is supplied to the primary switching transistor Q2 217 through depletion mode MOSFET Q6 219 configured as a current source. Q6 219 ensures the base current is independent of input supply voltage or output load current. The maximum collector current of Q2 217 is determined by the fixed base current and Q2's hfe. The regulator control loop eliminates the variation in Q2 hfe affecting collector current.

Regulation is achieved by controlling the base drive current of primary switching transistor Q2 217. By reducing the base current the maximum supported collector current is lowered resulting in a reduction of energy transferred per cycle of operation. Transistor Q3 229 and diode D3 231 shunts current away from the base of Q2 217, transistor Q5 233 ensures that current is only shunted during Q2 217 ON time. Q3 229 is driven by current mirror Ql 235 and shunt regulator Ul 237 generates an error current which drives the diode half of Ql 235. Shunt regulator Ul 237 performs regulation in generating a 2.5V reference at the centre point of Rl 239 and R9 241. Ul 237 cathode current is varied to maintain a voltage of 2.5V at its REF pin (R1/R9 connection) . Ql 235 provides a near identical current at pin 2 as being driven on pin 1 this enables the drive of Q3 229 from Ul 237 without inversion.

Figures 4a and 4b are graphs which compare the performance of a JFET and a resistor. In figure 4a, graph 301 plots power 301 against input voltage 305. The response curve for a resistor 307 shows a significant increase in output power over the range 0 to 30 Volts. Whereas, the response curve for the JFET 309 shows a relatively small rise over the range 0 to 30 Volts.

In figure 4b, the graph 311 plots current 313 against voltage 315. The response curve for a resistor 317 shows a significant increase in output current over the range 0 to 30 Volts. Whereas, the response curve for the JFET 319 shows a substantially constant value over the range 0 to 30 Volts.

The circuit of figure 3 is configured to provide a 4.47V output. This value maybe programmed to another output voltage through the formulae:

Vout = 2.5* (1+(R1/R9) ) At low line voltage and maximum output power the peak input current can reach in excess of 1.5A it is therefore necessary to keep the path length to the power source (battery/cell terminals) as short as possible to minimise line inductance. The input leads should also be appropriately rated to minimise voltage drop.

The design point for the coupled inductor Ll is at or near the extreme of operation namely at minimum input voltage and maximum output power.

Energy Stored, E, in the core of an inductor:

E=0.5*L*I²

Where L is winding inductance and I is winding current

The voltage, V, across the inductance is expressed as:

V=L*di/dt

Where L is the winding inductance and di/dt is the rate of change of winding current .

Output Power of the converter P is defined as:

P=Vo*Io=E*f

Where Vo is the output voltage, Io the output current and f the operational frequency of the converter. Substitution and rearrangement of the above three equations provide expressions for the inductor winding current and its critical maximum inductance value :-

L_crit= (0.5*V²)/(P*f)

At minimum input voltage, V, operational frequency is at its minimum (for complete demagnetisation mode) We select a minimum operational frequency twice that of upper human hearing range 40khz.

Allowing for 50% efficiency and Vce_sat of 15OmV on a minimum input supply of 0.8V we need the following maximum (critical) primary inductance:

Lcrit= (0.5*0.65²) /(0.5*40e3) = 10.6uH

Peak primary inductor current:

I=V/ (L*f) = 1.53A Maximum.

Under output short circuit conditions the peak primary inductor winding current will be significantly higher, as supported by the switching device hfe and its base current.

The converters operational frequency in short circuit will fall into the sub khz region due to extended discharge of core energy into secondary circuit as secondary winding voltage collapses to less than 10OmV. Start up at a minimum input voltage of 0.8V requires the feedback winding to multiply the primary winding voltage by at least two times to generate at least 1.2V of base emitter drive. This equates to around 20 Turns for the feedback winding. To keep the Vbe reverse voltage below the 6V device rating it would be desirable for the secondary inductor winding to have the same number of turns as the feedback winding.

A check on secondary discharge time at maximum output power:

An inductor of 10.6uH with 8T provides 66uH with 2OT

Secondary peak current, Is = SQRT (- (Vo²*Io*To) / (Ls-O .5*Vo*Ls) ) =

0.561A

Secondary discharge period, Ts = Ls*Is/ (Vo+Vd) = 6.7uS

Primary charge time, Tp = Lp*Ip/ (0.5*Vp) = 24.95uS

Minimum operational frequency = 1/(6.7e-6 +24.95e-6) = 31.59khz

Inductor core selected on its ability to store required energy in small volume and exhibiting low core losses and the ability to accommodate all windings.

Wire diameter sized on rms current requirement and losses.

Advantageously, the present invention can provide:

A self oscillating flyback converter with wide operational input voltage range;

Use of two current sources to restrict primary switching transistor drive current and start up current; . Significantly reduced range of operating frequency over input voltage range due to fixed on drive current and start up bias current; .

Use of diode configured JFET device to provide current source;

Use of diode configured depletion mode MOSFET to provide current source;

25OmW converter operational from 0.5V to 35V primary input voltage; and Programmable output voltage from 3V to 5.5V at 5OmA output current .

Improvements and modifications may be incorporated herein without deviating from the scope of the invention.

Claims

1. A self oscillating fly back circuit comprising: an input; a switching transistor; a primary winding for receiving energy from a primary power source; a feedback circuit comprising a feedback winding for providing a feedback signal to the base of the switching transistor; and an inductor winding for transferring energy to an output; wherein the input has a first active component which modulates an input signal and provides a controlled signal to the circuit and/or the feedback circuit has a second active component which modulates the signal to the base of the switching transistor.

2. A self oscillating fly back circuit as claimed in claim 1 wherein, the first active component is a first active current source .

3. A self oscillating fly back circuit as claimed in claim 1 or claim 2 wherein, the second active component is a second active current source.

4. A self oscillating fly back circuit as claimed in any preceding claim wherein, the first active current source comprises a constant current source which provides a substantially constant current to the circuit over a range of input voltages.

5. A self oscillating fly back circuit as claimed in any preceding claim wherein, , the second active current source comprises a constant current source which provides a substantially constant current to the circuit over a range of input voltages.

6. A self oscillating fly back circuit as claimed in any preceding claim wherein, the first and/or second active current source comprises a field effect transistor (FET).

7. A self oscillating fly back circuit as claimed in claim 6 wherein, the FET is a Junction Field Effect Transistor (JFET)

8. A self oscillating fly back circuit as claimed in claim^' 6 or claim 7 wherein, the FET is a Metal Oxide Semiconductor Field

Effect Transistor (MOSFET) .

9. A self oscillating fly back circuit as claimed in any preceding claim wherein, the circuit is regulated by controlling the base drive current of the switching transistor.

10. A self oscillating fly back circuit as claimed in any preceding claim wherein, the circuit is adapted to lowering the maximum supported collector current by reducing the base current .

11. A self oscillating fly back circuit as claimed in any preceding claim wherein, , the circuit further comprises a shunt which carries current away from the base of the switching transistor.

12. A self oscillating fly back circuit as claimed in claim 11 wherein, the shunt comprises a transistor and a diode.

13. A self oscillating fly back circuit as claimed in claim 11 or claim 12 wherein, , the shunt is adapted to operate only when the switching transistor is on.

14. A self oscillating fly back circuit as claimed in any of claims 11 to 13 wherein, the shunt is driven by a current mirror circuit.

15. A self oscillating fly back circuit as claimed in any of claims 11 to 14 wherein, the shunt is driven by a shunt regulator.