WO2009101315A1

WO2009101315A1 - Power weighting of a multicarrier signal on reception in a communication system

Info

Publication number: WO2009101315A1
Application number: PCT/FR2009/050155
Authority: WO
Inventors: Gautier Avril; Fabienne Moulin; Ahmed Zeddam
Original assignee: France Telecom
Priority date: 2008-02-04
Filing date: 2009-02-02
Publication date: 2009-08-20
Also published as: FR2927202A1

Abstract

A reception device (DRa) comprising an analogue-digital converter adapts the power of a multicarrier analogue signal (SA) of OFDM type so as to obtain a summed analogue signal (SAS) whose amplitude is adapted to an amplitude threshold. N bandpass filters (Ft1-FtN) filter the analogue signal respectively into N filtered analogue signals (SAF1-SAFN) each including at least one weighters (Fp1-FpI). N power weighters (Pd1-PdN) weight by N gains respectively the N filtered analogue signals. A summator (SM) sums the weighted filtered analogue signals into the summed analogue signal. The frequency bands of the filters may be fixed and the gains of the weighters may be variable, or vice versa.

Description

Power weighting of a multicarrier signal at reception in a communication system

The present invention relates to a power weighting of a multicarrier signal on reception in a communication system.

More particularly, the present invention relates to a power frequency weighting of a multicarrier signal. For example, the communication system supports a multi-carrier modulation of digital signals according to OFDM orthogonal frequency division multiplexing (Orthogonal Frequency Division Multiplexmg).

With reference to FIG. 1, a known multicarrier digital communication system comprises a transmission device DT and a reception device DR. An OFDM type multicarrier digital communication consists in transmitting an initial digital signal SNI including data in a wide frequency band from the transmission device DT. According to the principle of the OFDM, a modulator MD in the device DT modulates the bits of the signal SNI into complex symbols, for example according to a modulation of amplitude and phase QAM. The symbols are grouped in packets of I successive symbols. The I symbols of each group are put in parallel to form I digital components CNIχ,

CNI ₁ , CNIj distinct input of a synthesizer module Fouπer MI _FFT . The MI _FFT module performs a fast IFFT (Inverse Fast Fourier Transform) Fourier transform, ie a frequency-time conversion, for modulating I carriers with narrow frequency bands of the same width respectively by the I parallel symbols of each group. Each digital component is thus associated with a respective carrier, also called "sub-carrier". In MIFF Module _I ¹ / I modulated carriers undergo parallel-serial conversion to the symbol frequency to be combined into a time-domain digital signal SN T. The latter comprises a digital time-level period of each corresponding symbol has a sum of symbols on the carriers. The digital time signal is converted into an analog broadband multicarrier signal SA_T1 by means of a digital-to-analog converter CNA and is optionally amplified by an amplifier. A coupler CP_T adapted to a transmission channel CT of a communication network connecting a plurality of communication devices between them transmits the amplified analog signal SA_T2 which is transmitted to a reception device DR of a second communication system via the transmission channel CT.

Such a communication network linking a transmission device of a first communication system to a receiving device of a second communication system may be a telephone network, for example of the ADSL ("Asymmetric Digital Subscriber Lme") type, a radio network for example for broadcasting type DAB (in English "Digital Audio Broadcastmg") or for broadcast type DVB (Digital Video Broadcast "in English), or a communication network on power line type CPL The CT transmission channel for each of these examples is then: the copper pair for the telephone network, the air for the radio network and the power lines for the power line systems. During transmission via the transmission channel CT, the transmitted signal SA_T2 generally undergoes disturbances. The received analog signal SA_R1 by the receiving device DR is shaped in a reception coupler CP_R adapted to the channel CT and comprising a low noise amplifier for amplifying the shaped signal. The signal SA_R2 at the output of the coupler CP_R is converted into a digital signal SN R in a CAN-to-digital converter. After having been paralleled on I input channels of a Fourier analyzer module M _F pτ, the latter performs a Fast Fourier Transform (FFT), that is to say a conversion time-frequency, to demodulate the I carrier I symbols parallel to each symbol period and thus recover I digital components CNF] _, CNF ₁ , CNF ₁ . These digital components are put in series at the output of the module M _FFT so as to reconstitute the digital signal of symbols. This is demodulated in a DM demodulator to produce a final digital signal SNF a priori identical to the initial digital signal transmitted SNI.

The analog-to-digital converters CAN have a limited operating voltage range, for example between -1 V and +1 V. If the amplitude of the signal SA R2 coming from the coupler CP R exceeds this voltage range and is for example included in FIG. the voltage range [-3 V, +3 V], the digital signal SN_R is then outputted, thus generating errors conversion. Conversely, if the amplitude of the SA_R2 signal from the coupler is less than the operating voltage range of the CAN converter and is for example in the voltage range [- 0.2 V, +0.2 V], a part the operating range of the CAN converter can not be used, which increases the impact of the quantization noise of the CAN converter. The quantization noise corresponds to the floor noise of the CAN converter. The level of this noise depends on the number of bits of the converter and the voltage range on which it operates.

To overcome this problem, an Automatic Gain Control (AGC) gain controller (AGC) between the output of the receive coupler CP_R and the input of the analog-to-digital converter CAN first adjusts the amplitude of the received analog signal SA_R2 to the operating range of the CAN converter to obtain a suitable signal SA GV to the operating range of the CAN converter. This operation does not modify the signal / noise ratio received, the actual noise received being amplified or attenuated in the same way as the useful signal received over the entire frequency band. The signal-to-noise ratio seen by the system on each carrier i of the received signal SA_R2 can be expressed according to the following relation, with 1 <i <I:

Rsb, = ^C l * ^G CAG

Br ₁ G CAG + Br C, AN

with: - C ₁ : the useful component of the signal on the carrier i,

Br ₁ : the real noise of the signal on the carrier i, induced by the transmission channel CT, ~ GCAG: I ^e coefficient of amplification of the AGC controller, and

- BΓÇAN the noise of the analog-digital converter. By increasing the amplification coefficient GCAG of the controller CAG, the noise Brc _AN ^ u converter becomes negligible.

However, the amplification coefficient G _QAG depends on the total amplitude of the signal SA_R2 received by the coupler. The higher the amplitude of this signal, the less the coefficient G ^ AG is. Laboratory tests on PLC systems have shown that the noise of the CAN converter is mainly greater than the real noise of the CT transmission channel. Most communication systems, because of this limitation of the CAN converter, operate below the capacity of the transmission channel. In addition, when a few carriers have a much higher power level than other carriers, the amplitude of the received signal SA_R2 is largely related to the signal power on these carriers. Consequently, the AGC controller amplifies the received signal less because of the very high power level of these few carriers. This then implies a decrease in the signal-to-noise ratio including on the lower power carriers, and thus a reduction in the rate that it is possible to transmit on these carriers.

The present invention overcomes the drawbacks mentioned above by a method of receiving a multicarrier analog signal by a reception device comprising a converter analog / digital. The method is characterized in that it comprises the steps of:

N simultaneous bandpass filterings of the analog signal respectively in N filtered analog signals each including at least one carrier,

N weightings by N gains respectively of the N filtered analog signals, and a summation of the weighted filtered analog signals into a summed analog signal whose amplitude is adapted to an amplitude threshold

The amplitudes of the carriers of the received signal are thus weighted selectively to be adjusted optimally taking into account the characteristics of the analog-digital converter of the receiving device. These weights are determined to increase the total throughput by decreasing the quantization noise introduced by the analog-to-digital converter during the conversion of the summed analog signal. The invention makes it possible, without increasing the complexity of the remainder of the system, to increase the real data rate of the data signal.

According to a first embodiment of the invention, the carriers of the analog signal have frequencies selected by the N simultaneous bandpass filtering as a function respectively of N fixed frequency bands in order to produce the N filtered signals, and the N gains applied respectively to the filtered signals are variable and determined according to a maximum total bit rate of the summed analog signal and the amplitude of the summed analog signal adapted to the amplitude threshold.

According to a second embodiment of the invention, the carriers of the analog signal have frequencies selected according to N simultaneous bandpass filtering as a function respectively of N bands of variable frequency determined according to respective fixed gains to be applied to the filter signals, the maximum total rate of the sum analog signal and the amplitude of the analog signal sum adapted to the amplitude threshold.

The invention also has forcget a device for receiving a multicarrier analog signal comprising an analog / digital converter. The device is characterized in that it further comprises:

N bandpass filters for filtering the analog signal respectively in N analog filter signals each including at least one carrier, N power weighting means for weighting N respectively the N analog filter signals, and an adder for summing the weighted analog filter signals into an analog signal sum whose amplitude is adapted to an amplitude threshold.

According to a first embodiment of the device for weighting, the N bandpass filters are capable of filtering the carriers of the analog signal as a function respectively of N fixed frequency bands in order to produce the N filter signals, the N powerweighters respectively have N gains. variables to be applied to the filter signals. The device further comprises a gain controller for determining the N variable gains as a function of a maximum total rate of the analog sum signal and the amplitude of the analog sum signal adapted to the amplitude threshold.

According to a second embodiment of the device for weighting, the N bandpass filters are able to filter the carriers of the analog signal by function respectively of N variable frequency bands. The device further comprises a filter controller for determining the N variable frequency bands according to respective fixed gains to be applied respectively to the filter signals by the N power weighting (Pd] _- Pd ^), the maximum total throughput of the filter. sum signal and the amplitude of the analog signal sum adapted to the amplitude threshold.

Other characteristics and advantages of the present invention will appear more clearly on reading the following description of several embodiments of the invention given by way of non-limiting examples, with reference to the corresponding appended drawings in which:

FIG. 1 is a schematic block diagram of a multicarrier communication system as discussed above; FIG. 2 is a schematic block diagram of a reception device according to a first embodiment of the invention;

FIG. 3 is an algorithm of a variable gain determination process in a gain controller of the reception device; and

- Figure 4 is a schematic block diagram of a receiving device according to a second embodiment of the invention.

According to an example which will be referred to hereafter, the invention is applicable to networks and PLC communication systems, acronym for line carrier current. Nevertheless, the invention can be generalized to all OFDM networks and communication systems (Orthogonal Frequency Multiplexmg division (in English) and also to networks and broadband communication systems of UWB type ("Ultra WideBand" in English).

As shown in FIG. 2 or 4, in the communication system a reception device DRa according to a first or second embodiment of the invention comprises, in a similar manner to the known reception device shown in FIG. 1, a coupler CP receiving since the CT transmission channel an analog multi-carrier signal of the OFDM type, an analog-digital converter ADC, a quick Fouπer transformation module M FT ^ ^e t a demodulator DM. According to the invention, a power weighting device DPP is interconnected between the output of the CP coupler and the input of the CAN-to-digital converter. The device DPP processes at the output of the coupler CP the analog multicarrier signal SA which has possibly undergone in the coupler CP a low gain amplification. The analog signal SA comprises I carriers Pt ^ a Ptj which have been subjected to modulations of different amplitudes and phases. Each carrier Pt ₁ , with 1 <i <I, corresponds to at least a component of the digital signal associated with a carrier frequency Fp ₁ .

The power weighting device DPP, according to the two embodiments of the invention, comprises a power distributor RP receiving the analog signal SA, N selection stages Ft _n -Pd _n each comprising a bandpass filter Ft _n and a weighting device of power Pd _n , with 1 <n ≤ N, and an adder SM to obtain at the output of the DPP device an analog signal sum SAS with carriers having weighted powers. The entities RP, Fti to Ft _N , Pd ₁ to Pd _N and SM are represented in the form of functional blocks, most of which provide functions relating to the invention and may correspond to software and / or hardware modules.

The power distribution unit RP distributes the power of the analog signal SA which delivers by the coupler CP in N analog signals SA _{1 to} SA ₁ respectively to the inputs of the N stages of selection. Each analog signal SA _n comprises the I modulated carriers Ptχ-Ptj of the signal SA.

In each selection stage Ft _n -Pd _n , the bandpass filter Ft _n selects in the respective analog signal SA _n one or more carrier frequencies according to a selection relating to the first embodiment of the invention, or to the second embodiment of the invention described later. A filter signal SAF _n at the output of the filter FT _n only includes the carriers relative to the frequencies selected by the filter. The power weighting device Pd _n applies to the filter signal SAF _n a variable gain GV _n attenuating or amplifying the power of the carriers in the signal SAF _{n in} order to adjust the amplitude of the signal SAF _n as a function of the amplitudes of the other filter signals SAFχ. SAF _N and as a function of a LIM _CAN amplitude limit of the _CAN converter. To increase the overall throughput of the channel, the gain applied to a signal SAF _n high power is less than the gain applied to a signal SAF _n lower power. Each power weighting instrument Pd _n provides an SAP _n weighted signal whose amplitude of the selected carriers has been adjusted. The summator SM sum the weighted filter signals SAP ₁ to SAP _N into a single signal sum SAS comprising all the carriers Pti ~ Pt _N. At the output of the power scaling device DPP, the sum analog signal SAS provides a maximum total bit rate and has an amplitude smaller than the amplitude limit LIMQ ^ N ^ U CAN converter to which the SAS signal is applied to be converted into a signal numeric SN_R.

With reference to the first embodiment illustrated in FIG. 2, each bandpass filter Ft of the weighting device selects the carriers of the signal SA _n having frequencies comprised in a respective fixed frequency band delimited by a low cutoff frequency Fcb _n and a high cutoff frequency Fch _n . The frequency bands of the bandpass filters Fti to Ft are substantially disjoint and contiguous two by two and together cover the frequencies of the carriers Pt _{1 to} Pt ₁ . For example, the DPP device operates in a frequency band between 2 MHz and 28 MHz and comprises 5 bandpass filters Fti to Ft ₅ having cut-off frequencies distributed according to the following table:

As a variant, the power distributor RP and the N bandpass filters Ft ^ a Ft ^ are replaced by a frequency divider dividing the wide signal. band received in N narrow-band signals (Fcb] _, FCh ₁ ) a (Fcb _N , Fch _N ).

Each carrier Pt ₁ of the signal SA _n at the input of the filter Ft _n is associated with a selection coefficient Qn i equal to 1 if the frequency Fp ₁ of the corresponding carrier is in the respective fixed frequency band (Fcb _n , Fch _n ). In the opposite case, if Fp ₁ <Fcb _n or Fch _n <Fp ₁ , then the selection coefficient associates Q _{n ^ 1} is 0. Only the carriers of the signal SA _n whose frequencies are associated with equal selection coefficients a 1 are present in the filter signal SAF _n provided by the filter Ft _n and are processed by the power weighting instrument Pd _n . The other carriers are not processed by the weighting instrument Pd _n .

A variable gain GV _n is applied to the filter signal SAF _n in the power weighting instrument Pd _n . All the weighting modulators Pd ₁ to Pd _N of the N stages of selection are connected by a gain control bus to a gain controller CG which determines the variable gains GVi to GV _N as a function of a maximum total flow rate D _Tmax and the amplitude amp (P _τ ) of the analog signal sum SAS to obtain at the output of the device DPP. The gain controller CG is connected to or incorporates an MR memory able to record in particular all the variable gains GV _] _ a GV _N determmines.

With reference to FIG. 3, a method for determining the variable gains GV 1 to GV _N relative to the first embodiment of the invention and implemented in the gain controller CG comprises steps E 1 to E 12. The process successively assigns each of J predetermined gain values ai a otj to the gain of each of the power weighting Pd] _ a Pd _N so that all combinations of predetermined gain values are tested by the gain controller CG. The method comprises N recursion loops Bx a B _N included in each other and respectively relating to the variations of the power factor weight gain Pdi to Pd _N. In the last loop executed B _N included in all the other loops, values of the variable gains GV] _ a GV _N are selected for which the total rate D _τ of the SAS signal at the output of the device DPP is maximum and for which the amplitude of the SAS signal is less than the LIMCAN amplitude limit of the CAN converter. To simplify the representation of the algorithm of the proceeds, in FIG. 3, only 3 recursion loops Bi, B _n and B _N among the N loops are represented. A first recursion loop B1 including all the other loops corresponds to the steps El a ElI and tests the J possible values ai a αj for a gain Gi = CCKI of the first power weighting Pdi with the index Ki initially at the value 1 and incremente at each recurrence of the loop Bi from one unit to the number J. The two other recursion loops relating to an intermediate recurrence loop B _n to the steps E2 to ElO and to a last recurrence loop B _N in the steps E3a E9 test respectively for the gain G _n = a ^ _n of the power weighting Pd _n with 1 <K _n <J and for the gain ^ N ^{= α} _KN of the power weighting Pd _N with 1 <K _N <J, the J predetermined gain values. Each recurrence loop Bi, B _n , B _N is repeated as long as the index Ki, K _n , K _N associates is not equal to J + 1. The last recurrence loop B _N comprises the steps E4 to E8 corresponding to the determination of the variable gains GVi to GV _N according to conditions specific to the step E7.

At an initial step EO, the gain controller

CG initializes in zero in the MR memory the variable gains GVi to GV _N , and the maximum total flow D _Tmax of the analog signal summed SAS, used later during the execution of the process.

In step E4 in the last recurrence loop B _N , the gain controller CG assigns the predetermined gain values αjQ, ... αj <; _n , ... α ^^ respectively to the gains Gi ... G _n , ... G _N relative to the N stages of selection, with α] _ <ακi ≤ otj, ... ai - ^α Kn - ^α J f - ^has the - KN ^α - ^α J These gains G to G _N are stored in the memory MR.

In step E5, the controller CG determines the total signal-to-noise ratio Rsb _x of each carrier Pt ₁ , by adding the signal to noise ratios Rsb _{± / 1} to Rsb _N , _x of the carrier Pt ₁ determined respectively for each stage. selection Ft _n -Pd _n according to the gain G _n assigned to step E4.

The total signal-to-noise ratio RSb ₁ for the carrier i is determined according to the following relation:

NOT

n-1 with

PS * G * Ω - RSb _n ,, ≈ = ^r - ^

PB _n G _r Qn, i + PE

N: the number of selection stages of the power weighting device DPP,

PS _x : the power of the signal on the carrier Pt ₁ , PB ₁ : the power of the noise on the carrier Pt ₁ ,

PE ₁ : the sampling noise power of the CAN converter for the carrier Pt ₁ ,

- G _n : the gain for the selection stage Ft _n -Pd _n , and - Q _nr χ the selection coefficient of the bandpass filter Ft _n for the carrier Pt ₁ .

The gain controller CG also determines the total rate D ₁ of a carrier Pt ₁ as a function of the total ratio RSb ₁ obtained.

The gain controller CG sum all the determined rates 0χ to Dj of the carriers Pt] _ to Ptj to determine the total bit rate O ^ of the signal, ie: i D _τ = ΣD _X. i = l

In step E6, the gain controller CG determines the total power P <p of the signal SAS by summing the total power Pp _t x of each carrier Pt ₁ defined in the N stages of selection as a function respectively of the gains Gx to G ^ assigned to step E4 according to the following relation:

I I N

P _τ = Σ P _Ptl = Σ Σ PS, * G _n * Q _{n / 1} . i - li = ln = l

The gain controller CG applies an amp function to the total power P i previously determined in order to obtain the amplitude of the summed signal SAS as a function of the gains G x at G i attributed to the step E 4.

In step E7, if the total flow rate D _τ is greater than the maximum total flow D <p _max and if the amplitude amp (PT) of the signal SAS is less than the amplitude limit LIMQAN of the converter CAN, then the controller of gain awards to all winnings variables GV] _ GV _N respectively the values Gx a GJ _^ J previously allocated in step E4 and stored in the memory MR CG controller. The gain controller also assigns the maximum total bit rate D _Tmax _τ D the total flow rate determined in step E5. The variable gains GV] _ a GV _n and the flow rate Dχ _max are memorized in the memory MR by overwriting the old values.

As soon as all the combinations of the predetermined gain values in the N recursion loops have been tested, that is, if K i = J + 1, K _n = J + 1 and K _N = J + 1, the Variable gain values GV] _ a GV _N are considered definitively determined for a maximum total rate and a signal amplitude less than the amplitude limit of the CAN converter, and are stored in the MR memory.

At each selection stage Ft _n -Pd _n , the power weighting device Pd _n applies to the filter signal SAF _n the variable gain GV _n stored in order to obtain a weighted filter signal SAP _n .

As a variant, the gain weightings Pd] _ a Pd _N and the gain controller GC are replaced in the selection stages Ftχ-Pdχ to Ft _N -Pd _N respectively by automatic gain controllers. The automatic gain controller in each selection stage Ft _n -Pd _n automatically adjusts the amplitude of the respective filter signal SAF _n comprising only the carriers selected by the bandpass filter FT _n . The N automatic gain controllers have in common an amplitude adjustment threshold which must not be exceeded by the N powers of the SAP signals at SAP _N. This amplitude threshold may be equal to the amplitude limit of the CAN converter. If the amplitude threshold does not correspond to the amplitude limit LIMÇAN ^an automatic gain controller AGC is placed between the adder SM and the ADC converter to adjust the amplitude of the summed signal according to amplitude SAS LIMC _AN .

With reference to the second embodiment of the invention illustrated in FIG. 4, the gains GF] _ GFjvj of the power weighting Pd] _ to Pd ^ are fixed and the passbands of the bandpass filters are variable in order to select in each selection stage Ft _n -Pd _n of the signal carriers SA _n having substantially equal powers in a predetermined tolerance interval. The frequency bands of the bandpass filters F 1 to F 1 are substantially disjoint and contiguous two by two and together cover the carrier frequencies Pti to Pt ₁ . The low and high variable cutoff frequencies FVcbi and FVCh ₁ , ... FVcb _n and FVCh _{n ^} ... FVcb _N and FVch ^ bounding a variable passband of the bandpass filter Fti ... Ft _n , ... Ft ^ are determined by a filter controller CF connected to all bandpass filters Fti to FT _N.

Each bandpass filter Ft _n selects the carriers of the signal SA _n having frequencies within its bandwidth determined by the filter controller CF. Each carrier Pt ₁ of the signal SA _n at the input of the filter Ft _n is associated with a selection coefficient Qn, i equal to 1 if the corresponding carrier frequency Ft ₁ is included in the passband of the filter. In the opposite case, the associated selection coefficient Q _n , i ^is zero. Only the filtered carriers in the SA signal _n associated with a selection coefficient equal to 1 are processed in the selection stage Ft _n -Pd _n and are present in the filter signal SAF _n provided by the filter Ft _n . The other components are not processed.

The fixed gain GF _n in the power weighting Pd _n is a priori different from the fixed gains in the other power weightings and is applied to the filter signal SAF _n in the power weighting Pd _n . According to its predefined value, the fixed gain GF _n serves to weight, that is to say to amplify or attenuate the amplitude of the filter signal SAF _n and thus the power of the selected carriers. For high power, the gain GF _n attenuates the amplitude of the signal SAF _n . Conversely, for a low power, the gain GF _n amplifies the amplitude of the signal SAF _n .

Since the gains of the selection stages are fixed, the weighted signals SAPi ^a SAP ^ have substantially equal amplitudes, but are not adjusted according to the amplitude limit LIM _Q AN of the digital analog converter CAN. An automatic gain controller AGC is interconnected between the output of the summator SM and the input of the digital analog converter CAN in order to adjust the power of the total signal sum SAS as a function of the amplitude limit LIM _CAK [of the CAN converter.

The method of determining the bandpass filter cut-off frequencies is implemented in the filter controller CF and is analogous to the variable gain determination process, and successively assigns each of predetermined cutoff frequency values to the frequencies. to cut each of the bandpass filters Fti to Ft ^ so that all combinations of predetermined cutoff frequency values are tested by the CG gain controller. The method for determining the cut-off frequencies also comprises N recursive loops included in each other and relating respectively to the variations of the cut-off frequencies of the band-pass filters Ft] _ a Ft ^. In the last loop executed included in all the other loops, values of the pairs of distinct cut-off frequencies bounding the passbands of the filters Fti to Ft _N are selected to obtain a maximum total bit rate D _τ = D _Tmax and an amplitude amp (Pχ ) of the SAS signal less than the _{CAN CAN} amplitude limit of the _CAN converter. The flow rate and the amplitude are determined in particular according to the predefined fixed gains of the N stages of selection and the powers of the filtered carriers in each selection stage according to the variable cutoff frequencies tested.

The selective power weighting device DPP can also operate by combining the two previously described embodiments comprising for each selection stage a bandpass filter having a variable frequency band and a power weighting applying a variable gain.

The selective power weighting device DPP may be integrated in the receiving device of the communication system as present in the preceding examples, or with an input coupler be distinct from it.

Claims

1 - Method for receiving a multicarrier analog signal (SA) by a receiving device (DRa) having an analog-to-digital converter (ADC), implementing:

Simultaneous bandpass filtering of the analog signal respectively in N analog filter signals (SAFi-SAF) each including at least one carrier (Fp] _-Fpχ),

N weightings by N gains (GV ^ -GV _N , GFχ-GF _N ) respectively of the N analog filter signals, and a summation of the weighted analog filter signals into an analog signal sum (SAS), characterized in that it sets in addition: a determination of one of the characteristics of the filterings and weightings as a function of a maximum total flow rate (Dτ _max ) of the sum analog signal (SAS) and of the amplitude (amp (PiO) of the analog signal sum adapted to an amplitude limit (LIM ^ N) of the analog-to-digital converter (ADC).

2 - Method according to claim 1, wherein the carriers (Fpχ-Fpχ) of the analog signal

(SA) have frequencies selected by the N simultaneous bandpass filtering as a function respectively of N fixed frequency bands to produce the N filter signals (SAFχ-SAF _N ), and the N gains (GVχ-GV _N ) applied respectively to the filter signals are variable and determined according to the maximum total flow (D _Tmax ) of the sum analog signal (SAS) and the amplitude (amp (Pτ)) of the analog signal sum adapted to the amplitude limit ( _CAN LIM) . 3 - Method according to claim 1, wherein the carriers (Fpχ-Fpj) of the analog signal (SA) have frequencies selected according to N simultaneous bandpass filtering respectively as a function of N variable frequency bands determined according to respective gains fixed (GF] _- GFN) to apply on the filter signals (SAF- SAF _N ), the maximum total flow rate (Oτmaχ) of the analog signal sum (SAS) and the amplitude (amp (P _τ )) of the analog signal sum adapted to the amplitude limit (LIMς _AN ).

4 - Reception device (DRa) of a multicarrier analog signal (SA) comprising an analog-to-digital converter, N bandpass filters (Ftχ-Ft _N ) for filtering the analog signal respectively in N analog filter signals (SAFI-SAFN ) each including at least one carrier (Fp ₁ -Fp ₁ ),

N power weighting (Pdi-Pd ^) for weighting by N gains (GVχ-GV _t j, GFχ-GF _N ) respectively the N analog filter signals, and a summator (SM) for summing the weighted analog signal signals into a signal analogue sum (SAS), characterized in that it further comprises a means (CG; CF) for determining at least one of the characteristics of the filters and weightings according to a maximum total flow rate (D <r _max ) The analog signal sum (SAS) and the amplitude (amp (Pχ)) of the analog sum signal have to be adapted to an amplitude limit (LIMCAN) of the analog-to-digital converter (ADC). 5 - Device according to claim 4, wherein the N bandpass filters (Fti-Ft ^) are able to filter the carriers (Fpj-Fpj) of the analog signal (SA) as a function respectively of N fixed frequency bands so to produce the N filtered signals (SAF) _- SAFN), the N power weighting units (Pdχ-Pd _N ) have respectively N variable gains (GVJ-GV _N ) to be applied to the filtered signals, and said means (CG) is capable of determining the N variable gains as a function of the maximum total rate (D _Tmax ) of the summed analog signal (SAS) and the amplitude (amp (Pτ)) of the summed analog signal adapted to the amplitude limit (LIMQAN) •

6 - Device according to claim 4, wherein the N bandpass filters (Ftχ-Ft _f j) are able to filter the carriers (Fpχ-Fpi) of the analog signal (SA) as a function respectively of N variable frequency bands , and said means (CF) is able to determine the N variable frequency bands as a function of respective fixed gains (GF 1 -GF 1) to be applied respectively to the filtered signals (SAFI-SAFN) by the N power weighting (Pd) _-Pdjj), the maximum total rate (D _Tmax ) of the summed signal (SAS) and the amplitude (amp (P _τ )) of the summed analog signal adapted to the amplitude limit (LIM ^ AN) •