WO2008057126A1

WO2008057126A1 - Precision sampling circuit

Info

Publication number: WO2008057126A1
Application number: PCT/US2007/000649
Authority: WO
Inventors: Hae-Seung Lee
Original assignee: Cambridge Analog Technology, Llc
Priority date: 2006-11-09
Filing date: 2007-01-10
Publication date: 2008-05-15

Abstract

A sampling circuit includes an input voltage source; a first switch having an input operatively connected to the input voltage source; a sampling capacitor operatively connected to an output of the first switch; an operational amplifier having an inverting input operatively connected to the sampling capacitor; a second switch operatively connected across the inverting input of the operational amplifier and an output of the operational amplifier; and a second capacitor operatively connected to the output of the first switch. The first switch has a variable parasitic capacitance, and the second capacitor has a substantially more linear capacitance than the variable parasitic capacitance and is in parallel with the variable parasitic capacitance. A combined variable parasitic capacitance and capacitance of said switch capacitor is more linear than the variable parasitic capacitance of the first switch.

Description

PRECISION SAMPLING CIRCUIT

PRIORITY INFORMATION

[0001] The present invention claims priority to U.S. Utility Application Serial No. 11/558,114, filed on November 9, 2006. The entire contents of 11/558,114 are incorporated herein by reference in its entirety.

FIELD OF THE PRESENT INVENTION

[0002] The present invention relates generally to a sampling circuit for switched-capacitor filters, analog-to-digital converters, and delta-sigma modulators. More particularly, the present invention relates to a ground-side sampling circuit which produces a removable constant channel charge error.

BACKGROUND OF THE PRESENT INVENTION

[0003] Most switched-capacitor analog circuits such as switched-capacitor filters, analog-to-digital converters, and delta-sigma modulators require precise sampling of analog voltages on a capacitor. The charge sampled on the sampling capacitor must be a precise- linear function of the voltage that has been sampled. A basic sampling circuit is illustrated in Figure 1.

[0004] The MOS transistor Mi is operated as a switch. When the gate voltage Vg is high, MOS transistor Mi is turned ON, and the output voltage V_out is equal to the input voltage Vm. The charge, q, on the sampling capacitor CL is equal to

V,_nC_L.

[0005] When V₉ goes low, MOS transistor Mi turns OFF, and sampling capacitor CL is isolated from the input. If MOS transistor Mi were an ideal switch, the charge, q, sampled on sampling capacitor CL would remain unaltered. However, the MOS transistor Mi is not an ideal switch and thus injects MOS transistor Mi charge onto sampling capacitor C_L.

[0006] Part of the injected charge is from the channel charge, and the rest is due to capacitive coupling from the gate terminal of MOS transistor Mi to the output node. The capacitive coupling generally gives a constant offset error and does not give rise to nonlinearity. The channel charge in MOS transistor Mh splits into two components, qi and q₂, when MOS transistor M₁ is turned OFF. The component q₂ causes an error with respect to the charge, q, on the sampling capacitor C_L. In other words, the charge, q, on the sampling capacitor C_L is equal to Vj_nCi. + q2. [0007] The channel charge in MOS transistor Mi is a function of the input voltage because the gate-to-source voltage V_gs of MOS transistor Mi is equal to V₉ - V_1n.

[0008] It is well known that the channel charge of a MOS transistor is a nonlinear function of the gate-to-source voltage. Since the channel charge is a nonlinear function of the input voltage, the resulting charge error is a nonlinear function of the input voltage.

[0009] Conventionally, ground-side sampling, as illustrated in Figure 2, removes the input dependence of the charge injection in the first order. In the sampling circuit, shown in Figure 2, two MOS transistors are employed, the source-side transistor Mh and the ground-side transistor M₂. When both transistors, M₁ and M₂, are turned ON, the input voltage Vm is applied across the sampling capacitor CL- The sample is taken when the gate voltage V_g2 of the ground-side transistor M₂ is lowered, thereby turning ground-side transistor M₂ OFF. The channel charge in ground-side transistor M₂ splits into two components as before, q-i and q₂, when ground-side transistor M₂ is turned OFF. The component q₂ causes an error with respect to the charge, q, on the sampling capacitor C_L- In other words, the charge, q, on the sampling capacitor C_L is equal

[0010] In contrast to the circuit of Figure 1 , the channel charge in ground-side transistor M₂ is independent of the input voltage Vi_n, at least to the first order. This is because the source and the drain voltages are at ground potential when ground- side transistor M₂ is ON. Thus, if the ratio of the channel charge split between q-i and q₂ is constant, the charge error, q_2> will be constant rather than a nonlinear function of the input voltage. Such a constant offset error can be readily removed or minimized.

[0011] For high accuracy application, even a small amount of nonlinearity in q₂, due to second order effects, is often a limiting factor. For example, the channel charge in ground-side transistor M₂ may be dependent of the input voltage Vm due to a second order effect. [0012] The second order effect is due to the impedance variation in source-side transistor M-i. The ON resistance of source-side transistor Mi varies with the input voltage Vi_n. In addition, the parasitic capacitance associated with source-side transistor Mi is a nonlinear function of the input voltage Vi_n. [0013] In Figure 3, the variable ON resistance of source-side transistor Mi is shown as RON, and the variable parasitic capacitance Cp. As the gate voltage V_g2 is lowered to turn OFF ground-side transistor M₂, the channel charge splits into q₁ and q₂. This process is not instantaneous, but takes a finite amount of time on the order of the transit time of carriers in ground-side transistor M₂. [0014] As q₂ leaves ground-side transistor M₂, current, corresponding to i=dq2/dt, flows into the network consisting of C_L, RON, and C_P. This current creates a time-dependent voltage at the drain node of ground-side transistor M₂, in turn creating an electric field between the drain and the source of ground-side transistor M₂. This effect alters the split ratio between qi and q₂. Since the time-dependent voltage on the drain of ground-side transistor M₂ is a function of the composite impedance given by CL, RON, and CP, the charge split ratio between qi and q₂ is dependent on the input voltage Vi_n. Since RON, and Cp are nonlinear functions of the input voltage V_in, the injected charge q₂ is a nonlinear function of V_1n. [0015] Although source-side transistor Mi and ground-side transistor M₂ are shown as NMOS transistors in Figure 2, a parallel connection of NMOS and PMOS transistors, commonly referred to as a complementary switch, is often conventionally employed. The complementary switches somewhat alleviate the nonlinear charge injection, but not to a satisfactory extent with respect to utilization in high accuracy circuits. [0016] It is noted that if the time constant, RONCP, is much faster than the carrier transit time, the impedance presented by RON dominates within the time scale of the charge injection. In this situation, the instantaneous incremental voltages v-i and v₂ at Nodes 1 and 2, respectively, are Vi = (dq₂/dt)RoN and V₂ = Vi + (q₂/C_L) = (oq₂/ot)RoN + (q2/C_L). [0017] On the other hand, if the time constant, RONCP, is much slower than the carrier transit time, the impedance presented by C_P dominates within the time scale of the charge injection. In this situation, the instantaneous incremental voltages Vi and V₂ at Nodes 1 and 2, respectively, are V₁ = q2/C_P and V₂ = V₁ + (q2/C_L) = qj>/Cp + q₂/C_L = q₂(1/ C_P + 1/ C_L). [0018] In either situation,- the instantaneous changes in V₂ create an electric filed across ground-side transistor M₂- This effect alters the split ratio between qi and qz in turn affecting the magnitude of q₂- Since V₂ is nonlinearly dependent on the input voltage Vm through either R_ON or Cp, the resulting <\₂ is also a nonlinear function of the input voltage Vi_n.

[0019] Therefore, it is desirable to provide a sampling circuit that accurately samples an input voltage without suffering from the nonlinear error introduced by charge injection. It is desirable to provide a sampling circuit that accurately samples an input voltage without suffering from the nonlinear error introduced by charge injection and provides differentia! signal paths for sampled-data circuits.

Furthermore, it is desirable to provide a sampling circuit that reduces the effect of power supply, substrate, and common-mode noise by symmetric differential signal processing. Also, it is desirable to provide a sampled-data circuit that increases the signal range by incorporating differential signal paths.

SUMMARY OF THE PRESENT INVENTION

[0020] One aspect of the present invention is a sampling circuit. The sampling circuit includes an input voltage source; a first switch having an input operatively connected to the input voltage source; a sampling capacitor operatively connected to an output of the first switch; a second switch having an input operatively connected to the sampling capacitor; and a second capacitor operatively connected to the output of the first switch.

[0021] Another aspect of the present invention is a sampling circuit. The sampling circuit includes an input voltage source; a first switch having an input operatively connected to the input voltage source; a sampling capacitor operatively connected to an output of the first switch; an operational amplifier having an inverting input operatively connected to the sampling capacitor; a second switch operatively connected across the inverting input of the operational amplifier and an output of the operational amplifier; and a second capacitor operatively connected to the output of the first switch.

[0022] A further aspect of the present invention is a sampling circuit. The sampling circuit includes a first input voltage source; a first switch having an input operatively connected to the first input voltage source; a first sampling capacitor operatively connected to an output of the first switch; a differential amplifier having an inverting input operatively connected to the first sampling capacitor; a second switch operatively connected across the inverting input of the differential amplifier and a non-inverting output of the differential amplifier; and a second capacitor operatively connected to the output of the first switch.

BRIEF DESCRIPTION OF THE DRAWINGS

[0023] The present invention may take form in various components and arrangements of components. The drawings are only for purposes of illustrating a preferred embodiment and are not to be construed as limiting the present invention, wherein: [0024] Figure 1 illustrates a prior art sampling switch;

[0025] Figure 2 illustrates another prior art sampling switch;

[0026] Figure 3 illustrates a circuit model of the prior art sampling switch illustrated in Figure 2;

[0027] Figure 4 illustrates a sampling switch according to the concepts of the present invention;

[0028] Figure 5 illustrates a closed-loop sampling switch according to the concepts of the present invention;

[0029] Figure 6 illustrates a fully-differential closed-loop sampling switch according to the concepts of^'the present invention; [0030] Figure 7 illustrates another fully-differential closed-loop sampling switch according to the concepts of the present invention;

[0031] Figure 8 illustrates a third fully-differential closed-loop sampling switch according to the concepts of the present invention;

[0032] Figure 9 illustrates fourth fully-differential closed-loop sampling switch according to the concepts of the present invention;

[0033] Figures 10-12 illustrates further fully-differential closed-loop sampling switches according to the concepts of the present invention.

DETAILED DESCRIPTION OF THE PRESENT INVENTION [0034] The present invention will be described in connection with preferred embodiments; however, it will be understood that there is no intent to limit the present invention to the embodiments described herein. On the contrary, the intent is to cover all alternatives, modifications, and equivalents as may be included within the spirit and scope of the present invention, as defined by the appended claims. [0035] For a general understanding of the present invention, reference is made to the drawings. In the drawings, like reference have been used throughout to designate identical or equivalent elements. It is also noted that the various drawings illustrating the present invention may not have been drawn to scale and that certain regions may have been purposely drawn disproportionately so that the features and concepts of the present invention could be properly illustrated. [0036] It is noted that, in the various Figures, the earth symbol indicates the system's common-mode voltage. For example, in a system with 2.5 V and -2.5 V power supplies, the system's common-mode voltage may be at ground. In a system with a single 2.5 power supply, the system's common-mode voltage may be at 1.25 V.

[0037] As noted above, Figure 4 illustrates an example of a sampling circuit. As illustrated in Figure 4, a capacitor Ci₁ which is substantially more linear than a variable parasitic capacitance Cp of a source-side transistor Mi, is connected between Node 1 and a constant voltage. The constant voltage may be ground. In this embodiment, capacitor C₁ and variable parasitic capacitance C_P are effectively in parallel and the capacitance value of capacitor Ci may be significantly larger than the variable parasitic capacitance C_P. In this situation, the instantaneous incremental voltages V₁ and V₂ at Nodes 1 and 2, respectively, are V₁ = q₂/(Cp+C-t) and

qa((1/ (Cp+d)) + 1/ C_L).

[0038] The addition of the capacitor Ci to the sampling circuit has several effects. First, the effect of input dependent R_ON can be greatly reduced by making the time constant, Ro_N(Cp+C₁),much longer than the carrier transit time. Second, the combined capacitance (Cp+Cι ) is much more linear than Cp. Third, the magnitude of the voltage V₂ is reduced by the addition of C-j, further reducing the dependence of charge split ratio on the input voltage Vi_n. Since C-i is much larger than Cp, vi is much smaller and less dependent on the input voltage V_in. Therefore, it follows that V₂ is much less dependent on Vt_n, giving a substantially constant split ratio between q! and q₂. Furthermore, the sampling circuit of Figure 4 reduces the effect of substrate noise.

[0039] Figure 5 illustrates an example of a closed-loop sampling circuit. As illustrated in Figure 5, a sampling switch M₂, when turned ON, connects the inverting input and the output of an operational amplifier 3. With an input switch M₁ turned ON₁ a first plate 4 of sampling capacitor CL receives the input voltage Vin, while a second plate 5 of the sampling capacitor C_L settles to an offset voltage of the operational amplifier 3. A capacitor Ci, which is substantially more linear than a variable parasitic capacitance C_P of the input switch Mi, is connected between Node 1 and a constant voltage. The constant voltage may be ground. [0040] In this embodiment, capacitor Ci and variable parasitic capacitance C_P are effectively in parallel and the capacitance value of capacitor Ci may be significantly larger than the variable parasitic capacitance Cp. When the sampling switch M₂ is turned OFF, charge is sampled on the capacitor C_L. The non-linear input dependent charge injection is greatly reduced in the similar manner to the embodiment of Figure 4. The closed-loop sampling of Figure 5 also removes the effect of the offset voltage associated with the operational amplifier 3, thereby enabling the closed-loop sampling circuit of Figure 5 to be appropriate for analog- to-digital converters and other precision sampled-data circuits. Furthermore, the sampling circuit of Figure 5 reduces the effect of substrate noise. [0041] Figure 6 illustrates an example of a fully differential closed-loop sampling circuit being employed in order to improve power supply and substrate rejection.

As illustrated in Figure 6, a sampling switch M₂, when turned ON, connects the inverting input and the non-inverting output of a differential amplifier 30. With an input switch Mi turned ON, a first plate 4 of sampling capacitor Cu receives the input voltage Vi_n+, while a second plate 5 of the sampling capacitor Cu settles to an offset voltage of the differential amplifier 30. A capacitor Ci, which is substantially more linear than a variable parasitic capacitance C_PI of the input switch M₁, is connected between Node 1 and a constant voltage. The constant voltage may be ground. [0042] Moreover, as illustrated in Figure 6, a sampling switch M₄, when turned ON, connects the non-inverting input and the inverting output of a differential amplifier 30. With an input switch M₃ turned ON, a first plate 6 of sampling capacitor C|_₂ receives the input voltage Vj_n., while a second plate 7 of the sampling capacitor Cι_₂ settles to a common-mode voltage - of the differential amplifier 30. A capacitor C₂, which is substantially more linear than a variable parasitic capacitance CP3 of the input switch M₃, is connected between Node 10 and a constant voltage. The constant voltage may be ground. [0043] In this embodiment, capacitor Ci and variable parasitic capacitance Cpi are effectively in parallel and the capacitance value of capacitor Ci may be significantly larger than the variable parasitic capacitance CPI. When the sampling switch M₂ is turned OFF, charge is sampled on the capacitor Cu. The non-linear input dependent charge injection is greatly reduced in the similar manner to the embodiment of Figure 5.

[0044] Moreover, capacitor C₂ and variable parasitic capacitance Cp₃ are effectively in parallel and the capacitance value of capacitor Ci may be significantly larger than the variable parasitic capacitance Cpa. When the sampling switch M₄ is turned OFF, charge is sampled on the capacitor CL2. The non-linear input dependent charge injection is greatly reduced in the similar manner to the embodiment of Figure 5. [0045] The closed-loop sampling of Figure 6 also removes the effect of the offset voltage associated with the differential amplifier 30, thereby enabling the fully differential closed-loop sampling circuit of Figure 6 to be appropriate for analog-to-digital converters and other precision sampled-data circuits. Furthermore, the sampling circuit of Figure 6 reduces the effect of substrate noise. [0046] Figure 7 illustrates another example of a fully differential closed-loop sampling circuit being employed in order to improve power supply, substrate rejection, and removal of a differential component of the nonlinear input dependent charge injection. As illustrated in Figure 7, a sampling switch M₂, when turned ON, connects the inverting input and the non-inverting output of a differential amplifier 30. With an input switch Mi turned ON, a first plate 4 of sampling capacitor Cu receives the input voltage Vi_n, while a second plate 5 of the sampling capacitor Cu settles to a common-mode of the differential amplifier 30.

[0047] Moreover, as illustrated in Figure 7, a sampling switch M4, when turned ON, connects the non-inverting input and the inverting output of a differential amplifier 30. With an input switch M₃ turned ON, a first plate 6 of sampling capacitor Cι_₂ receives the input voltage Vi_n, while a second plate 7 of the sampling capacitor Cι_₂ settles to a common-mode voltage of the differential amplifier 30. A capacitor Ci , which is substantially more linear than a variable parasitic capacitance Cp₃ of the input switch M₃ or a variable parasitic capacitance CPI of the input switch M-t, is connected between Node 1 and Node 10. [0048] In this embodiment, an effective capacitance, twice the value of Ci, and variable parasitic capacitance Cpi are effectively in parallel and the capacitance value of capacitor Ci may be significantly larger than the variable parasitic capacitance C_Pi. When the sampling switch M₂ is turned OFF, charge is sampled on the capacitor Cι_i. The non-linear input dependent charge injection is greatly reduced in the similar manner to the embodiment of Figure 5. [0049] Moreover, an effective capacitance, twice the value of Ci, and variable parasitic capacitance C_P3 are effectively in parallel and the capacitance value of capacitor C₁ may be significantly larger than the variable parasitic capacitance C_P3. When the sampling switch M₄ is turned OFF, charge is sampled on the capacitor CL2- The non-linear input dependent charge injection is greatly reduced in the similar manner to the embodiment of Figure 5.

[0050] The closed-loop sampling of Figure 7 also removes the effect of the offset voltage associated with the differential amplifier 30 and removes the differential component of the nonlinear input dependent charge injection associated with the differential amplifier 30, thereby enabling the fully differential closed-loop sampling circuit of Figure 7 to be appropriate for analog-to-digital converters and other precision sampled-data circuits. Furthermore, the sampling circuit of Figure 7 reduces the effect of substrate noise. [0051] Figure 8 illustrates a third example of a fully differential sampling circuit being employed in order to improve power supply, substrate rejection, and removal of a differential component of the nonlinear input dependent charge injection. As illustrated in Figure 8, when an input switch M₁ is turned ON, a first plate 4 of sampling capacitor Cu receives the input voltage Vi_n+. A sampling switch M₂, when turned ON, connects a second plate 5 of the sampling capacitor Cu to a constant voltage, preferably a common-mode voltage.

[0052] Moreover, as illustrated in Figure 8, when an input switch M₃ is turned ON, a first plate 6 of sampling capacitor Ci_₂ receives the input voltage Vi_n-. A sampling switch IVI₂, when turned ON, connects a second plate 7 of the sampling capacitor C_L2 to a constant voltage, preferably a common-mode voltage. A capacitor C_L which is substantially more linear than a variable parasitic capacitance Cp₃ of the input switch M₃ or a variable parasitic capacitance Cp₁ of the input switch Mi, is connected between Node 1 and Node 10. [0053] In this embodiment, an effective capacitance, twice the value of C₁, and variable parasitic capacitance Cp₁ are effectively in parallel and the capacitance value of capacitor Ci may be significantly larger than the- variable parasitic capacitance C_P1. When the sampling switch M₂ is turned OFF, charge is sampled on the capacitor Cu. The non-linear input dependent charge injection is greatly reduced in the similar manner to the embodiment of Figure 5. [0054] Moreover, an effective capacitance, twice the value of Ci, and variable parasitic capacitance Cp3 are effectively in parallel and the capacitance value of capacitor Ci may be significantly larger than the variable parasitic capacitance CP3. When the sampling switch M₄ is turned OFF, charge is sampled on the capacitor Cι_2- The non-linear input dependent charge injection is greatly reduced in the similar manner to the embodiment of Figure 5. Furthermore, the sampling circuit of Figure 8 reduces the effect of substrate noise.

[0055] Figure 9 illustrates a fourth example of a fully differential closed-loop sampling circuit being employed in order to improve power supply, substrate rejection, and removal of a differential component of the nonlinear input dependent charge injection. As illustrated in Figure 9, a sampling switch M2, when turned ON, connects the inverting input and the non-inverting output of a differential amplifier

30. With an input switch Mi turned ON, a first plate 4 of sampling capacitor Cu receives the input voltage V_1n+, while a second plate 5 of the sampling capacitor Cu settles to a common-mode voltage of the differential amplifier 30. [0056] Moreover, as illustrated in Figure 9, a sampling switch MU, when turned ON, connects the non-inverting input and the inverting output of a differential amplifier 30. With an input switch M₃ turned ON, a first plate 6 of sampling capacitor CL2 receives the input voltage Vm, while a second plate 7 of the sampling capacitor Cι_2 settles to a common-mode voltage of the differential amplifier 30. A capacitor Ci, which is substantially more linear than a variable parasitic capacitance Cp3 of the input switch M₃ or a variable parasitic capacitance Cpi of the input switch Mi , is connected between Node 1 and Node 10. [0057] In this embodiment, an effective capacitance, twice the value of Ci, and variable parasitic capacitance CPI are effectively in parallel and the capacitance value of capacitor Ci may be significantly larger than the variable parasitic capacitance C_P1. When the sampling switch M₂ is turned OFF,. charge is sampled on the capacitor Cu .

[0058] The parasitic capacitance C_Pτ is made substantially independent of the input voltage by bootstrapping the back gate of the input switch M-|. A buffer amplifier 20 is connected between the input voltage and the back gate of the input switch M-). The voltage across the parasitic capacitance Cpi is approximately zero regardless of the input voltage.

[0059] Alternatively, the buffer amplifier 20 can be made to have a constant offset voltage such that a prescribed DC voltage is maintained across the parasitic capacitance C_p1. [0060] Moreover, an effective capacitance, twice the value of Ci, and variable parasitic capacitance CP3 are effectively in parallel and the capacitance value of capacitor Ci may be significantly larger than the variable parasitic capacitance Cp₃. When the sampling switch M4 is turned OFF, charge is sampled on the capacitor CL2. A second buffer amplifier 25 is connected between the input voltage and the back gate of the input switch M3. The voltage across the parasitic capacitance Cp₃ is approximately zero regardless of the input voltage.

[0061] Alternatively, the buffer amplifier 25 can be made to have a constant offset voltage such that a prescribed DC voltage is maintained across the parasitic capacitance C_p3.

[0062] The closed-loop sampling of Figure 9 also removes the effect of the offset voltage associated with the differential amplifier 30 and removes the differential component of the nonlinear input dependent charge injection associated with the differential amplifier 30, thereby enabling the fully differential closed-loop sampling circuit of Figure 9 to be appropriate for analog-to-digital converters and other precision sampled-data circuits. Furthermore, the sampling circuit of Figure 9 reduces the effect of substrate noise.

[0063] Figure 10 illustrates a fifth example of a fully differential closed-loop sampling circuit being employed in order to improve power supply, substrate rejection, and removal of a differential component of the nonlinear input dependent charge injection. As illustrated in Figure 10, a sampling switch Wk, when turned ON, connects the inverting input and the non-inverting output of a differential amplifier 30. With an input switch Mi turned ON, a first plate 4 of sampling capacitor Cu receives the input voltage Vm, while a second plate 5 of the sampling capacitor Cu settles to a common-mode voltage of the differential amplifier 30.

[0064] Moreover, as illustrated in Figure 10, a sampling switch M₄, when turned ON, connects the non-inverting input and the inverting output of a differential amplifier 30. With an input switch M₃ turned ON, a first plate 6 of sampling capacitor Cι_₂ receives the input voltage Vm, while a second plate 7 of the sampling capacitor C_L2 settles to a common-mode voltage of the differential amplifier 30. A capacitor Ci, which is substantially more linear than a variable parasitic capacitance Cp3 of the input switch M3 or a variable parasitic capacitance CP_I of the input switch Mi, is connected between Node 1 and Node 10. [0065] In this embodiment, an effective capacitance, twice the value of C-i, and variable parasitic capacitance Cpi are effectively in parallel and the capacitance value of capacitor Ci may be significantly larger than the variable parasitic capacitance C_Pi. When the sampling switch M₂ is turned OFF, charge is sampled on the capacitor Cu- [0066] Both, the parasitic capacitance C_pi and the R_ONI resistance of the input switch Mi is kept constant. The parasitic capacitance C_pi is kept constant by keeping the voltage across C_p1 constant in the same manner as in Figure 9. The RON_I resistance of the input switch Mi is kept constant by bootstrapping the gate of input switch Mi when it is turned ON by tying the gate of to the output of a third buffer amplifier 21 The third buffer amplifier 21 biases the gate of the input switch Mi at a voltage that is offset by a constant amount from the input voltage V,_n. This keeps the gate-to-source voltage of the input switch Mi constant. Since the threshold voltage of the input switch Mi is kept constant by the back gate bootstrapping, the RONI resistance of the input switch Mi is constant regardless of the input voltage. [0067] Moreover, the parasitic capacitance C_p3 and the R_ON3 resistance of the input switch M₃ is kept constant. The parasitic capacitance C_P3 is kept constant by keeping the voltage across C_P3 constant in the same manner as in Figure 9. The RON₃ resistance of the input switch M₃ is kept constant by bootstrapping the gate of input switch M₃ when it is turned ON by tying the gate of to the output of a fourth buffer amplifier 26 The fourth buffer amplifier 26 biases the gate of the input switch

M₃ at a voltage that is offset by a constant amount from the input voltage V_iri. When the input switch M₃ is turned OFF, the gate is tied to a constant voltage, for example, ground potential. This keeps the gate-to-source voltage of the input switch M₃ constant. Since the threshold voltage of the input switch M₃ is kept constant by the back gate bootstrapping, the RON₃ resistance of the input switch M₃ is constant regardless of the input voltage.

[0068] The closed-loop sampling of Figure 10 also removes the effect of the offset voltage associated with the differential amplifier 30 and removes the differential component of the nonlinear input dependent charge injection associated with the differential amplifier 30, thereby enabling the fully differential closed-loop sampling circuit of Figure 10 to be appropriate for analog-to-digital converters and other precision sampled-data circuits. Furthermore, the sampling circuit of Figure 10 reduces the effect of substrate noise. [0069] Figure 11 illustrates another example of a fully differential sampling circuit being employed in order to improve power supply, substrate rejection, and removal of a differential component of the nonlinear input dependent charge injection. As illustrated in Figure 1 1 , when an input switch Mh is turned ON, a first plate 4 of sampling capacitor Cu receives the input voltage V_1n+. A sampling switch M2, when turned ON, connects a second plate 5 of the sampling capacitor Cu to a constant voltage, preferably a common-mode voltage.

[0070] Moreover, as illustrated in Figure 11 when an input switch M₃ is turned ON, a first plate 6 of sampling capacitor CL2 receives the input voltage Vi_n-. A sampling switch M2, when turned ON, connects a second plate 7 of the sampling capacitor Cι_2 to a constant voltage, preferably a common-mode voltage. A capacitor Ci, which is substantially more linear than a variable parasitic capacitance CP3 of the input switch M₃ or a variable parasitic capacitance CPI of the input switch Mi, is connected between Node 1 and Node 10. [0071] In this embodiment, an effective capacitance, twice the value of Ci, and variable parasitic capacitance Cpi are effectively in parallel and the capacitance value of capacitor Ci may be significantly larger than the variable parasitic capacitance C_P1. When the sampling switch M₂ is turned OFF, charge is sampled on the capacitor Cu .

[0072] The parasitic capacitance CPI is made substantially independent of the input voltage by bootstrapping the back gate of the input switch Mi . A buffer amplifier 20 is connected between the input voltage and the back gate of the input switch Mi. The voltage across the parasitic capacitance Cpi is approximately zero regardless of the input voltage.

[0073] Alternatively, the buffer amplifier 20 can be made to have a constant offset voltage such that a prescribed DC voltage is maintained across the parasitic capacitance C_pi .

[0074] Moreover, an effective capacitance, twice the value of Ci, and variable parasitic capacitance Cp3 are effectively in parallel and the capacitance value of capacitor Ci may be significantly larger than the variable parasitic capacitance Cp₃. When the sampling switch M₄ is turned OFF, charge is sampled on the capacitor CL2- A second buffer amplifier 25 is connected between the input voltage and the back gate of the input switch M₃. The voltage across the parasitic capacitance C_P3 is approximately zero regardless of the input voltage.

[0075] Alternatively, the buffer amplifier 25 can be made to have a constant offset voltage such that a prescribed DC voltage is maintained across the parasitic capacitance C_P3. [0076] The sampling circuit of Figure 11 removes the nonlinear input dependent charge injection, thereby enabling the fully differential closed-loop sampling circuit of Figure 11 to be appropriate for analog-to-digital converters and other precision sampled-data circuits. Furthermore, the sampling circuit of Figure 11 reduces the effect of substrate noise.

[0077] Figure 12 illustrates another example of a fully differential sampling circuit being employed in order to improve power supply, substrate rejection, and removal of a differential component of the nonlinear input dependent charge injection. As illustrated in Figure 12, when an input switch Mi is turned ON, a first plate 4 of sampling capacitor Cu receives the input voltage Vi_n+. A sampling switch WI2, when turned ON, connects a second plate 5 of the sampling capacitor Cu to a constant voltage, preferably a common-mode voltage. [0078] Moreover, as illustrated in Figure 12 when an input switch M3 is turned ON, a first plate 6 of sampling capacitor Cι_₂ receives the input voltage Vj_n-. A sampling switch M₂, when turned ON, connects a second plate 7 of the sampling capacitor C|_2 to a constant voltage, preferably a common-mode voltage. A capacitor C₁, which is substantially more linear than a variable parasitic capacitance C_P3 of the input switch M₃ or a variable parasitic capacitance CPI of the input switch Mi, is connected between Node 1 and Node 10. [0079] In this embodiment, an effective capacitance, twice the value of C-i, and variable parasitic capacitance C_PI are effectively in parallel and the capacitance value of capacitor Ci may be significantly larger than the variable parasitic capacitance C_PI. When the sampling switch Λ/I₂ is turned OFF, charge is sampled on the capacitor Cu. [0080] Both, the parasitic capacitance C_pi and the R_ONI resistance of the input switch Mi is kept constant. The parasitic capacitance C_p1 is kept constant by keeping the voltage across Cpi constant in the same manner as in Figure 9. The Ro_Ni resistance of the input switch Mi is kept constant by bootstrapping the gate of input switch Mh when it is turned ON by tying the gate of to the output of a third buffer amplifier 21 The third buffer amplifier 21 biases the gate-of the input switch

M-i at a voltage that is offset by a constant amount from the input voltage Vi_n. This keeps the gate-to-source voltage of the input switch Mh constant. Since the threshold voltage of the input switch Mi is kept constant by the back gate bootstrapping, the RONI resistance of the input switch Mh is constant regardless of the input voltage. [0081] Moreover, the parasitic capacitance C_P3 and the R_ON₃ resistance of the input switch M₃ is kept constant. The parasitic capacitance C_p3 is kept constant by keeping the voltage across C_P3 constant in the same manner as in Figure 9. The R_ON₃ resistance of the input switch M₃ is kept constant by bootstrapping the gate of input switch M₃ when it is turned ON by tying the gate of to the output of a fourth buffer amplifier 26 The fourth buffer amplifier 26 biases the gate of the input switch M₃ at a voltage that is offset by a constant amount from the input voltage Vi_n. When the input switch M₃ is turned OFF, the gate is tied to a constant voltage, for example, ground potential. This keeps the gate-to-source voltage of the input switch M₃ constant. Since the threshold voltage of the input switch M₃ is kept constant by the back gate bootstrapping, the RON3 resistance of the input switch M₃ is constant regardless of the input voltage.

[0082] The sampling circuit of Figure 12 also removes the nonlinear input dependent charge injection, thereby enabling the fully differential closed-loop sampling circuit of Figure 12 to be appropriate for analog-to-digital converters and other precision sampled-data circuits. Furthermore, the sampling circuit of Figure 12 reduces the effect of substrate noise.

[0083] In summary, by including a capacitor, which is substantially more linear than a variable parasitic capacitance of a source-side transistor, is connected in parallel to the variable parasitic capacitance of a source-side transistor to reduce the dependence of charge split ratio on the input voltage. Furthermore, the capacitor reduces the effect of substrate noise.

[0084] While various examples and embodiments of the present invention have been shown and described, it will be appreciated by those skilled in the art that the spirit and scope of the present invention are not limited to the specific description and drawings herein, but extend to various modifications and changes.

Claims

What is claimed is:

1. A sampling circuit, comprising: an input voltage source; a first switch having an input operatively connected to said input voltage source; a sampling capacitor operatively connected to an output of said first switch; a second switch having an input operatively connected to said sampling capacitor; and a second capacitor operatively connected to said output of said first switch.

2. The sampling circuit as claimed in claim 1, wherein said first switch has a variable parasitic capacitance and said second capacitor has a capacitance greater than the variable parasitic capacitance of said first switch.

3. The sampling circuit as claimed in claim 1, wherein said first switch has a variable parasitic capacitance and said second capacitor is in parallel with the variable parasitic capacitance of said first switch.

4. The sampling circuit as claimed in claim 3, wherein said second capacitor has a capacitance greater than the variable parasitic capacitance of said first switch.

5. The sampling circuit as claimed in claim 1 , wherein said first switch has a variable parasitic capacitance and a combined variable parasitic capacitance and capacitance of said second capacitor is more linear than the variable parasitic capacitance of said first switch.

6. The sampling circuit as claimed in claim 1 , wherein said second capacitor is connected to a constant voltage source.

7. The sampling circuit as claimed in claim 1, further comprising: a second input voltage source; a third switch having an input operativeiy connected to said second input voltage source; a second sampling capacitor operatively connected to an output of said third switch; and a fourth switch having an input operatively connected to said second sampling capacitor; said second capacitor being operatively connected across said output of said third switch and said output of said first switch.

1 8. The sampling circuit as claimed in claim 7, wherein said third switch has a variable parasitic capacitance and said second capacitor has a capacitance greater than the variable parasitic capacitance of said third switch.

1 9. The sampling circuit as claimed in claim 7, wherein said second capacitor has a capacitance greater than the variable parasitic capacitance of said third switch.

1 10. The sampling circuit as claimed in claim 7, wherein said third switch has a variable parasitic capacitance and a combined variable parasitic capacitance and capacitance of said second capacitor is more linear than the variable parasitic capacitance of said third switch.

1 11. The sampling circuit as claimed in claim 1, further comprising a first buffer amplifier operatively connected between said first input voltage source and a back-gate of said first switch.

1 12. The sampling circuit as claimed in claim 1 , further comprising a first amplifier operatively connected to a gate of said first switch to bias the gate of said first switch at

3 a voltage that is offset by a constant amount from an input voltage of said first input voltage ^■ source, i

2 13. The sampling circuit as claimed in claim 7, further comprising:

3 a first buffer amplifier operatively connected between said -first input voltage source and a back-gate of said first switch; and

5 a second buffer amplifier operatively connected between said second input

6 voltage source and a back-gate of said third switch.

i

14. The sampling circuit as claimed in claim 13, further comprising: a first amplifier operatively connected to a gate of said first switch to bias the gate of said first switch at a voltage that is offset by a constant amount from an input voltage of said first input voltage source; and a second amplifier operatively connected to a gate of said third switch to bias the gate of said third switch at a voltage that is offset by a constant amount from an input voltage of said second input voltage source.

15. A sampling circuit, comprising: an input voltage source; a first switch having an input operatively connected to said input voltage source; a sampling capacitor operatively connected to an output of said first switch; an operational amplifier having an inverting input operatively. connected to said sampling capacitor; a second switch operatively connected across the inverting input of said operational amplifier and an output of said operational amplifier; and a second capacitor operatively connected to said output of said first switch.

16. The sampling circuit as claimed in claim 15, wherein said first switch has a variable parasitic capacitance and said second capacitor has a capacitance greater than the variable parasitic capacitance of said first switch.

17. The sampling circuit as claimed in claim 15, wherein said first switch has a variable parasitic capacitance and said second capacitor is in parallel with the variable parasitic capacitance of said first switch.

18. The sampling circuit as claimed in claim 17, wherein said second capacitor has a capacitance greater than the variable parasitic capacitance of said first switch.

19. The sampling circuit as claimed in claim 15, wherein said first switch has a variable parasitic capacitance and a combined variable parasitic capacitance and capacitance of said second capacitor is more linear than the variable parasitic capacitance of said first switch.

20. The sampling circuit as claimed in claim 15, wherein said second capacitor is connected to a constant voltage source.

21. A sampling circuit, comprising: a first input voltage source; a first switch having an input operatively connected to said first input voltage source; a first sampling capacitor operatively connected to an output of said first switch; a differential amplifier having an inverting input operatively connected to said first sampling capacitor; a second switch operatively connected across the inverting input of said differential amplifier and a non-inverting output of said differential amplifier; and a second capacitor operatively connected to said output of said first switch.

22. The sampling circuit as claimed in claim 21 , wherein said first switch has a variable parasitic capacitance and said second capacitor has a capacitance greater than the variable parasitic capacitance of said first switch.

23. The sampling circuit as claimed in claim 21 , wherein said first switch has a variable parasitic capacitance and said second capacitor is in parallel with the variable parasitic capacitance of said first switch.

24. The sampling circuit as claimed in claim 23, wherein said second capacitor has a capacitance greater than the variable parasitic capacitance of said first switch.

25. The sampling circuit as claimed in claim 21 , wherein said first switch has a variable parasitic capacitance and a combined variable parasitic capacitance and capacitance of said second capacitor is more linear than the variable parasitic capacitance of said first switch.

26. The sampling circuit as claimed in claim 21 , wherein said second capacitor is connected to a constant voltage source.

27. The sampling circuit as claimed in claim 21 , further comprising: a second input voltage source; a third switch having an input operatively connected to said second input voltage source; a second sampling capacitor operatively connected to an output of said third switch; a fourth switch operatively connected across a non-inverting input of said differential amplifier and an inverting output of said differential amplifier; and a fourth capacitor operatively connected to said output of said third switch.

28. The sampling circuit as claimed in claim 27, wherein said third switch has a variable parasitic capacitance and said fourth capacitor has a capacitance greater than the variable parasitic capacitance of said third switch.

29. The sampling circuit as claimed in claim 27, wherein said third switch has a variable parasitic capacitance and said fourth capacitor is in parallel with the variable parasitic capacitance of said third. switch.

30. The sampling circuit as claimed in claim 29, wherein said fourth capacitor has a capacitance greater than the variable parasitic capacitance of said third switch.

31. The sampling circuit as claimed in claim 17, wherein said third switch has a variable parasitic capacitance and a combined variable parasitic capacitance and capacitance of said fourth capacitor is more linear than the . variable parasitic capacitance of said third switch.

32. The sampling circuit as claimed in claim 27, wherein said fourth capacitor is connected to a constant voltage source.

33. The sampling circuit as claimed in claim 21 , further comprising: a second input voltage source; a third switch having an input operatively connected to said second input voltage source; a second sampling capacitor operatively connected to an output of said third switch; and a fourth switch operatively connected across a non-inverting input of said differential amplifier and an inverting output of said differential amplifier; said second capacitor being operatively connected across said output of said third switch and said output of said first switch.

34. The sampling circuit as claimed in claim 33, wherein said third switch has a variable parasitic capacitance and said second capacitor has a capacitance greater than the variable parasitic capacitance of said third switch.

35. The sampling circuit as claimed in claim 34, wherein said second capacitor has a capacitance greater than the variable parasitic capacitance of said third switch.

36. The sampling circuit as claimed in claim 33, wherein said third switch has a variable parasitic capacitance and a combined variable parasitic capacitance and capacitance of said second capacitor is more linear than the variable parasitic capacitance of said third switch.

37. The sampling circuit as claimed in claim 21, further comprising a first buffer amplifier operatively connected between said first input voltage source and a back-gate of said first switch.

38. The sampling circuit as claimed in claim 21 , further comprising a first amplifier operatively connected to a gate of said first switch to bias the gate of said first switch at a voltage that is offset by a constant amount from an input voltage of said first input voltage source.

39. The sampling circuit as claimed in claim 27, further comprising: a first buffer amplifier operatively connected between said first input voltage source and a back-gate of said first switch; and a second buffer amplifier operatively connected between said second input voltage source and a back-gate of said third switch.

40. The sampling circuit as claimed in claim 39, further comprising: a first amplifier operatively connected to a gate of said first switch to bias the gate of said first switch at a voltage that is offset by a constant amount from an input voltage of said first input voltage source; and a second amplifier operatively connected to a gate of said third switch to bias the gate of said third switch at a voltage that is offset by a constant amount from an input voltage of said second input voltage source.

41. The sampling circuit as claimed in claim 33, further comprising: a first buffer amplifier operatively connected between said first input voltage source and a back-gate of said first switch; and a second buffer amplifier operatively connected between said second input voltage source and a back-gate of said third switch.

42. The sampling circuit as claimed in claim 41 , further comprising: a first amplifier operatively connected to said gate of said first switch to bias the gate of said first switch at a voltage that is offset by a constant amount from an input voltage of said first input voltage source; and a second amplifier operatively connected to said gate of said third switch to bias the gate of said third switch at a voltage that is offset by a constant amount from an input voltage of said second input voltage source.