WO2008047445A1

WO2008047445A1 - Transmission device

Info

Publication number: WO2008047445A1
Application number: PCT/JP2006/320917
Authority: WO
Inventors: Shoichi Fujita; Takashi Izumi; Yuta Seki
Original assignee: Panasonic Corporation
Priority date: 2006-10-20
Filing date: 2006-10-20
Publication date: 2008-04-24

Abstract

It is an object to provide a transmission device, which uses a LINC system, is easily miniaturized and manufactured with low price and low power consumption, holds the linearity (leakage power outside a frequency band width and modulation accuracy) of a signal wave with an envelope fluctuation in the case that a signal frequency band width is limited, and amplifies a transmission signal with high power efficiency. This device (100) resolves a transmission IQ complex base band signal into a plurality of fixed envelope signals, and synthesizes them after nonlinear amplification of each of them. A digital orthogonal modulation unit (110) digitally up-converts a BB signal and generates a Low-IF digital signal, which is a scalar signal. A fixed envelope conversion unit (120) multiplexes a signal and an orthogonal component in the Low-IF digital signal, i.e., the converted scalar signal, to generate first and second envelope signals (S1, S2). Non-linear amplifiers (161) and (162) amplify the first and second envelope signals (S1, S2), respectively. A lossless synthesizer (170) synthesizes signals obtained by the non-linear amplifiers (161) and (162).

Description

Transmitter

Technical field

The present invention relates to a transmission apparatus having an output amplifier circuit (PA: Power Amplifier) for digital wireless communication.

Background art

Conventionally, the transmission signal S (t) is replaced with two constant envelope signals S (t), S (for the problem of achieving both efficiency and linearity in an output amplifier circuit (PA) for digital wireless communication. t)

0 01 02

In addition, a LINC method (Linear Amplification with Nonlinear Component) is known in which each is amplified by a highly efficient nonlinear amplifier and then synthesized (for example, Patent Document 1).

FIG. 1 is a diagram for explaining a conventional LINC amplification method.

[0004] In the LINC method, when the transmission signal (modulation signal) is expressed by the following equation (1), it is decomposed into two constant envelope signals S (t) and S (t) shown in FIG.

01 02

[0005] S (t) = V (t) Xcos {o> ct + φ (t)} '"(l)

o

S (t) and S (t) are expressed by the following equations (2) and (3), where the maximum amplitude of S (t) is Vmax.

01 02 0

[0006] S (t) = Vmax / 2 X cos {oct + φ (t)} (2)

01

S (t) = Vmax / 2 X cos {coct + 0 (t)} (3)

02

Φ (t) = φ (t) + a (t)

Θ (t) = (t) a (t)

When these S (t) and S (t) are amplified and vector synthesized, as shown in Fig. 1.

01 02

A signal obtained by amplifying s (t) is obtained.

0

[0007] Here, since S (t) and S (t) are constant envelope signals, a highly efficient nonlinear amplifier circuit is used.

01 02

Can be. Also, by using a lossless synthesizer, it is possible to eliminate the power consumption of components that are canceled out during synthesis. As described above, the LINC output amplifier circuit (PA) enables highly efficient and linear amplification.

Patent Document 1: Japanese Patent No. 1892840 Disclosure of the invention

Problems to be solved by the invention

However, in the conventional LINC method, constant envelope signals S (t) and S (t) are wideband signals.

01 02

Therefore, the following problems arise.

[0009] 1) High speed operation of constant envelope signals S and S generation circuit (high sampling rate operation, computational complexity)

01 02

Large) is required, making it difficult to reduce size, cost, and power consumption.

[0010] 2) The synthesizer must also have a wide bandwidth, and the broadband loss of the lossless synthesizer used in the conventional LINC method is difficult, and the efficiency of the synthesizer decreases and the synthesizer itself It is difficult to reduce the size and cost.

[0011] 3) When the signal band is limited by the constant envelope signal S 1, S generator and synthesizer

01 02

The effect of this is increased and the linearity (out-of-band leakage, modulation accuracy) is greatly degraded.

Here, when the conventional method is used, the constant envelope signals s (t) and s (t) become wideband signals.

01 02

The point will be described. In the conventional method, the constant envelope signal S at the zero crossing of the transmission signal S (t)

0

, S phase becomes discontinuous.

01 02

This will be described with reference to FIG. Transmit signal S (t) crosses zero and changes to S (t + At)

0 0

Therefore, S (t) is phase-inverted to S (t + A t), and S (t) is phase-inverted to S (t + A t).

01 01 02 02

As a result, the phase changes discontinuously. In other words, in Fig. 2, S (t) crosses 0 and starts from the first quadrant.

0

If we change to the 3rd quadrant, S (t) crosses 0 from the 2nd quadrant, and S (t +

01 01

At (t), the phase changes instantaneously discontinuously, and S (t) extends from the 4th quadrant to the 2nd quadrant across 0.

02

The phase changes instantaneously discontinuously in S (t + A t).

02

[0014] Accordingly, S and S become very wideband signals. Figure 3 shows S and S frequency scans.

01 02 01 02

An example of a vector is shown. Figure 3 shows the frequency spectrum of the constant envelope signals S and S when the primary modulation is 16QAM with 864 subcarrier OFDM, and the transmission signal bandwidth

01 02

Compared to, the signal is very wide.

[0015] Next, the influence when the band of the S and S signals is limited will be described. Twice the transmission signal bandwidth

01 02

Fig. 4 and Fig. 5 show an example of the output cascading and the reception constellation (demodulation by an ideal receiver) when the bandwidth is limited to the specified bandwidth.

Note that FIG. 4 shows the case where 864 subcarrier OFDM and primary modulation is 16QAM. Fig. 5 shows the frequency spectrum of the final output after synthesis of the constant envelope signals S, S and S and S after nonlinear amplification in this case.

01 02 01 02

Figure 4 shows the reception constellation corresponding to Fig. 4.

In FIG. 4, a large leakage of about 20 dB occurs outside the band, and the constellation shown in FIG. 5 is also deteriorated. In Fig. 4, the bands of S and S are widened.

01 02

This is because high-frequency components are restored by nonlinear amplification.

[0018] An object of the present invention is to easily reduce the size, reduce the cost, and reduce the power consumption by using the LINC method. Even when the signal band is limited, the linearity of a signal wave having an envelope variation is achieved. It is an object of the present invention to provide a transmission apparatus and a transmission method capable of amplifying a transmission signal while maintaining (out-of-band leakage power, modulation accuracy) and having high power efficiency.

Means for solving the problem

[0019] A transmitting apparatus of the present invention is an amplifying apparatus using a LINC method that decomposes an IQ complex baseband signal to be transmitted into a plurality of constant envelope signals, and synthesizes each after nonlinear amplification, A scalar signal converting means for converting the IQ complex baseband signal into a scalar signal, and a constant envelope for generating the first and second constant envelope signals by multiplexing the signals on orthogonal components of the converted scalar signal. The signal obtained by the line signal generating means, the first and second nonlinear amplifiers for amplifying the first and second constant envelope signals, respectively, and the signals obtained by the first and second nonlinear amplifiers are synthesized. A configuration having a synthesizer is adopted.

The invention's effect

[0020] According to the present invention, it is easy to reduce the size, reduce the cost, and reduce the power consumption. Even when the signal band is limited, the linearity of the signal wave having the envelope variation (the out-of-band leakage power) , Modulation accuracy) and transmit power with high power efficiency.

Brief Description of Drawings

[0021] [Fig. 1] Diagram for explaining the conventional LINC method

[Figure 2] Diagram for explaining the conventional LINC system

[Figure 3] Diagram for explaining the conventional LINC system

[Figure 4] Diagram for explaining the conventional LINC system

[Figure 5] Diagram for explaining the conventional LINC system FIG. 6 is a diagram for explaining the principle of the present invention.

FIG. 7 is a diagram for explaining the principle of the present invention.

FIG. 8 is a diagram for explaining the principle of the present invention.

FIG. 9 is a diagram for explaining the principle of the present invention.

FIG. 10 is a diagram for explaining the principle of the present invention.

FIG. 11 is a block diagram showing the configuration of the transmitting apparatus according to Embodiment 1 of the present invention.

FIG. 12 is a circuit diagram showing an example of a specific configuration of a digital quadrature modulation unit

FIG. 13 is a block diagram showing the configuration of the transmitting apparatus according to Embodiment 2 of the present invention.

FIG. 14 shows an example of subcarriers to which symbols are assigned in the subcarrier arrangement section.

FIG. 15 is a block diagram showing the configuration of the transmitting apparatus according to Embodiment 3 of the present invention.

[Fig.16] Diagram for explaining OFDM signal in which NULL is placed by NULL insertion part

[Fig.17] Diagram for explaining OFDM signal after IFFT by IFFT section

FIG. 18 is a diagram for explaining an OFDM signal in which NULL is inserted in the fourth embodiment.

FIG. 19 is a diagram for explaining the OFDM signal converted by the IFFT unit in the fourth embodiment.

FIG. 20 is a diagram showing a final output OFDM signal when using a sum component without constraint of 2 × Repetition, which is finally output in the quadrature modulation unit in the fourth embodiment.

FIG. 21 is a block diagram showing the configuration of the transmitting apparatus according to Embodiment 5 of the present invention.

FIG. 22 is a diagram for explaining the spectrum of a signal that has been LINC amplified in the two amplification systems in the transmitter of the fifth embodiment.

FIG. 23 is a diagram for explaining the operation of a constant envelope conversion unit as a modification of the transmitting apparatus according to Embodiment 5.

FIG. 24 is a block diagram showing the configuration of the transmitting apparatus according to the sixth embodiment of the present invention.

FIG. 25 is a diagram for explaining an IFFT unit in the transmitting apparatus according to the seventh embodiment.

BEST MODE FOR CARRYING OUT THE INVENTION

Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings.

[0023] The present invention relates to an output amplifier circuit (PA) for digital wireless communication, In a LINC amplifier circuit that aims to achieve both formability and high efficiency, the linearity (out-of-band leakage power, modulation accuracy) is improved by applying a new decomposition method to the constant envelope signal. The

[0024] (1) Basic principle of the present invention

The basic principle of the present invention is to convert an IQ complex baseband signal into a scalar signal, multiplex the signals with the orthogonal components to generate constant envelope signals (S1, S2), and amplify and synthesize these signals. To generate a transmission signal.

[1] 1) Constant envelope signal (S1, S2) generation method

FIG. 6 is a diagram for explaining the principle of the present invention and shows a method for generating a constant envelope signal according to the present invention.

In FIG. 6, a scalarized modulation signal (hereinafter referred to as “scalar signal”) is S (t), and the scalar signal S (t) is a signal that changes on the I axis.

[0027] In Fig. 6, for a scalar signal S (t), considering a circle whose radius is the maximum amplitude of the scalar signal S (t) or a larger amplitude, S (t), S ( t)

1 2

The alternating components A (t), — A (t) are multiplexed to generate a constant envelope signal. These constant envelope signals S1 and S2 are expressed by the following equations (4) and (5).

[0028] That is, if the constant envelope amplitude is C (C≥S (t) maximum amplitude),

A (t) = sqrt (C ² -S ² (t))

S (t) = S (t) + j * A (t) "-(4)

S (t) = S (t) —j * A (t) "-(5)

2

It becomes.

[0029] (1-2) Scalar signal S (t) crosses zero

FIG. 7 is a diagram for explaining the principle of the present invention, and is an explanatory diagram when the scalar signal S (t) crosses zero.

[0030] As shown in Fig. 7, when the transmission signal (scalar signal) S (t) crosses zero and changes to S (t + At), S (t) changes to S (t + At), S (t t) changes to S (t + At).

1 1 2 2

That is, according to the present invention using equations (4) and (5), when the transmission signal S (t) converted to a scalar changes to S (t + At) across 0, S (t t) from the first quadrant to the second quadrant S (t + At) As a result, S (t) changes from the 4th quadrant to the 3rd quadrant S (t + At).

twenty two

[0032] This change is a continuous change that moves on the circumference, and there is no phase change of 180 degrees, and no phase discontinuity occurs. As a result, the spectral broadening of the constant envelope signals SI and S2 is reduced, and the influence of the band limitation is greatly reduced.

FIGS. 8 to 10 are diagrams for explaining the principle of the present invention, FIG. 8 is a diagram showing the frequency spectrum of the constant envelope signals SI and S2 before nonlinear amplification in the present invention, and FIG. FIG. 5 is a diagram showing an S1, S2 spectrum after nonlinear amplification and a final output (after synthesis) spectrum when band-limiting in the present invention. FIG. 10 is a diagram showing an example of a reception constellation when the band is limited in the present invention. FIG. 8 shows a spectrum without band limitation, and FIGS. 9 and 10 show an example in which the spectrum is limited to twice the signal band.

[0034] As shown in FIGS. 8 to 9, in the amplification method of the present invention, the leakage outside the band is reduced to −80 dB or less even when the bandwidth is limited. As shown in FIG. There is almost no degradation of Chillon. As described above, the principle of the present invention can be greatly improved as compared with the conventional method.

A transmission apparatus according to an embodiment that realizes such a principle will be described below.

(2) Embodiment 1

FIG. 11 is a block diagram showing a configuration of transmitting apparatus 100 according to Embodiment 1 of the present invention.

[0037] The transmission apparatus 100 shown in FIG. 11 includes a digital quadrature modulation unit (scalar signal conversion unit) 110, a constant envelope conversion unit 120, 07 eight conversion units (“0 eight”) 131, 132, a low-pass filter (“LPF”). 14 and 142, quadrature modulation sections 151 and 152, nonlinear amplifiers 161 and 162, and a synthesizer (lossless synthesizer) 170. The synthesizer 170 is shown as a (lossless) synthesizer in FIG.

A digital quadrature modulation (Low-IF) unit 110 performs scalar conversion processing on an input digital baseband (hereinafter “BB” t) signal.

Digital quadrature modulation section 110 digitally upconverts the BB signal to generate a Low-IF digital signal, and outputs the low-IF digital signal to constant envelope conversion section 120. The digital BB signal input to the digital quadrature modulation unit 110 may be a power OFDM baseband signal that is a complex baseband signal modulated by QPSK or 16QAM! / !. FIG. 12 is a circuit diagram showing an example of a specific configuration of the digital quadrature modulation unit 110.

As shown in FIG. 12, the digital quadrature modulation unit 110 multiplies the input complex BB signal I signal by cos co t, multiplies the input complex BB signal Q signal by sin co t, The low-IF (low intermediate frequency) signal (Real Part real number) is generated by calorie calculation.

[0042] The Low-IF digital signal generated by the digital quadrature modulation unit 110 is a scalar signal (real number) Real Part. This scalar signal (real number) Real Part corresponds to the scalar signal S (t) described in the basic principle in the present embodiment.

[0043] The constant envelope converter 120 is a scalar signal (Low

The signal (scalar signal) is multiplexed on the quadrature component of the (IF signal) to generate two constant envelope signals SI and S2, and output them to the DZA converters 131 and 132. These constant envelope signals Sl and S2 correspond to the constant envelope signals S (t) and S (t) described in the basic principle in this embodiment.

1 2

[0044] The DZA conversion units 131 and 132 convert the constant envelope signals S1 and S2 output from the constant envelope conversion unit 120 into analog signals from the digital signal, respectively, and output the analog signals to the LPFs 141 and 142.

LPFs 141 and 142 remove sampling frequency components and aliasing components, which are unnecessary components included in the signals input from DZA conversion units 131 and 132, and output them to quadrature modulation units 151 and 152. .

[0046] The quadrature modulation sections 151 and 152 orthogonally modulate the I and Q components of the analog constant envelope signals SI and S2 with the Lo signal to form an RF band signal and output the RF band signal to the nonlinear amplifiers 161 and 162.

[0047] The frequency of the Lo signal that orthogonally modulates the constant envelope signals SI and S2 in the orthogonal modulation units 151 and 152 is a frequency that is offset by the Low-IF frequency with respect to the transmission frequency (center). Of the sum or difference frequency components, components different from the transmission frequency are removed.

Nonlinear amplifiers 161 and 162 nonlinearly amplify (limiter amplification) the constant envelope signal up-converted to the RF band, and output the nonlinearly amplified signal to synthesizer (lossless synthesizer) 170. As the nonlinear amplifiers 161 and 162, a highly efficient amplifier is used because linearity is not necessary.

[0049] The (lossless) synthesizer 170 is configured to add two non-linearly amplified constant envelope signals SI and S2 Synthesize. By combining, components multiplexed due to the constant envelope 匕 cancel each other, and only the transmission signal is output. The power of the components that cancel out is not consumed when a lossless synthesizer is used, and high efficiency can be achieved.

[0050] Thus, according to the present embodiment, digital quadrature modulation section 110 as a scalar signal conversion means for converting an IQ complex baseband signal into a scalar signal, and first and second orthogonal signals of the scalar signal. A constant envelope converter (constant envelope signal generator) 120 that generates the first and second constant envelope signals by multiplexing the scalar signal with the component, and the first and second constant envelope signals First and second nonlinear amplifiers 161 and 162 for amplifying, respectively, and a synthesizer (lossless synthesizer) 170 for synthesizing signals obtained by the first and second nonlinear amplifiers 161 and 162 are provided. Therefore, miniaturization, cost reduction, and low power consumption are easy, and even when the signal band is limited, the linearity (out-of-band leakage power, modulation accuracy) of the signal wave with envelope fluctuation is maintained. Realize transmitter 100 that can amplify transmission signals with high power efficiency it can.

[0051] (3) Embodiment 2

FIG. 13 is a block diagram showing a configuration of transmitting apparatus 200 according to Embodiment 2 of the present invention.

[0052] This transmitting apparatus 200 uses the above-described basic principle for OFDM (Orthogonal Frequency Division Multiplexing), and here, double-repetition in which one primary modulation symbol is redundantly transmitted by two subcarriers. Send.

The transmission apparatus 200 shown in FIG. 13 includes a coordinate conversion unit 211, a subcarrier arrangement unit 213, an IFFT (Inverse. Fast Fourier Transform) unit 216, a constant envelope conversion unit 220, 07 eight transform units (“0 eight”). ) 131, 132, Rhonos filters (“1 ^”) 141, 142, quadrature modulation units 151, 152, nonlinear amplifiers 161, 162, and synthesizer (lossless synthesizer) 170.

Note that this transmitting apparatus 200 has the same basic configuration as that of transmitting apparatus 100 corresponding to Embodiment 1 shown in FIG. 11. In the configuration of transmitting apparatus 100, digital orthogonal modulation section 110 is used instead. Thus, a coordinate conversion unit 211, a subcarrier arrangement unit 213, and an IFFT unit 216 are provided. Therefore, in the transmission apparatus 200, the same components as those of the transmission apparatus 100 are denoted by the same names and reference numerals, and description thereof is omitted. In transmission apparatus 200, coordinate conversion section 211, subcarrier arrangement section 213, and IFFT section 216 constitute scalar signal conversion section 210.

[0056] The coordinate conversion unit 211 converts the IQ plane coordinates of the subcarrier symbols to the OFDM baseband signal that is first-order modulated by QPSK, 16QAM, etc.

Output to 213.

[0057] The coordinate conversion unit 211 uses any one of the following four types of conversion methods. The IQ coordinate of modulation symbol n is (In, Qn). (1) Complex conjugate coordinate transformation: For example, (In, Q n) → (ln, —Qn) transformation to I axis symmetry (2) Complex conjugate inversion coordinate transformation: (In, Qn) → (- (In, Qn) is converted to Q axis symmetry (3) IQ exchange coordinate conversion: For example, (In, Qn) → (Qn, In) is converted symmetrically to the straight line Q = I (4) IQ permutation inversion coordinate transformation: For example, (In, Qn) → (-Qn, In). Symmetrical transformation with respect to the straight line Q = —1 Assign to. Specifically, the subcarrier arrangement unit 213 uses the coordinate-transformed OFDM baseband signal input from the coordinate transformation unit 211 to convert the coordinate-transformed symbol and its original symbol to DC on the frequency axis. Are assigned to subcarriers at symmetrical positions.

FIG. 14 is a diagram showing an example of subcarriers to which symbols are assigned by subcarrier arrangement section 213. FIG. 14 shows a case where the original symbol is converted with a complex conjugate relationship. The post-arrangement OFDM signal shown in Fig. 14 corresponds to double repetition in the subcarrier direction.

[0059] IFFT section 216 converts the frequency domain OFDM signal input by subcarrier arrangement section 213 into a time domain signal and outputs the time domain signal to constant envelope conversion section 220.

In IFFT section 216, when a signal in which symbols subjected to coordinate transformations (1) to (4) are arranged on symmetrical subcarriers is IFF, the output is as follows due to the nature of Fourier transformation.

[0061] In the coordinate transformation (1), only the real part signal (imaginary part = 0)

In coordinate transformation (2), only the imaginary part signal (real part = 0)

Real part = imaginary part in coordinate transformation (3)

In coordinate transformation (4), real part = one imaginary part Therefore, use (1) the real part, (2) the imaginary part, and (3) (4) the real part force imaginary part.

[0062] Thereby, the OFDM baseband signal can be converted into a scalar signal.

[0063] The constant envelope conversion unit 220 multiplexes the signal into the orthogonal components of the scalar signal such as only the real part, only the imaginary part, real part = imaginary part, real part = one imaginary part, etc. input from the IFFT part 216. Two constant envelope signals S 1 and S 2 are generated and output to the DZA converters 131 and 132.

[0064] The operations of constant envelope conversion section 220, DZA conversion sections 131 and 132, LPF 141 and 142, quadrature modulation sections 151 and 152, nonlinear amplifiers 161 and 162, and synthesizer 170 are the same as those in Embodiment 1 described above. Since it is the same as that of a thing, description is abbreviate | omitted. However, the Lo signal frequency should be the same as the transmission signal frequency (center). Also, the coordinate-converted symbol is transmitted as it is. This is equivalent to double Repetition transmission.

[0065] Thus, according to the present embodiment, the OFDM primary modulation symbol is converted to an I-axis symmetric symbol, converted to a Q-axis symmetric symbol, converted to a symbol in which IQ is replaced, or IQ is changed. A sub-array in which the coordinate conversion unit 211 that converts to a reversed and inverted symbol, the symbol converted by the coordinate conversion unit 211, and the symbol before coordinate conversion are arranged on subcarriers at symmetrical positions with the DC component in between By providing carrier arrangement section 213 and IFFT 216 for converting the symbols arranged by subcarrier arrangement section 213 into time domain signals, the same effects as in Embodiment 1 can be obtained when transmitting OFDM signals. This comes out.

[0066] (4) Embodiment 3

FIG. 15 is a block diagram showing a configuration of transmitting apparatus 300 according to Embodiment 3 of the present invention.

[0067] This transmitting apparatus 300 is used in OFDM (Orthogonal Frequency Division Multiplexing), and transmits twice the repetition.

Transmitting apparatus 300 shown in FIG. 15 is a modification of transmitting apparatus 200 according to Embodiment 2, and includes subcarrier arrangement section 313, NULL insertion section 314, IFFT (Inverse. Fast Four Transform) Unit 316, constant envelope conversion unit 320, 07 eight conversion units (“0 eight”) 131, 132, mouth-pass filters (“^”) 141, 142, quadrature modulation units 151, 152, nonlinear amplifier 161, 1 62, and a synthesizer (lossless synthesizer) 170.

[0069] It should be noted that this transmitting apparatus 300 has the same basic configuration as transmitting apparatus 200 corresponding to Embodiment 2 shown in FIG. 13, and in the configuration of transmitting apparatus 200, coordinate conversion section 211 is omitted. In addition, a NULL insertion unit 314 is provided between the subcarrier arrangement unit 313 and the IFFT unit 316. Therefore, in the transmission apparatus 300, the same components as those of the transmission apparatus 200 are denoted by the same names and reference numerals, and the description thereof is omitted.

In transmitting apparatus 300, subcarrier arrangement section 313, null insertion section 314, and IFFT section 316 constitute scalar signal conversion section 310.

[0071] Subcarrier arrangement section 313 receives the input OFDM primary modulation signal (primary modulation symbol).

The number of primary modulation symbols is arranged on the frequency axis in the order of subcarriers. NULL insertion part 3

Output to 14.

[0072] NULL insertion section 314 inserts NULLs corresponding to the number of input primary modulation symbols. The NULL insertion unit 314 also inserts a DC subcarrier and a guard band NULL as necessary.

FIG. 16 is a diagram for explaining an OFDM signal in which NULLs are arranged by the NULL insertion unit 314.

[0074] As shown in Fig. 16, in the NULL insertion unit 314, the primary modulation symbols are arranged in the positive frequency subcarrier (S1 to S6 in Fig. 16), and NULL (N1 to N6) is assigned to the negative frequency subcarrier. Place.

IFFT section 316 converts the frequency domain OFDM signal that has been null-inserted by null insertion section 314 into a time-domain signal, and outputs the time domain signal to constant envelope transform section 320.

FIG. 17 is a diagram for explaining an OFDM signal after IFFT by the IFFT unit.

[0077] IFFT section 316 IFFTs the signal of only the positive frequency subcarrier. A signal with only positive frequency subcarriers has a complex conjugate symbol at a symmetrical position with respect to DC (complex conjugate symmetry) as shown in Fig. 17, and a complex conjugate inversion symbol at a symmetrical position with respect to DC. It can be thought of as a signal obtained by adding a certain signal (complex conjugate antisymmetric).

Therefore, a signal obtained by IFFT of the IFFT is a sum of a signal obtained by IFFT of the complex conjugate symmetric signal and a signal obtained by IFFT of the complex conjugate antisymmetric signal by the addition theorem of Fourier transform. former Is a signal with only real part due to the nature of Fourier transform, and the latter with only imaginary part

. In other words, the real part of the output is a signal with complex conjugate symbols at symmetrical positions around DC.

(Complex conjugate symmetric) and the imaginary part has a complex conjugate inversion symbol at the symmetric position

It becomes (complex conjugate antisymmetric).

As described above, according to the configuration of the third embodiment, a scalar signal equivalent to that of the second embodiment can be obtained by using either the real part or the imaginary part.

[0080] Therefore, after arranging complex conjugate symbols at symmetrical positions as in the second embodiment, IF

Even without FT, the amplitude is 1Z2, but the same signal is generated.

[0081] Since constant envelope conversion section 320 and thereafter are exactly the same as those in the second embodiment, description thereof is omitted.

Transmitting apparatus 300 according to the third embodiment transmits complex conjugate symbols (some! / Is an inversion thereof) as it is, corresponding to double repetition transmission, similar to transmitting apparatus 200 according to the second embodiment.

[0083] (5) Embodiment 4

In the third embodiment, the complex conjugate symbol at the symmetric position is transmitted as it is as equivalent to the double Repetition transmission, whereas in the fourth embodiment, the OFDM signal is transmitted as a symmetric position symbol without the restriction of the double Repetition. Is transmitted.

The transmission apparatus of the fourth embodiment is used for OFDM, and has the same block configuration as that of the transmission apparatus 300 of the third embodiment shown in FIG. That is, the transmission apparatus of the fourth embodiment is similar to the third embodiment in that the subcarrier arrangement unit 313, the NULL insertion unit 314, the IFFT (Inverse. Fast Fourier Transform) unit 316, the constant envelope conversion units 320, 07 Eight converters (“0 eight”) 131, 132, low-pass filters (“1 ^”) 141, 142, quadrature modulators 151, 152, nonlinear amplifiers 161, 162, combiner (lossless combiner) 170 Have

In the transmitting apparatus 300 of the third embodiment as described above, the processing of the NULL insertion unit and the number of IFFT points in the processing of the IFFT unit are different from those of the transmitting apparatus of the fourth embodiment. Hereinafter, only the difference between the configuration of transmitting apparatus 300 of Embodiment 3 and the configuration of transmitting apparatus 300 of Embodiment 3 will be described.

[0086] FIG. 18 is used to explain the OFDM signal in which NULL is inserted in the fourth embodiment. FIG.

As shown in FIG. 18, NULL insertion section 314 in the transmission apparatus of Embodiment 4 sets NULL so that the primary modulation symbols (S11 to S16) become a part of the positive frequency (Low-IF signal).揷 Enter.

[0088] Also, IFFT section 316 in the transmission apparatus of Embodiment 4 converts the frequency domain OFDM signal into which N ULL has been inserted by the NULL insertion section, into a time domain signal and outputs the time domain signal to constant envelope conversion section 320. . The IFFT unit in Embodiment 4 performs IFFT with a sufficiently large number of points with respect to the number of primary modulation symbols.

FIG. 19 is a diagram for explaining the OFDM signal converted by IFFT section 316 in the fourth embodiment.

[0090] As shown in FIG. 19, the frequency spectrum of the IFFT signal is the same as in Embodiment 3, but the real part is a scan with complex conjugate symbols (S11 * to S16 *) at symmetrical positions around DC. The imaginary part becomes a spectrum with a complex conjugate inversion symbol at the symmetrical position. Note that the complex conjugate inversion symbol at the target position is obtained by inverting the positions of S11 * to S16 * and becomes —S11 * to S16 *, but is omitted in FIG. In this way, by using either the real part or the imaginary part (hereinafter also referred to as “Img”) of the OFDM signal output from the IFFT unit 316, a low-IF signal (scalar signal) ) Can be obtained.

[0091] The constant envelope converter and subsequent parts are exactly the same as in the first to third embodiments.

Note that, in the transmission apparatus of the fourth embodiment, the frequency of the Lo signal is set to a frequency that is offset by the Low-IF frequency with respect to the transmission frequency (center). In addition, the quadrature modulation unit removes and outputs a component different from the transmission frequency from the sum (Lo + IF) or difference (Lo – IF) frequency components. In this case, filtering is performed with an analog filter, so the IF frequency is set high enough for removal.

FIG. 20 shows the final output OFDM signal when using the sum component without the constraint of the 2 × Repetition, which is finally output in the quadrature modulation unit.

[0094] (6) Embodiment 5

The fifth embodiment is limited to OFDM, and the transmission apparatus according to the fourth embodiment is modified. Good shape. In the fourth embodiment, unlike the case where either the real part or the imaginary part output from the IFFT part is used (in the above, the real part is used), the IFF T part is used in the fifth embodiment. The real part (Real Part) and imaginary part (Img Part) output from 416 are used.

In this fifth embodiment, the image envelope configuration is removed by removing the image component when the constant envelope signals Sl and S2 performed in the fourth embodiment are orthogonally modulated and upconverted to the RF band. Is what you do.

FIG. 21 is a block diagram showing a configuration of transmitting apparatus 500 according to Embodiment 5 of the present invention.

A transmission apparatus 500 shown in FIG. 21 shows a modification of transmission apparatus 200 in Embodiment 2, and includes subcarrier arrangement section 313, NULL insertion section 414, IFFT (Inverse. Fast Fourier Transform) section. 416, constant envelope converter 521, 523, DZA converter (“DZA”) 131 to 134, low-pass filter (“: LPF”) 141 to 144, quadrature modulator 551 to 554, nonlinear amplifier 561 to 564, lossless It has combiners 571 and 575, a combiner 580, and a π Z2 phase shifter 590. Note that this transmitting apparatus 500 has the same basic configuration as that of transmitting apparatus 300 corresponding to Embodiment 3 shown in FIG. 15, and the same components are given the same names and reference numerals for explanation. Omitted.

Transmitting apparatus 500 includes subcarrier arrangement section 313, NULL insertion section 414, IFFT section 416 power S scalar signal conversion section 510.

In transmitting apparatus 500, the configuration from subcarrier arrangement section 313 to IFFT section 416 is the same as that in Embodiment 4, and the same processing is performed.

[0100] That is, in transmitting apparatus 500, subcarrier arrangement section 313 to which an OFDM primary modulation signal is input arranges a primary modulation symbol at a part of the positive frequency of the OFDM primary modulation signal, and a NULL insertion section Output to 414.

[0101] NULL insertion section 414 inserts NULL so that the primary modulation symbol becomes a part of the positive frequency (Low_IF signal), and outputs the result to IFFT section 416.

[0102] IFFT section 416 has frequency domain OF in which NULL is inserted by NULL insertion section 414.

The DM signal is converted into a time domain signal and output to the constant envelope conversion units 521 and 523, respectively. Note that IFFT section 416 has an IFF with a sufficiently large number of points relative to the number of primary modulation symbols. Doing T.

More specifically, IFFT section 416 generates a Low-IF signal by performing IFF on the input signal. The real part of the IFFT output by IFFT section 416 is a spectrum with a complex conjugate symbol at a symmetric position around DC, and the imaginary part is a spectrum with a complex conjugate inversion symbol at a symmetric position.

IFFT section 416 outputs the real part (Real Part) to constant envelope conversion section 521 and the imaginary part (Img Part) to constant envelope conversion section 523.

[0105] The constant envelope converter 521 is composed of D / A 131-1, 131-2, 132-1, 132-2, LPF1 41-1, 141-2, 142-1, 142-2, quadrature modulator Together with 551 and 552, nonlinear amplifiers 561 and 562, and a combiner (lossless combiner) 571, an amplification system 520A for the real part is configured. The constant envelope converter 523 consists of D / A133-1, 133-2, 134-1, 1, 134-2, LP F143-1, 143-2, 144-1, 1, 144-2, quadrature modulator 553, 554 The non-linear amplifiers 563 and 564 and the synthesizer (lossless synthesizer) 575 constitute an amplification system 520B for the imaginary part.

[0106] Using the amplification system 520A for the real part and the amplification system 520B for the imaginary part, in the fifth embodiment, the real part (real part) and imaginary part (Img part) of the IFFT output from the IFFT part 416 are used. ) Is generated to generate a constant envelope signal for LINC amplification.

[0107] The multiplexing unit 580 is arranged at the final stage, synthesizes the LINC-amplified signal using the amplification system 520A for the real part and the amplification system 520B for the imaginary part, and outputs it as a transmission signal.

By the way, in the transmission device 500, the Lo signal phase used for the quadrature modulation of the constant envelope signal differs between the amplification system 520A for the real part and the amplification system 5 20B for the imaginary part, and the imaginary part side is shifted by πΖ2 phase shift. The phase is shifted by + πZ2 [rad] by the part 590.

That is, in transmitting apparatus 500, IFFT output is processed by removing the phase rotation (+ π 2) of the imaginary part of the amplification part of the imaginary part, which is the real part + j * imaginary part.

[0110] For this reason, the quadrature modulation units 553 and 554 in the amplification system 520Β with respect to the imaginary part use the π Ζ 2 phase shift unit 590 for the input signal in the quadrature modulation, and perform the corresponding phase rotation. I can see you. Figure 22 shows the spectrum of the LINC amplified signal.

FIG. 22 shows two amplification systems 520A and 520B in transmitting apparatus 500 of the fifth embodiment. It is a figure where it uses for description of the spectrum of the LINC-amplified signal.

As shown in FIG. 22, in the output of the real part side amplification system 520A in the transmission device 500, the difference frequency component becomes a complex conjugate signal of the sum frequency component, and in the output of the imaginary part side amplification system 520B. The difference frequency component becomes a complex conjugate inversion signal of the sum frequency component.

Therefore, when both signals are combined by combining section 580, the difference frequency component is canceled and only the sum component is output.

[0114] According to the fifth embodiment, the signal having the sum frequency component output from the real part side amplification system 520A and the difference frequency component that is a complex conjugate signal thereof, and the imaginary part side amplification system 520B Since the signal having the sum frequency component to be output and the difference frequency component which is the complex conjugate inversion signal is synthesized by the multiplexing unit 580, the difference component is canceled and only the sum component is output. Therefore, when transmitting an OFDM signal, the same effect as in the first embodiment can be obtained.

[0115] In transmission apparatus 500 of the fifth embodiment, amplification system 520B for the imaginary part operates in the same manner as amplification system 520A for the real part, and πΖ2 phase shift unit 590 performs quadrature modulation units 553 and 554. Although the phase difference is adjusted, the constant envelope conversion unit 523 for the imaginary part may be configured to add the phase difference to thereby eliminate the πΖ2 phase shift unit 590.

In this way, in the configuration of transmitting apparatus 500 of the fifth embodiment, a configuration in which ππ2 phase shift section 590 is omitted will be described as a modified example of transmitting apparatus 500.

FIG. 23 shows the operation of constant envelope conversion section 523 for imaginary part with phase difference added (see FIG. 21) when π Ζ2 phase shift section 590 is omitted in transmitting apparatus 500 of Embodiment 5. FIG.

[0117] In the modification of the transmission apparatus 500 in which the π 5002 phase shift unit 590 is omitted in the transmission apparatus 500, the operation of the constant envelope conversion unit 523 on the imaginary part side is different.

[0118] That is, in the constant envelope conversion unit 523 in the transmission device 500 having the π Ζ 2 phase shift unit 590, similarly to the constant envelope conversion unit 521, the IFFT output is set as the I axis, and the signal is multiplexed on the Q axis to be constant envelope In the modified example of the transmitter 500 without the π Ζ2 phase shift unit 590, the IFFT output is used as the Q-axis signal, and the signal is multiplexed on the I-axis to form a constant envelope. I do.

[0119] Specifically, the constant envelope conversion unit 523 on the imaginary part side in this modification example (see Fig. 21) uses the imaginary part signal of the input IFFT part 416 as the Q-axis signal, and sets A on the I-axis side. (t), — A (t) is multiplexed to generate constant envelope signals SI and S2. SI and S2 correspond to the constant envelope signals S (t) and S (t) described in the basic principle in the present embodiment.

1 2

[0120] S (t) and S (t) generated in the constant envelope converter 523 in the modified example are expressed by the following equations (

1 2

6) and (7).

[0121] S (t) = A (t) + j * S (t) "'(6)

S (t) = —A (t) + j * S (t) "-(7)

2

As shown in FIG. 23, in the processing on the imaginary part side in the modification, the phase difference of + ππ2 is increased with respect to the processing on the real part side due to the operation of the constant envelope conversion unit 523 for the imaginary part to which the phase difference is added. Therefore, there is no need to cover the phase rotation by the quadrature modulation units 553 and 554, and the πΖ2 phase shift unit 590 is not necessary.

[0122] (7) Embodiment 6

The basic idea of the sixth embodiment is that the transmission devices 100, 200, 30 of the above-described embodiments.

At 0 and 500, the local leak of the quadrature modulation unit is reduced.

[0123] In the mixer circuit of the quadrature modulation unit in each of the transmission devices 100, 200, 300, and 500, a local leak in which the Lo signal leaks to the output occurs. In the sixth embodiment, it is assumed that the local leak reduction configuration is applied to the second embodiment.

[0124] FIG. 24 is a block diagram showing a configuration of transmitting apparatus 600 according to Embodiment 6 of the present invention.

[0125] In transmission apparatus 600, coordinate transformation section 611 transforms IQ plane coordinates of subcarrier symbols for an OFDM baseband signal that is first-order modulated by QPSK, 16Q AM, etc., in the same manner as coordinate transformation section 211. Then, the data is output to the subcarrier arrangement unit 613. The subcarrier arrangement unit 613 performs the same processing as the subcarrier arrangement unit 213.

That is, subcarrier arrangement section 613 assigns primary modulation symbols to subcarriers using the primary modulated OFDM baseband signal. Further, the subcarrier arrangement unit 613 uses the OFDM baseband signal subjected to coordinate conversion from the coordinate conversion unit 611 to The converted symbol and its original symbol are assigned to subcarriers located symmetrically across DC.

IFFT section 616 converts the frequency domain OFDM signal input from subcarrier arrangement section 213 into a time domain signal, and outputs the real part (Real Part) to constant envelope conversion section 620.

The coordinate conversion unit 611, the subcarrier arrangement unit 613, and the IFFT unit 616 constitute a scalar signal conversion unit 610.

[0129] The constant envelope conversion unit 620 generates two constant envelope signals SI and S2 by multiplexing the scalar signal with the orthogonal component of the scalar signal of the real part in the OFDM signal input from the IFFT unit 616, and outputs it. . At this time, the phase of the constant envelope signal S2 is inverted by the inversion units 625-1, 625-2 and output to the DZA conversion units 632-1, 632-2.

[0130] Then, for the constant envelope signal S2, the quadrature modulation unit 652, the DZA conversion unit 632, LP

For the signal input via F642, the Lo signal phase is inverted to perform quadrature modulation and output to the nonlinear amplifier 662.

[0131] As described above, in the transmission device 600, by inverting one phase of the constant envelope signals Sl and S2, and by inverting the Lo signal phase of quadrature modulation, the constant envelope signal is converted into the original phase. However, the phase of the local leak signal is reversed between S1 and S2. Therefore, cancellation occurs due to multiplexing, and the amount of leakage can be reduced.

[0132] In the transmission apparatus 600 shown in Fig. 24, the S2 side is inverted, but the S1 side may be inverted. Of course, the basic idea of the sixth embodiment can be applied not only to the second embodiment but also to other embodiments.

(8) Embodiment 7

The transmitting apparatus of the seventh embodiment shows a configuration in which the IFFT circuit calculation amount is reduced in each of the second, third, and fourth embodiments described above, and only the configuration of the IFFT unit is different.

That is, in Embodiments 2, 3, and 4, the OFDM signal is generated using only one of the IFFT outputs. In this case, the calculation amount can be reduced. The case where only the real part is used will be described.

FIG. 25 is a diagram for explaining the IFFT unit in the transmission apparatus according to the seventh embodiment.

[0136] IFFT section 716 shown in Fig. 25 performs the following processing. [0137] If the input of the N-point IFFT is z (n) = x (n) + jy (n) (where n = l to N), the IFFT output d (k) is expressed by the following equation (8). It is.

[0138] [Equation 1]

χ () cos (—2 mk / N ヽ一) n) sini -2mik I N)

+ j {x (n) sin (-2mik / N) + y (n) cos (-2 mk / N)}

... (8)

[0139] In this equation (8), since the imaginary part is not used, the calculation is omitted, and the following equation (9) is obtained. Thereby, the amount of calculation can be halved.

[0140] [Equation 2]

(9)

[0141] IFFT section 716 using such a calculation method is replaced with IFFT sections 116, 21 of the second, third, and fourth embodiments.

By applying to 6, 416, the configuration of the seventh embodiment is obtained.

[0142] The embodiment of the present invention has been described above.

[0143] The transmission apparatus according to the present invention is not limited to the above embodiments, and can be implemented with various modifications.

[0144] Although a case has been described with the above embodiment as an example where the present invention is configured with nodeware, the present invention can be implemented with software. For example, an algorithm of the output amplification method according to the present invention is described in a program language, and this program is stored in a memory and executed by information processing means, thereby realizing the same function as the apparatus according to the present invention. be able to.

Industrial applicability

[0145] The transmission device according to the present invention is easy to reduce in size, cost and power consumption, and retains the linearity of a signal wave having an envelope variation even when the signal band is limited, And, It has the effect of amplifying a transmission signal with high power efficiency and is useful as a device used in an OFDM wireless transmission device.

Claims

The scope of the claims

[1] An LINC method amplifying apparatus that decomposes an IQ complex baseband signal to be transmitted into a plurality of constant envelope signals, and synthesizes each after nonlinear amplification,

A scalar signal converting means for converting the IQ complex baseband signal to a scalar signal, and a constant envelope for generating the first and second constant envelope signals by multiplexing the signals on the orthogonal components of the converted scalar signal Signal generating means;

First and second nonlinear amplifiers for amplifying the first and second constant envelope signals, respectively;

And a synthesizer for synthesizing signals obtained by the first and second nonlinear amplifiers.

2. The transmission device according to claim 1, wherein the scalar signal converting means is a digital quadrature modulator that digitally upconverts the IQ complex baseband signal and converts it to a Low-IF digital signal.

[3] The IQ complex baseband signal is an OFDM primary modulation signal,

The scalar signal conversion means includes

Coordinate conversion means for converting the OFDM primary modulation symbol into an I-axis symmetric symbol, a Q-axis symmetric symbol, an IQ interchanged symbol, or an IQ interchanged and inverted symbol.

Subcarrier arrangement means for arranging the symbol transformed by the coordinate transformation means and the symbol before coordinate transformation on subcarriers at positions symmetrical to each other across the DC component, and symbols arranged by the subcarrier arrangement means Frequency / time conversion means to convert to time domain signal,

The transmitting device according to claim 1, further comprising:

[4] The IQ complex baseband signal is an OFDM primary modulation signal,

The scalar signal conversion means includes

Subcarrier arrangement means for arranging the OFDM primary modulation symbols only on subcarriers in one direction, positive or negative with respect to the DC component! Null insertion means for arranging nulls on subcarriers that are not arranged with symbols by the subcarrier arrangement means;

A frequency and time conversion means for converting a symbol obtained by the subcarrier arrangement means and the null insertion means into a signal in a time domain;

The transmitting device according to claim 1, further comprising:

The IQ complex baseband signal to be transmitted is decomposed into a plurality of constant envelope signals, each is nonlinearly amplified, and then synthesized and transmitted. Is converted to a scalar signal,

By multiplexing the signal to the orthogonal component of the converted scalar signal, the first and second constant envelope signals are generated,

Nonlinearly amplifying the first and second constant envelope signals,

A transmission method for combining and transmitting the first and second constant envelope signals subjected to nonlinear amplification.