WO2008029459A1

WO2008029459A1 - Radio receiving apparatus and radio communication system

Info

Publication number: WO2008029459A1
Application number: PCT/JP2006/317662
Authority: WO
Inventors: Katsuya Oda; Takashi Enoki
Original assignee: Panasonic Corporation
Priority date: 2006-09-06
Filing date: 2006-09-06
Publication date: 2008-03-13

Abstract

A radio receiving apparatus wherein AFC between a base station and the terminal can be accomplished and a phase noise characteristic can be improved, thereby ensuring a high communication quality. In this apparatus, a quadrature demodulator (257) uses a pilot signal from an amplifier (260) to perform a quadrature demodulation of a signal from a delay corrector (256) and cancel the phase noise. A frequency deviation amount determining part (270) uses a signal distributed by a second distributor (258) to determine the amount of a frequency deviation from a reference frequency of a transmitting base station. A frequency control part (262) controls, in accordance with the frequency deviation amount, a frequency-variable reference signal reception oscillator (210) and a frequency-variable reference signal transmission oscillator (211) to have respective output oscillation frequencies synchronized with the reference frequency of the transmitting base station.

Description

Radio receiving apparatus and radio communication system

Technical field

TECHNICAL FIELD [0001] The present invention relates to a radio reception apparatus having a phase noise cancellation function and an automatic frequency control (AFC) function, and a radio communication system.

Background art

Conventionally, various measures have been taken in order to provide a wireless communication system with excellent phase noise characteristics, and the technology of this type of wireless communication system is disclosed in Patent Document 1, for example. FIG. 1 is a block diagram showing an example of a wireless receiver in a conventional wireless communication system with improved phase noise characteristics. The radio receiver shown in Fig. 1 has a local noise canceller to improve the phase noise characteristics. FIG. 2 is a characteristic diagram showing frequency characteristics of each component of the local noise canceller in the wireless reception device of FIG. In other words, the symbol of each black circle in Fig. 1 corresponds to the characteristic diagram of each symbol in Fig. 2.

Therefore, the operation of the local noise canceller in the wireless reception device of FIG. 1 will be described with reference to FIG. The input signal (A) input to the wireless receiver shown in Fig. 1 is multiplexed with the modulated IF signal (BBT-OFDM) and pilot 'carrier (PI LOT) as shown in Fig. 2 (A). It is assumed that the input phase noise (thick diagonal line) is superimposed. Where the pilot 'carrier (PILOT) frequency in the input signal is f, IF

PLT

If the frequency of the signal (BBT—OFDM), that is, the input signal is f and the input phase noise is Θ (t), the input phase noise 0 (t) is superimposed on f and f. Is it

PLT sig PLT sig is expressed as follows.

f Z Θ (t)

PLT

f Z Θ (t)

sig

[0004] The input signal (A) is distributed by distributor 1, one is output to the pilot branch, and the other is output to the signal branch. In the pilot branch, one of the signals distributed by the distributor 1 is band-limited by the bandpass filter (BPF) 2 and pilot 'carrier (PIL Only the component OT) passes through and is extracted, and is further amplified by the limiter amplifier 3. At this time, the IF signal component is removed from the frequency characteristics of the output signal (Β) output from BPF2 and the output signal (C) output from limiter amplifier 3, as shown in Fig. 2 (B'C). Only the pilot carrier (PILOT) component and the input phase noise Θ (t) superimposed on it.

[0005] At this time, a delay occurs in BPF2, and if this delay time is τ, the input pilot key

BPF1

Carrier frequency f (t— τ)

PLT has a delay time τ

Input phase noise delayed by BPF1 0

Since BPF1 is superimposed !, the incoming canolot carrier frequency f is expressed by the following equation.

PLT

f Z Θ (t- τ)

PLT BPF1

On the other hand, in the signal branch distributed by distributor 1, local oscillator signal (D) is output from local oscillator 4. Here, the frequency characteristics of the local oscillation signal (D), in which the local oscillator 4 power is also output, are as shown in Fig. 2 (D), and the local oscillation phase (LO) signal superimposed on the local oscillation frequency (LO) signal. Noise. Here, the local oscillation signal frequency in the system is expressed as f

Let LO be the local oscillation signal phase noise in the system, and let Φ (t) be the local oscillation signal frequency f in the system.

Since LO is superimposed with local oscillation signal phase noise Φ (t) in the system, local oscillation signal frequency f

LO

Is expressed as:

f

LO)

[0007] In the signal branch, the input signal output from the distributor 1 is frequency-converted by the multiplier (mixer) 5 with the local oscillator signal (D) of four local oscillators, and the multiplier 5 Signal (E) is output. Here, the frequency characteristics of the signal (E) output from the multiplier 5 are the sum and difference components of the input signal (A) and the local oscillation signal (D) as shown in Fig. 2 (E). Exists. Therefore, the relationship between each signal component included in the signal) and the superimposed phase noise is as follows.

f -f Ζ Θ (t)-(t)

PLT LO

f f Z Θ (t)-(t)

sig LO

f + f Z Θ (t) + (t)

PLT LO

f + f Z Θ (t) + (t)

sig LO

[0008] The frequency-converted signal (E) has only a difference component by a bandpass filter (BPF) 6. Since the band is limited to pass, the signal (F) is output from BPF6. In the frequency characteristics of this signal (F), as shown in Fig. 2 (F), the sum component in the signal) is removed and only the difference component exists. At this time, a delay occurs in BPF6.

BPF

Assuming 2, the phase noise superimposed on the extracted difference component has a delay time τ

A delay occurs only for BPF2. At this time, the relationship between each signal component included in the signal (F) and the superimposed phase noise is as follows.

f -f Z Θ (t- τ)-(t- τ)

PLT LO BPF2 BPF2

f -f Ζ θ (t- τ)-(t- τ)

sig LO BPF2 BPF2

[0009] Then, a delay amount is added to the signal) in the delay corrector 7 so as to be equivalent to the delay time in the BPF2 of the pilot branch, and the signal (G) is output from the delay corrector 7. Here, with respect to the delay time τ of the bandpass filter (BPF) 2, the bandpass filter

BPF1

(BPF) 6 delay time is τ and delay corrector 7 delay time is At.

BPF2

τ = τ + At

BPFl BPF2

Thus, the delay compensator 7 adds a delay At to the signal) to equalize the delay time difference from the pilot branch.

[0010] As a result, the frequency characteristic of the signal (G) does not change, and the waveform is as shown in Fig. 2 (G). The relationship between each signal component included in the signal (G) and the superimposed phase noise Is obtained by adding the delay Δt to the phase noise.

f -ί Ζ Θ (t- τ At)-(t- τ-At)

PLT LO BPF2- BPF2

f f Z Θ (t- τ -At)-(t- τ-At)

sig LO BPF2 BPF2

[0011] The signal (G) of the signal branch and the pilot branch signal (C) output from the limiter amplifier 3 are frequency-converted by the frequency change 8, and the signal (H) Is output.

[0012] Here, the frequency characteristic of the signal (H) output from the frequency shift 8 is the sum and difference components of the signal (G) and the signal (C) as shown in Fig. 2 (H). Exists. Therefore, the relationship between each signal component contained in signal (H) and the superimposed phase noise is as follows. f — (f -f) Ζ θ (t-τ)-{0 (t- T -At)-(t- T -At)}

PLT PLT O BPFl BPF2 BPF2

f — (f f) Z Θ (t-τ)-{0 (t- T -At)-(t- T -At)}

PLT sig LO BPFl BPF2 BPF2 f + (f -f) Z Θ (t- τ) + {0 (t- T -Μ) -φ (ί ~ τ -At)}

PLT PLT O BPF1 BPF2 BPF2

f + (f f) Z Θ (t- τ) + {0 (t- T -At)-(t- T -At)}

PLT sig LO BPF1 BPF2 BPF2

[0013] Here, as described above, the delay time of the delay corrector 7 is

τ = τ + At

BPFl BPF2

Thus, the delay Δt is added to equalize the delay time difference between the signal branch and the pilot branch, so the above equation can be rearranged as follows.

f Z (t- τ-At)

LO BPF2

f one (f one f) Z (t- τ-At)

LO sig PLT BPF2

2Xf f Z2X Θ (t- τ)-(t- τ-At)

PLT LO BPFl BPF2

f + (f -f) Z2X Θ (t- τ)-(t- τ At)

PLT sig LO BPFl BPF2

[0014] Here, focusing on the difference component, the frequency of the output signal component is the frequency of the local oscillation signal (f) in the system related to the frequency of the input signal.

LO, that is, constant. In addition, when the pilot 'carrier is focused, the sideband of the signal is inverted at the input and output. Also, the phase noise of the output signal is canceled by the input phase noise Θ (X) and becomes the phase noise φ (X) of the local oscillation signal in the system instead. That is, the phase noise φ of the local oscillation signal in the system

If (X) is sufficiently small, the phase noise of the input signal is sufficiently reduced and output.

[0015] Therefore, the signal (H) frequency-converted by the frequency change ^^ 8 is band-limited so that only the difference component and only the signal component pass through the band-pass filter (BPF) 9, and from BPF 9, Signal (I) is output. As shown in Fig. 2 (1), the frequency characteristics of this signal (I) are such that only the signal component of the difference component exists by removing the pilot / carrier component in the sum and difference components of signal (H). . At this time, the relationship between the signal component included in the signal (I) and the superimposed phase noise is as follows.

f one (f one f) Ζ (t- τ-At)

LO sig PLT BPF2

[0016] Due to the frequency synchronization and noise removal principle of the above local noise canceller, for example, even if there is a frequency deviation in the input signal, the local oscillation frequency having high stability and high frequency generated by the local oscillator 4 Since an output signal with a frequency according to is obtained, the frequency deviation of the input signal can be eliminated. The phase noise of the output signal is The phase noise Θ (X) superimposed on the input signal is canceled and instead only the phase noise φ (X) of the local oscillation signal in the system, so the phase noise φ (X) of the local oscillation signal in the system If X) is sufficiently small, the phase noise of the input signal is sufficiently reduced and output.

[0017] Also, in the mobile communication field, in order to avoid characteristic deterioration due to a difference in local oscillation frequency between the base station and the terminal, the difference between the frequency of the signal received from the base station and the local oscillation frequency of the terminal Terminal frequency is also fed back to the base station, and automatic frequency control (AFC) is required to synchronize the local oscillation frequency of the terminal with the base station reference frequency.

The following methods are known as methods for realizing general AFC control. In other words, using a known replica symbol or the like, a difference between the reception frequency from the base station and the local oscillation frequency of the terminal is detected in the baseband part of the terminal, and is fed back to the base station using an uplink transmission signal. At the same time, the reference frequency of the reference oscillator of the terminal is shifted according to the detected frequency deviation amount, and the oscillation frequency between the base station and the terminal is synchronized.

Patent Document 1: JP 2002-152158 A

Disclosure of the invention

Problems to be solved by the invention

[0019] However, in the AFC control method in the conventional wireless communication system, since the signal is received by the terminal antenna and the received frequency shift is canceled before the input of the baseband unit, The frequency deviation amount is always 0 at the time of input of the band portion, and it becomes impossible to detect the frequency deviation information that should originally be obtained. Therefore, the actual frequency shift amount becomes unclear, and frequency synchronization between the base station and the terminal cannot be obtained, resulting in inferior characteristics.

[0020] Hereinafter, the characteristic deterioration caused by AFC will be described in more detail with reference to FIG. 3 and FIG. Fig. 3 is a system configuration diagram when the AFC technology is applied when the phase noise cancellation technology is not used in the conventional wireless communication system. Fig. 4 is a system configuration diagram when the AFC technology is applied when the phase noise cancellation technology is used in the conventional wireless communication system. Note that in both FIG. 3 and FIG. 4, only the radio reception unit of the terminal is shown by simplifying the configuration of the entire radio communication system.

First, the frequency when the phase noise cancellation technique is not used in the wireless communication system of FIG. The operation of number synchronization will be described. The description of the same components as those described so far is omitted.

A radio reception apparatus 10 shown in FIG. 3 includes a reception unit 11, a reception antenna 30, a transmission unit 12, and a transmission antenna 31. The receiving antenna 30 receives a wireless signal (RF signal) from the wireless transmission device of the communication partner, and the transmission antenna 31 transmits the wireless signal (RF signal) to the wireless transmission device of the communication partner.

The receiving unit 11 includes a first receiving baseband unit 13 including a receiving baseband processing unit 14 and a frequency shift amount detecting unit 16, a frequency variable reference signal oscillator 18, a first receiving local oscillation unit 21, and an amplifier 51. A multiplier 52, a band-pass filter 53, an amplifier 54, a quadrature demodulator 57, and a frequency control unit 62.

In addition, the transmission unit 12 includes a first transmission baseband unit 15, a third transmission local oscillation unit 23, a fourth transmission local oscillation unit 24, a quadrature modulator 80, a multiplier 81, and an amplifier 82. I have.

In an actual environment, a signal is received by the reception antenna 30 of the wireless reception device 10 in a state in which a frequency shift is included due to the influence of fading or the like. At this time, it is assumed that the signal frequency when the receiving antenna 30 is receiving a signal is represented by the following equation.

frx = fo + Af

However,

frx: Signal frequency under antenna reception

f

0: Original signal frequency

Δ f: Amount of frequency deviation generated in the propagation path

[0026] The signal having the signal frequency frx is subjected to predetermined processing such as amplification and frequency conversion by the amplifier 51, the multiplier (mixer) 52, the band-pass filter 53, and the amplifier 54, and the second receiving local oscillation unit The signal oscillated at 22 is input to the quadrature demodulator 57 and demodulated into a baseband signal. At this time, since the frequency oscillated by the second reception local oscillation unit 22 is initially set to fo, the signal frequency at the time of input to the reception baseband processing unit 14 is Δ1 ”. The signal frequency Δ1 ″ needs to be corrected because it is a difference between the reference frequency of the transmitter 12 and the reference frequency of the receiver 11. Therefore, the frequency shift amount Ai¾S is detected by the frequency shift amount detection unit 16 in the first reception burst band unit 13. No detected frequency The information of the amount Δ ί is transferred to the frequency control unit 62, and the frequency control unit 62 performs control to shift the frequency of the frequency variable reference signal oscillator 18 in the reception unit 11 by Af. As a result, the oscillation frequencies of the first reception local oscillation unit 21 and the second reception local oscillation unit 22, and the third transmission local oscillation unit 23 and the fourth transmission local oscillation unit 24 are shifted by Δί. The transmitter 11 and the transmitter 12 can be synchronized. In this way, automatic frequency control (AF C) is realized. However, the AFC control method described here is an example, and AFC can be realized by other control methods. What is important is that the baseband signal demodulated by the quadrature demodulator 57 is input to the reception baseband processing unit 14 in a state including the amount of frequency shift (here, Δf).

Next, the operation of the AFC when the phase noise cancellation technique is used for mobile communication in the wireless communication system of FIG. 4 will be described.

The radio reception device 10a in FIG. 4 includes a reception unit lla, a reception antenna 30, a transmission unit 12a, and a transmission antenna 31. The receiving antenna 30 receives a wireless signal (RF signal) from the wireless transmission device of the communication partner, and the transmission antenna 31 transmits the wireless signal (RF signal) to the wireless transmission device of the communication partner.

[0029] The reception unit 11a includes a reception baseband processing unit 14 and a frequency shift amount detection unit 16, a first reception baseband unit 13, a frequency variable reference signal oscillator 18, a first reception local oscillation unit 21, and an amplifier 51, a multiplier 52, a band pass filter 53, an amplifier 54, a distributor 55, a delay corrector 56, a quadrature demodulator 57, a band pass filter (BPF) 59, an amplifier 60, and a frequency control unit 62.

[0030] The transmission unit 12a includes a first transmission baseband unit 15, a third transmission local oscillation unit 23, a first transmission baseband unit 15,

4 transmission local oscillators 24, a quadrature modulator 80, a multiplier 81, and an amplifier 82.

[0031] The input signal input to the reception unit 11a of the wireless reception device 1 Oa via the reception antenna 30 is represented by the following equation as described above.

frx = fo + Af

However,

frx: Signal frequency under antenna reception

f: Original signal frequency Δ f: Amount of frequency deviation generated in the propagation path

[0032] This input signal is subjected to amplification and frequency conversion processing, and then distributed to the signal branch and pilot branch by the distributor 55. When the quadrature demodulator 57 performs quadrature demodulation according to the operation principle described in FIGS. All phase noise will be cancelled. However, what is noted here is the relationship between the signal signal frequency at FF in the diagram of Fig. 4 and the pilot signal frequency at EE in the diagram. Since the signal signal frequency at FF remains the input signal of the receiving antenna 30, the following equation is obtained.

fFF = fl + Af

However,

IFF: Signal frequency at FF in Figure 4

fl: Value converted from the original signal frequency by the oscillation frequency of the first receiving local oscillator 21

Δ f: Amount of frequency deviation generated in the propagation path

[0033] Also, in the figure, the pilot signal frequency at EE is given by the following equation.

fEE = fl + Af

However,

EE: Signal frequency at EE in Figure 4

Δ f: Amount of frequency deviation generated in the propagation path

Here, since the orthogonal demodulation unit 57 performs frequency conversion using the signal signal frequency fFF and the pilot signal frequency ffiE, the frequency of the input signal input to the reception baseband processing unit 14 is always the following. It is expressed by the following formula.

1BB = 0

However,

1BB: Signal frequency at output of quadrature demodulator 57 (0 = baseband frequency)

[0035] As this expression force is also divided, the term of the frequency shift amount Δ1 "disappears, that is, the reception baseband processing unit 14 does not include the frequency shift amount generated in the propagation path. It will be Therefore, it becomes impossible to supply information on the frequency deviation amount to the frequency control unit 62, and as a result, it becomes impossible to vary the frequency of the frequency variable reference signal oscillator 18. Therefore, it is impossible to control the AFC with a wireless receiver having a conventional configuration as shown in FIG.

[0036] An object of the present invention is to provide a radio receiver capable of realizing high frequency and high communication quality by realizing automatic frequency control (AFC) between a base station and a terminal and improving phase noise characteristics. And providing a wireless communication system.

Means for solving the problem

[0037] The radio reception apparatus of the present invention is a radio transmission apparatus as a communication partner, wherein a pilot signal is superimposed and reception means for receiving the transmitted signal, extraction means for extracting the pilot signal from the reception signal, Orthogonal demodulation means for performing orthogonal demodulation on the received signal using the extracted pilot signal, and frequency deviation amount detecting means for detecting the frequency deviation amount of the oscillation frequency of the local oscillator with respect to a reference frequency using the received signal And automatic frequency control means for performing control to synchronize the oscillation frequency of the local oscillator with the reference frequency using the detected frequency deviation amount.

[0038] The radio communication system of the present invention is a radio communication system including a radio transmission device and a radio reception device, and the radio transmission device transmits a radio signal in which a pilot signal is superimposed on the center of a multicarrier signal. The wireless reception device includes a reception unit that receives a signal transmitted from the wireless transmission device, an extraction unit that extracts the pilot signal from the reception signal, and the extraction for the reception signal. Quadrature demodulating means for performing quadrature demodulation using the pilot signal, frequency deviation amount detecting means for detecting the frequency deviation amount of the oscillation frequency of the local oscillator with respect to a reference frequency using the received signal, and the detected frequency deviation And automatic frequency control means for performing control to synchronize the oscillation frequency of the local oscillator with the reference frequency using a quantity.

The invention's effect

[0039] According to the present invention, both the phase noise canceller and the automatic frequency control (AFC) can be achieved, so that it is possible to realize both the low power consumption and the improvement in the reception sensitivity. The frequency of the received local signal can be obtained without using the frequency deviation information of the band signal. It becomes possible to synchronize the wave number and the frequency of the transmission local signal. In addition, since there is no possibility that the pilot signal will deteriorate even if a frequency shift occurs, the phase noise canceling function can be optimally operated. This makes it possible to improve the phase noise characteristics and maintain good communication quality while realizing automatic frequency control between the base station and the terminal.

Brief Description of Drawings

FIG. 1 is a block diagram showing an example of a wireless receiver in a conventional wireless communication system with improved phase noise characteristics

2 is a characteristic diagram showing the frequency characteristics of each component of the local noise canceller in the wireless receiver of FIG.

[Figure 3] System configuration diagram when applying AFC technology without using phase noise cancellation technology in a conventional wireless communication system

[Fig.4] System configuration diagram when applying AFC technology when phase noise cancellation technology is used for mobile communication in a conventional wireless communication system

FIG. 5 is a block diagram showing a configuration of a wireless transmission device of the wireless communication system according to Embodiment 1 of the present invention.

FIG. 6 is a block diagram showing a configuration of a radio reception apparatus of the radio communication system according to Embodiment 1 of the present invention.

7 is a characteristic diagram showing frequency characteristics of each signal in the wireless transmission device shown in FIG. 5 and the wireless reception device shown in FIG.

[Fig.8] Internal configuration of quadrature demodulator shown in Fig.6

FIG. 9 is a block diagram showing a configuration of a radio reception apparatus of the radio communication system according to Embodiment 2 of the present invention.

FIG. 10 is a block diagram showing a configuration of a wireless reception device of the wireless communication system according to the third embodiment of the present invention.

FIG. 11 is a block diagram showing a configuration of a wireless reception device of the wireless communication system according to the fourth embodiment of the present invention.

FIG. 12 shows a configuration of a radio receiving apparatus of a radio communication system according to Embodiment 5 of the present invention. Block diagram

BEST MODE FOR CARRYING OUT THE INVENTION

The wireless communication system of the present invention is provided with a function of further distributing the output signal of the pilot branch in the phase noise cancellation technique to detect the frequency shift amount. Then, the reference oscillation frequency in the receiver is changed according to the amount of frequency deviation detected by this function, and the frequency tracking is performed by setting the input frequency of the frequency converter (mixer) to a state that takes into account the amount of deviation. To realize. This makes it possible to achieve both phase noise cancellation and AFC.

Next, an embodiment of the wireless communication system of the present invention will be described in detail. Note that, in the drawings used in the following embodiments, the same components are denoted by the same reference numerals, and redundant descriptions are omitted as much as possible. Furthermore, when explaining the principle of phase noise cancellation, it is assumed that there is no signal frequency deviation due to the spatial propagation path in order to simplify the explanation.

FIG. 5 is a block diagram showing a configuration of a radio transmission apparatus of the radio communication system according to Embodiment 1 of the present invention. FIG. 6 is a block diagram showing the configuration of the radio reception apparatus of the radio communication system according to Embodiment 1 of the present invention. That is, the radio communication system according to the first embodiment includes radio transmission apparatus 100 shown in FIG. 5 and radio reception apparatus 200 shown in FIG. Note that the radio reception apparatus in FIG. 6 shows a single conversion type in which an input signal is frequency-converted by a local signal.

First, the configuration of radio transmitting apparatus 100 shown in FIG. 5 will be described. The wireless transmission device 100 includes a transmission baseband unit 101 that generates a baseband signal, a transmission unit 105 that performs predetermined processing on the baseband signal and transmits the signal as an RF signal, and an antenna 130. It has become. The transmission baseband unit 101 combines the modulated signal and the pilot signal and transmits them to the transmission unit 105. Transmitter 105 transmits to the outside a radio signal obtained by multiplexing a modulated signal with no signal at the center frequency and a pilot signal having the same center frequency as the center frequency. The antenna 130 sends a radio signal (RF signal) to the outside. I believe.

The transmission baseband unit 101 includes a signal synthesis unit 102, a modulation signal generation unit 103, and a pilot signal generation unit 104. Modulation signal generation section 103 generates a modulation signal such as a multicarrier that does not carry a signal on the center frequency portion, and outputs the modulation signal to signal synthesis section 102. Pilot signal generation section 104 generates a pilot signal and outputs it to signal synthesis section 102. The signal synthesis unit 102 synthesizes the modulation signal (M-CD MA) received from the modulation signal generation unit 103 and the pilot signal (PILOT) received from the pilot signal generation unit 104 and outputs the synthesized signal to the transmission unit 105.

In addition, the transmission unit 105 includes a reference signal oscillator 110, a first transmission local oscillation unit 121, a second transmission local oscillation unit 122, a quadrature modulator 152, a multiplier (mixer) 153, and an amplifier 154. I have. The reference signal oscillator 110 generates a reference signal and outputs it to the first transmission local oscillation unit 121 and the second transmission local oscillation unit 122. First transmission local oscillator 121 generates a local oscillation signal using the reference signal received from reference signal oscillator 110 and outputs the local oscillation signal to multiplier 153. Second transmission local oscillation unit 122 generates a local oscillation signal using the reference signal received from reference signal oscillator 110 and outputs the local oscillation signal to quadrature modulator 152. The quadrature modulator 152 uses the local oscillation signal from the second transmission local oscillation unit 122 to orthogonally modulate the synthesized signal of the modulation signal output from the signal synthesis unit 102 of the transmission baseband unit 101 and the pilot signal. And output to the multiplier 153. Multiplier 153 converts the signal subjected to quadrature modulation by quadrature modulator 152 into a radio signal (RF signal) using the local oscillation signal received from first transmission local oscillation unit 121. Amplifier 154 amplifies the radio signal received from multiplier 153 and outputs the amplified signal to antenna 130.

Next, the operation of radio transmitting apparatus 100 shown in FIG. 5 will be described. In the transmission baseband unit 101, the modulation signal generation unit 103 generates a modulation signal and outputs it to the signal synthesis unit 102. Here, the modulation signal is described as multi-carrier CDMA, the signal is placed on the center frequency portion on the power frequency axis, and any modulation signal can be handled, for example, It may be an OFDM signal or the like.

[0048] Further, the signal synthesis unit 102 synthesizes the modulation signal (M-CDMA) received from the modulation signal generation unit 103 and the pilot signal (PILOT) received from the pilot signal generation unit 104. Transmit to transmission section 105. Note that the pilot signal generated by the pilot signal generation unit 104 is positioned at the center on the frequency axis of the modulation signal, and f = 0 [Hz], where f is the frequency of the pilot signal. ing.

PILOT PILOT

Next, in transmission section 105, second transmission local oscillation section 122 generates a local oscillation signal using the reference signal generated by reference signal oscillator 110, and this local oscillation signal is converted into quadrature modulator 152. Output to. Then, the quadrature modulator 152 uses the local oscillation signal received from the second transmission local oscillation unit 122, and combines the modulated signal output from the signal synthesis unit 102 of the transmission baseband unit 101 and the pilot signal. Are orthogonally modulated and output to the multiplier 153.

Multiplier 153 uses the local oscillation signal received from first transmission local oscillation unit 121 to convert the signal subjected to quadrature modulation by quadrature modulator 152 into a radio signal (RF signal). The radio signal is amplified by the amplifier 154 and then transmitted to the radio reception device via the antenna 130. Note that here, the first transmission local oscillation unit 121 generates a local oscillation signal using the reference signal received from the reference signal oscillator 110, and the first transmission local oscillation unit 121 and the local oscillation signal from the first transmission local oscillation unit 121 The generation of the local oscillation signal by the two transmitting local oscillators 122 is synchronized.

Next, the configuration of radio receiving apparatus 200 shown in FIG. 6 will be described. The radio receiving apparatus 200 includes a receiving unit 201, a receiving antenna 230, a transmitting unit 202, and a transmitting antenna 231. Receiving antenna 230 receives a radio signal (RF signal) from radio transmitting apparatus 100 shown in FIG. 5, and transmitting antenna 231 transmits a radio signal (RF signal) to radio transmitting apparatus 100.

The reception unit 201 includes a first reception baseband unit 203 including a reception baseband processing unit 204, a frequency variable reference signal reception oscillator 210, a first reception local oscillation unit 221, an amplifier 251, and a multiplier 252. , Bandpass filter 253, amplifier 254, first distributor 255, delay corrector 256, quadrature demodulator 257, second distributor 258, bandpass filter (BPF) 259, amplifier 260, frequency control unit 262, and frequency A deviation amount detection unit 270 is provided.

[0053] The transmission unit 202 includes a first transmission baseband unit 205, a frequency variable reference signal transmission oscillator 211, a third transmission local oscillation unit 223, a fourth transmission local oscillation unit 224, and a quadrature modulator. 280, a multiplier 281, and an amplifier 282.

Amplifier 251 amplifies the radio signal (RF signal) received by receiving antenna 230 and outputs the amplified signal to multiplier 252. Multiplier 252 converts the frequency of the radio signal output from amplifier 251 with the local oscillation signal from first reception local oscillation unit 221 and outputs the result to bandpass filter 253. The band pass filter 253 extracts only the signal in the desired frequency band from the signal power frequency-converted by the multiplier 252 and outputs the signal power to the amplifier 254. The amplifier 254 amplifies the signal in the desired frequency band output from the band pass filter 253 and outputs the amplified signal to the first distributor 255. That is, the radio signal (RF signal) received by the receiving antenna 230 is subjected to predetermined processing such as amplification and frequency conversion by the amplifier 251, the multiplier 252, the band-pass filter 253, and the amplifier 254. 1 Input to distributor 255.

[0055] The first distributor 255 distributes the input signal in two directions and outputs the signals to the delay compensator 256 and the second distributor 258, respectively. The second distributor 258 further distributes the signal distributed by the first distributor 255 in two directions, and outputs them to the BPF 259 and the frequency shift amount detection unit 270, respectively. The BPF 259 extracts a signal component corresponding to the pilot signal from one signal distributed by the first distributor 255 and outputs it to the amplifier 260. The amplifier 260 amplifies the signal component corresponding to the pilot signal extracted by the BP F259 and outputs the amplified signal component to the quadrature demodulator 257.

The delay corrector 256 gives a delay to one of the signals distributed by the first distributor 255 and outputs the delayed signal to the quadrature demodulator 257. Quadrature demodulator 257 performs quadrature demodulation by frequency multiplying the output signal of amplifier 260 and the output signal of delay corrector 256, and outputs the result to first reception baseband section 203. The first reception baseband unit 203 performs baseband processing on the baseband signal demodulated by the quadrature demodulator 257 by the reception baseband processing unit 204.

Note that if the input signal level to quadrature demodulator 257 is kept constant only by amplifier 260, distortion occurs only in the pilot branch, and phase noise remains in the output of quadrature demodulator 257. Become. Therefore, the input signal level to the first distributor 255 is Pin [dBm], the power loss due to the first distributor 255 and the second distributor 258 is a [dB], and the power loss of the BPF259 is If the loss is β [dB] and the gain of the amplifier 260 is γ [dB], the input signal level Pin to the first distributor 255 is approximately proportional to the output level of the amplifier 260 (Pin + y-a ~ β). Set to be. Thereby, distortion in the pilot branch can be prevented.

The frequency shift amount detection unit 270 detects the frequency shift amount with respect to the reference frequency of the transmission base station using one of the signals distributed by the second distributor 258, and outputs it to the frequency control unit 262. The frequency control unit 262 controls the frequency variable reference signal reception oscillator 210 and the frequency variable reference signal transmission oscillator 211 based on the frequency shift amount detected by the frequency shift amount detection unit 270.

The frequency variable reference signal receiving oscillator 210 is controlled in output frequency by the frequency control unit 262 and outputs an oscillation frequency signal to the first receiving local oscillation unit 221. The first reception local oscillation unit 221 outputs an arbitrary frequency using the reference oscillation signal supplied from the frequency variable reference signal reception oscillator 210 as a reference.

The frequency variable reference signal transmission oscillator 211 outputs an oscillation frequency signal to the third transmission local oscillation unit 223 and the fourth transmission local oscillation unit 224, the output frequency of which is controlled by the frequency control unit 262. Third transmission local oscillator 223 outputs an arbitrary frequency to the multiplier ² 81 a reference oscillation signal supplied from the variable frequency reference signal transmission oscillator 211 as a reference. The fourth transmission local oscillation unit 224 generates a local oscillation signal using the reference oscillation signal supplied from the frequency variable reference signal transmission oscillator 211, and outputs the local oscillation signal to the quadrature modulator 280. The first transmission baseband unit 205 generates a transmission baseband signal and outputs it to the quadrature modulator 280. The quadrature modulator 280 uses the local oscillation signal from the fourth transmission local oscillation unit 224 to orthogonally modulate the transmission baseband signal output from the first transmission baseband unit 205 and the pilot signal, and thereby performs a multiplier. Output to 281.

Multiplier 281 converts the signal quadrature modulated by quadrature modulator 280 into a radio signal, using the local oscillation signal output from third transmission local oscillation unit 223. The amplifier 282 amplifies the radio signal output from the multiplier 281 and transmits the amplified signal to the transmission antenna 231.

Next, signal transmission / reception operations in radio transmitting apparatus 100 in FIG. 5 and radio receiving apparatus 200 in FIG. 6 will be described with reference to FIG. FIG. 7 is a characteristic diagram showing frequency characteristics of each signal in the wireless transmission device 100 shown in FIG. 5 and the wireless reception device 200 shown in FIG. Is the frequency and the vertical axis is the signal level. 7 (A) to (G) show the frequency characteristics of the signal of the part to which the corresponding alphabet symbol is added in FIGS. 5 and 6. FIG.

[0063] The combined signal (A) of the modulation signal output from the modulation signal generation unit 103 and the pilot signal output from the pilot signal generation unit 104 has a frequency characteristic as shown in Fig. 7 (A). . As described above, here, the pilot signal is positioned at the center on the frequency axis of the modulation signal, and if the frequency of the pilot signal is f, f = 0

PILOT PILOT

[Hz].

[0064] The combined signal (A) of the modulated signal and the pilot signal is frequency-converted to a radio signal by transmission section 105 and output from antenna 130. Radio frequency f of the modulation signal included in the radio signal output from the antenna 130

RF and radio frequency of pilot signal f

RF—PILOT is expressed as the following equation.

f = f + f + f

RF CDMA Lol Lo2

f = f + f + f

RF_PIPOT PILOT Lol Lo2

Note that the frequency of the modulation signal generated by the modulation signal generation unit 103 is f, and the first transmission

CDMA

The frequency of the local oscillation signal oscillated by the local oscillation unit 121 is f and the second transmission local oscillation

LOL

Let f be the frequency of the local oscillation signal oscillated by unit 122.

Lo2

Here, in transmission section 105, the combined signal (A) is the phase noise of second transmission local oscillation section 122 in quadrature modulator 152 and the position of first transmission local oscillation section 121 in multiplier 153. Phase noise is superimposed and output as a radio signal. In addition, phase noise is superimposed on the radio signal even in the propagation path from the output from the antenna 130 of the radio transmission device 100 of FIG. 5 to the reception of the reception antenna 230 of the radio reception device 200 of FIG. Is done.

[0067] Therefore, assuming that the total sum of the phase noise superimposed on the transmitter 105 and the propagation path is Θ (t), the radio signal (B) received by the receiving antenna 230 is the frequency shown in FIG. It is expressed as the following equation with characteristics.

f Z Θ (t)

RF

f Z Θ (t)

RF—PILOT

[0068] The radio signal (B) received by the receiving antenna 230 in FIG. 6 was amplified by the amplifier 251. Later, the frequency is converted by a multiplier 252. Here, since the first reception local oscillation unit 221 oscillates a local signal having phase noise Φ (t), this local signal (C) has a frequency characteristic as shown in FIG. 7 (C). It is expressed as the following formula.

f Z φ (t)

Therefore, the phase noise φ (t) of the first reception local oscillation unit 221 is superimposed on the signal frequency-converted by the multiplier 252 and given to the band pass filter 253. The bandwidth of this bandpass filter 253 is the frequency of the difference component output from the multiplier 252, that is, (f

-f) and (f-f) are set to be extracted. Therefore, amplifier 25

The signal (D) output from 4 has the frequency characteristics shown in Fig. 7 (D), and is expressed by the following equation.

f f Z Θ (t)-(t)

f -f Z Θ (t)-(t)

[0070] Next, the signal (D) is distributed by the first distributor 255, one of which is output as a modulated signal branch, and the other is output as a pilot branch. In the pilot branch, the signal is distributed by the second distributor 258 and then input to the BPF 259. Since this BPF259 is set to extract only the pilot signal component, the BPF259 extracts and outputs only the pilot signal component from the distributed signal (D).

At this time, the delay τ is superimposed on the signal (D) by passing through the BPF 259 and the amplifier 260. Therefore, the output signal (Ε) of the amplifier 260 is expressed by the following equation with frequency characteristics as shown in FIG.

f f Z Θ (t- τ)-(t- τ)

On the other hand, in the modulation signal branch, the signal (D) is transferred to the delay corrector 256 by Μ = τ

1

A delay amount such that + τ is superimposed. Note that τ is a delay amount generated inside the quadrature demodulator 257 described later. Therefore, the signal (F) output from the delay corrector 256 has a frequency characteristic as shown in FIG. 7 (F) and can be expressed as the following equation.

f -f Θ Θ (t A t) — φ (t- A t)

[0073] Signal (E) and signal (F) are multiplied by orthogonal demodulator 257 and then orthogonally demodulated. Note that the quadrature demodulator 257 is delayed as shown in the internal configuration diagram of the quadrature demodulator 257 shown in FIG. A delay corrector 401, a 90-degree phase shifter 402, a delay corrector 401 -side multiplier 403, and a 90-degree phase shifter 402 -side multiplier 404 are provided.

In FIG. 8, the signal) is input to the delay corrector 401 and the 90-degree phase shifter 402. As a result, the 90-degree phase shifter 402 shifts the phase of the signal (E) by 90 degrees and outputs it to the multiplier 404. At this time, a delay amount τ is generated in the 90-degree phase shifter 402. Meanwhile, delay compensation

2

The device 401 has a delay amount τ in the signal (Ε) as much as the delay amount generated by the 90-degree phase shifter 402.

2 Make corrections so that they occur.

The signal (F) is input to the multiplier 403 and the multiplier 404, multiplied by the output signal (信号) from the delay corrector 401 and the 90-degree phase shifter 402, and the signal from the quadrature demodulator 257. Output as (G). Note that the signal) is corrected in the delay corrector 256 in consideration of the delay amount in the quadrature demodulator 257, so that the multiplier 403 and the multiplier 404 are corrected.

2

The signal (Ε) and signal (F) that are multiplied in step 1 are in phase. Therefore, ideal recovery can be performed.

Therefore, signal (G) output from quadrature demodulator 257 has a frequency characteristic as shown in FIG. 7 (G), and can be expressed by the following equation.

(f 1 f) 1 (f 1 f)

RF Lol RF— PILOT Lol

Z Θ (t- τ-τ)-(t- τ — τ) one {0 (t one A t) one φ (t one A t)}

1 2 1 2

[0077] When this is rearranged using the conditions f = 0 Hz and A t = τ + τ,

PILOT 1 2

It becomes like this.

f Z 0

CDMA

[0078] These equations show that the modulation signal generated by the modulation signal generation unit 103 is completely canceled out by the phase noise superimposed on the transmission unit 105, the propagation path, and the first reception local oscillation unit 221. Is demodulated by the radio receiving apparatus 200. That is, the phase noise superimposed on the received signal can be removed, and the phase noise generated in the system of the reception radio section can be removed.

[0079] As described above, radio transmitting apparatus 100 multiplexes and transmits a pilot signal at the center frequency of the transmission signal, and radio receiving apparatus 200 transmits the same frequency error and phase noise as the received signal. Frequency multiplication is performed with a pilot signal with For sound, frequency multiplication is performed using a signal having the same phase noise. Therefore, the frequency error and phase error contained in the received signal can be removed, and the phase error generated in the system can be completely removed, so that a wireless communication system with excellent phase noise characteristics can be obtained. Can be realized.

Next, the principle of AFC operation when a frequency shift is added to the transmission signal frequency by the propagation path will be described. In FIG. 6, the signal distributed by the second distributor 258 is input to the frequency shift amount detection unit 270 in the frequency shift detection branch. Then, the frequency shift amount detection unit 270 detects the frequency shift amount with respect to the reference frequency of the transmission base station, and outputs it to the frequency control unit 262. Accordingly, the frequency control unit 262 controls the output oscillation frequencies of the frequency variable reference signal reception oscillator 210 and the frequency variable reference signal reception oscillator 211 according to the input frequency deviation amount. At this time, the frequency of the first reception local oscillation unit 221 is referred to the oscillation frequency signal of the frequency variable reference signal reception oscillator 210 as a reference, so an arbitrary oscillation frequency signal corresponding to the frequency deviation is sent to the multiplier 252. Supply.

On the other hand, the transmission baseband signal output from the first transmission baseband unit 205 is modulated by the quadrature modulator 280, but the fourth reference signal is based on the oscillation frequency signal of the frequency variable reference signal transmission oscillator 211. Since the modulation oscillation signal is supplied from the transmission local oscillation unit 224, the output of the quadrature modulator 280 takes the frequency deviation into consideration. In addition, the force not shown in the figure is subjected to signal amplification processing by appropriate amplification and filtering, and then converted to a terminal output frequency by a multiplier 281.

At this time, since the oscillation signal for frequency conversion is supplied from the third transmission local oscillation unit 223 using the oscillation frequency signal of the frequency variable reference signal transmission oscillator 211 as a reference, a multiplier (frequency conversion device) 281 The output of is added with the amount of frequency deviation. The output signal of the multiplier (frequency conversion device) 281 is subjected to appropriate processing such as amplification and is output from the transmission antenna 231. As described above, since the output signal of the transmission antenna 231 is a signal in which the amount of frequency deviation is added, the frequency synchronization between the wireless transmission device 100 in FIG. 5 and the wireless reception device 200 in FIG. 6 can be achieved. .

[0083] As described above, according to the present embodiment, the pilot in the phase noise canceling technique is used. The output of the branch is further distributed to provide a frequency shift detection function. Then, the reference oscillation frequency in the wireless receiver is changed according to the detected frequency deviation amount, and the input frequency of the frequency conversion (multiplier) is set to a state that takes into account the frequency deviation amount. Achieves frequency tracking. This makes it possible to achieve both phase noise cancellation and AFC.

[0084] When the pilot signal is extracted by BPF259, the phase noise outside the frequency band of BPF259 cannot be extracted. Therefore, the phase noise is included in the frequency characteristics shown in FIG. 6 (D). However, this phase noise can be suppressed by the second transmission local oscillation unit 122 of the wireless transmission device 100 shown in FIG. 5 and the first reception local oscillation unit 221 of the wireless reception device 200 shown in FIG.

For example, the first reception local oscillation unit 221 is configured as a PLL frequency synthesizer, and the loop bandwidth is designed to be equal to or less than the bandwidth of the BPF259. By doing so, the phase noise φ (t) outside the pass frequency band of the bandpass filter shown in Fig. 2 (D) can be suppressed, and the effect can be ignored. Note that the second transmission local oscillation unit 122 and the first transmission local oscillation unit 121 of the wireless transmission device 100 shown in FIG. 5 similarly suppress the phase noise 0 (t) outside the frequency band of the BPF259. can do.

Further, in the present embodiment, first transmission local oscillation unit 121 in radio transmission device 100 oscillates as a local frequency oscillated in first reception local oscillation unit 221 of radio reception device 200 in FIG. Signal with the same frequency (f) as the local oscillation signal

LOL

Force 2 XRF frequency (f + f) or less and different from RF frequency

Lol Lo2

Naturally, even if the same frequency (f) as that of the local oscillation signal oscillated by the second transmission local oscillation unit 122 is used, the phase noise can be similarly canceled.

Lo2

Further, in the present embodiment, the configuration of transmission baseband section 101 and transmission section 105 in radio transmission apparatus 100 in FIG. 5 is the superheterodyne system, but the pilot signal is centered on the frequency axis of the modulation signal. Any method can be used as long as it can transmit a signal having a frequency characteristic in which signals are arranged. For example, direct conversion, low IF, or the like may be used.

[0088] Also, according to the present embodiment, the radio signal received by the radio reception device is the center frequency. Therefore, the local oscillator 4 and the frequency change 8 in the signal branch of the conventional local noise canceller in FIG. 1 are multiplexed with a modulation signal that does not include signals in the number and a pilot signal having the same center frequency as the center frequency. Therefore, the phase noise included in the local oscillation signal generated by this local oscillation unit 4 is not included in the signal branch signal (signal F). Therefore, the phase error generated in the system can be completely removed, and a radio communication system having excellent phase noise characteristics can be realized. Furthermore, it becomes possible to realize AFC that performs frequency synchronization between the wireless transmission device and the wireless reception device.

FIG. 9 is a block diagram showing a configuration of a radio reception apparatus of the radio communication system according to Embodiment 2 of the present invention. The radio reception device 200a in FIG. 9 shows a single conversion type in which a received signal is frequency-converted by a multiplier 252. Note that the configuration of the wireless transmission apparatus of the wireless communication system according to Embodiment 2 is the same as the configuration of FIG.

[0090] The radio receiving apparatus 200a shown in FIG. 9 differs from the radio receiving apparatus 200 shown in FIG. 6 in that a second orthogonal demodulator 261 is added except for the frequency shift amount detection unit 270, and The difference is that a frequency shift amount detection unit 206 is added to the reception baseband unit 203a. The second orthogonal demodulator 261 uses the signal from the frequency variable reference signal reception oscillator 210 to orthogonally demodulate one of the signals distributed by the second distributor 258. The frequency shift amount detection unit 206 also detects the frequency shift amount Δ ί for the signal power output from the second quadrature demodulator 261. That is, in radio receiving apparatus 200a of the second embodiment shown in FIG. 9, second orthogonal demodulator 261 is provided as a method for realizing frequency deviation amount detection section 270 of the first embodiment shown in FIG. Is input to the frequency shift amount detection unit 206 in the first reception baseband unit 203a to detect the frequency shift amount.

[0091] Thus, according to the present embodiment, as in the case of Embodiment 1, it is possible to achieve low power consumption and improved reception sensitivity while making both the phase noise canceller and the AFC compatible. it can.

<Embodiment 3> FIG. 10 is a block diagram showing a configuration of a radio reception apparatus of the radio communication system according to Embodiment 3 of the present invention. The radio reception device 200b of FIG. 10 shows a single conversion type in which a received signal is frequency-converted by a multiplier 252. Note that the configuration of the wireless transmission apparatus of the wireless communication system according to Embodiment 3 is the same as the configuration of FIG.

The wireless receiving apparatus 200b shown in FIG. 10 is different from the wireless receiving apparatus 200a shown in FIG. 9 in that a frequency variable BPF263 is provided instead of the BPF259. This frequency variable B PF263 varies the band frequency to be passed according to the frequency deviation amount Δί. In other words, the filter that extracts the pilot signal of the pilot branch is the frequency variable BPF263 that makes the center frequency variable, and the pilot signal can be extracted according to the frequency deviation amount, thereby reducing the phase noise cancellation effect due to the frequency deviation. Suppress it.

That is, in radio receiving apparatus 200b shown in FIG. 10, frequency variable BPF 263 is a band to be passed in accordance with frequency deviation amount Δ ί input from frequency deviation amount detection unit 206 of first reception baseband unit 203a. The frequency is varied. As a result, even if a frequency shift occurs, the frequency variable BPF 263 can change the band frequency according to the frequency shift amount Δ ί, so there is no possibility that the pilot signal extracted by the frequency variable BPF 263 will deteriorate. . Therefore, the phase noise canceller can be reliably operated even if the frequency shift amount Δ ί fluctuates.

FIG. 11 is a block diagram showing a configuration of a radio reception apparatus of the radio communication system according to Embodiment 4 of the present invention. The radio receiving apparatus 200c in FIG. 11 shows a direct conversion type in which the received signal is directly converted into a baseband signal without frequency conversion. Note that the configuration of the wireless transmission apparatus of the wireless communication system according to Embodiment 4 is the same as the configuration of FIG. The wireless reception device 200c shown in FIG. 11 is that the wireless reception device 200a shown in FIG. 9 is changed from single conversion to direct conversion. That is, as in the radio reception apparatus 200c shown in FIG. 11, the phase noise canceller and the AFC can be compatible with each other even if the input signal is not frequency-converted and the direct conversion type is used. <Embodiment 5>

FIG. 12 is a block diagram showing a configuration of a radio reception apparatus of the radio communication system according to Embodiment 5 of the present invention. The radio reception device 200d in FIG. 12 shows a direct conversion type in which the received signal is directly converted into a baseband signal without frequency conversion. Note that the configuration of the radio transmission apparatus of the radio communication system according to Embodiment 5 is the same as the configuration of FIG. 5 shown in Embodiment 1, and thus the description thereof is omitted. The wireless reception device 200d shown in FIG. 12 is that the wireless reception device 200b shown in FIG. 10 is changed from the single conversion to the direct conversion.

That is, as in the radio reception device 200d shown in FIG. 12, even if it is a direct conversion type in which the input signal is not frequency-converted, it is possible to achieve both the phase noise canceller and the AFC as well as the frequency shift. Even if this occurs, the extracted pilot signal will not be degraded. Therefore, the phase noise canceller can be operated reliably even if a frequency shift occurs.

Note that the wireless communication system of the present invention can be realized by a Low-IF configuration in addition to direct conversion and single conversion. In addition to the AFC configuration, the wireless communication system of the present invention can also add a high-performance function of phase noise canceller technology. For example, it is possible to perform variable gain amplification that calculates the received power value of the received signal based on the amplitude of the output signal of the quadrature demodulator and performs amplification in accordance with the received power value. In addition, a temperature measuring means for measuring the temperature can be provided, and a function for correcting the delay amount change due to the temperature characteristic and the amplitude of the received signal can be added.

Furthermore, the radio communication system of the present invention can also realize an orthogonal demodulator by frequency-multiplying the signal component corresponding to the pilot signal extracted by the BPF and the output signal of the delay corrector. It is also possible to change the BPF bandwidth in the pilot branch based on the filter band control signal in the baseband part. It is also possible to provide amplification means for amplifying the signal distributed in the pilot branch and outputting it to the BPF. In addition, the BSF for suppressing pilot signal components in the signal branch is omitted.

Industrial applicability The wireless receiver and the wireless communication system according to the present invention can improve the phase noise characteristics while performing optimum automatic frequency control (AFC), and thus various wireless communication devices such as a mobile phone, a PHS, and a wireless LAN. In addition, the power can be effectively used for the wireless communication system configured.

Claims

The scope of the claims

[1] In the wireless transmission device of the communication partner, a receiving means for receiving the transmitted signal with the notched signal superimposed thereon;

Extracting means for extracting the pilot signal from the received signal;

An orthogonal demodulation means for performing orthogonal demodulation on the received signal using the extracted pilot signal;

A frequency deviation detecting means for detecting a frequency deviation of the oscillation frequency of the local oscillator with respect to a reference frequency using the received signal;

Automatic frequency control means for performing control to synchronize the oscillation frequency of the local oscillator with the reference frequency using the detected frequency deviation amount;

A wireless receiver comprising:

[2] comprising second orthogonal demodulation means for performing orthogonal demodulation on the received signal using a signal oscillated from the local oscillator;

The frequency shift amount detecting means detects the frequency shift amount from a signal quadrature demodulated by the second orthogonal demodulating means;

The wireless receiver according to claim 1.

[3] The extraction means includes a frequency variable bandpass filter that extracts a signal component corresponding to a pilot signal by band-limiting the received signal, and the frequency variable bandpass filter based on the frequency shift amount Change the center frequency of

The wireless receiver according to claim 1.

[4] A wireless communication system comprising a wireless transmission device and a wireless reception device,

The wireless transmission device

A transmission means for transmitting a radio signal in which a pilot signal is superimposed at the center of a multicarrier signal;

The wireless receiver is

Receiving means for receiving a signal transmitted from the wireless transmission device;

Extracting means for extracting the pilot signal from the received signal;

A direct demodulation is performed on the received signal using the extracted pilot signal. A demodulating means;

Automatic frequency control means for performing control to synchronize the oscillation frequency of the local oscillator with the reference frequency using the detected frequency deviation amount,

Wireless communication system.