JP2007295331A - Radio base station device - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a radio base station device which can be made inexpensive and compact and is capable of reducing computational complexity of a predistorter and quickly and precisely compensating a nonlinear distortion by resolving conventional issues that a circuit for receiving a feedback signal is large-scale to increase the device cost to hinder miniaturization of the device and that the computational complexity of the predistorter is large to require much time for processing. <P>SOLUTION: In the radio base station device, a transmission signal emitted from an antenna of system 0 (Ant0) for transmission/reception is received as a feedback signal by an antenna of system 1 (Ant1) only for reception, and a frequency multiplexed signal of an original reception signal and the feedback signal is subjected to A/D conversion and is subjected to quadrature detection by a digital quadrature detection part 46 and is separated into the reception signal and the feedback signal by time division FFT processing, and a distortion compensation part 53 updates a distortion compensation table 53 on the basis of feedback data. The device does not require a reception circuit only for feedback. <P>COPYRIGHT: (C)2008,JPO&INPIT

Description

  The present invention relates to a radio base station apparatus for mobile communication, and more particularly to a radio base station apparatus capable of simplifying the apparatus configuration and compensating for nonlinear distortion of a power amplifier at high speed and with high accuracy.

As a nonlinear distortion compensation method for a power amplifier in a conventional radio base station apparatus, there are a feedback distortion compensation system, a feedforward distortion compensation system, and a PD (Pre-Distortion) system.
The feedback method is not suitable for use in a radio base station because instability increases when used in a wide band like W-CDMA (Wideband Code Division Multiple Access).
The feed-forward method is configured to extract an error component in a transmission signal, amplify it with a sub-amplifier, and subtract it from the output signal of the main amplifier. Therefore, the circuit becomes complicated and power efficiency is improved by using the sub-amplifier. There is a problem of lowering.

  Recently, a PD method that does not require a sub-amplifier is used in order to give in advance a reverse characteristic of distortion generated in the main amplifier, and in particular, a DPD (Digital Pre-Distortion) method is often used. In the DPD method, it is difficult to generate a distortion compensation value having an accurate inverse characteristic compared to a feedback method or a feedforward method using the distortion signal itself of the main amplifier, but with a simple configuration and a relatively high accuracy. A high antiphase compensation value can be generated.

  In the DPD method, a part of the transmission output signal is received as a feedback signal, the predistorter detects the difference from the transmission signal, performs the process of optimizing the value of the reference table used for distortion compensation, and converges. The reference table is updated with the obtained values and used for the next distortion compensation.

Separately, in digital wireless communication systems, the reception diversity method is used as a method for improving the error rate of communication contents and improving the quality of communication lines, and is used for wireless base station devices and terminal devices. Has been.
In the reception diversity method, a plurality of receivers are prepared, signals transmitted through different transmission paths are received, and the quality of the received signal is improved by selection or maximum ratio combining.

As a conventional technique using the reception diversity system, Japanese Patent Laid-Open No. 11-239077 “Digital Radio” published on August 31, 1999 (Applicant: Hitachi Electronics Co., Ltd., Inventor: Kazue Kato) There is.
In this prior art, the reception path includes a switch for switching the first and second frequency bands to a reception-dedicated antenna that is not connected to the antenna duplexer among the two diversity receivers, and communicates via the base station. The control unit receives the signal received through the base station with diversity, and receives the signal through the direct communication between the terminals. The signal is controlled to receive non-diversity, and the transmission path is not provided with a changeover switch so that nonlinear distortion caused by the changeover circuit can be removed (see Patent Document 1).

JP-A-11-239077

  However, the conventional DPD method requires a separate feedback circuit for receiving a wideband transmission signal including distortion, increasing the device cost and hindering downsizing of the device.

  Further, in the conventional DPD method, the predistorter needs to converge the reference table value while repeating fine adjustment so that the transmission signal approaches asymptotically asymptotically, and the processing amount of the convergence process of the reference table is large. There was a problem that processing took time.

  In particular, when a feedback system is provided, new distortions and frequency-dependent memory effects on the AM-AM characteristics and AM-PM characteristics occur, and the amount of processing increases to converge them, and the processing time increases. There was a problem that it took.

  In addition, when amplifiers have frequency characteristics, many parameters and sufficient averaging time are required, and they cannot be processed at high speed. There was a problem that characteristics could not be obtained.

  The present invention has been made in view of the above circumstances, and provides a radio base station apparatus that can reduce the size of the apparatus, reduce the processing amount of the predistorter, and compensate for nonlinear distortion at high speed and with high accuracy. The purpose is to do.

  The present invention for solving the problems of the above-described conventional example multiplies the transmission signal before amplification by an amplifier that amplifies the transmission signal and a distortion compensation coefficient that compensates for nonlinear distortion generated in the amplifier, and after amplification. A predistorter that inputs a feedback signal of a transmission signal, detects distortion included in the feedback signal by fast Fourier transform, and updates the distortion compensation coefficient so that the distortion is minimized, and a radio signal with a radio terminal A radio base station apparatus that transmits and receives a delay signal that delays a transmission signal before amplification multiplied by a distortion compensation coefficient for a predetermined time, and a predistorter that is delayed by a feedback signal and a delay circuit It is characterized by performing a fast Fourier transform on the difference between the two.

  The present invention also includes a modulation unit that digitally modulates transmission data, an amplifier that amplifies the modulated transmission signal, a first antenna dedicated for transmission / reception or transmission, a second antenna dedicated for reception, and an amplifier. A predistorter that multiplies the transmission signal before amplification by a distortion compensation coefficient that compensates for the generated nonlinear distortion, inputs a feedback signal of the transmission signal after amplification, and updates the distortion compensation coefficient based on the feedback signal. A radio base station apparatus for transmitting and receiving radio signals to and from a radio terminal, wherein the second antenna receives a radio signal from the radio terminal and transmits a transmission signal output from the first antenna as a feedback signal The received signal from the wireless terminal received by the second antenna and the feedback signal are the same signal as a frequency multiplexed signal. An A / D converter that samples at the pulling frequency, a digital quadrature detector that digitally detects the frequency multiplexed signal digitally converted by the A / D converter, and fast Fourier transforms the digital quadrature detected signal in a time-division manner. A fast Fourier transform unit that separates a received signal from a wireless terminal and a feedback signal, an inverse fast Fourier transform unit that performs inverse fast Fourier transform on both separated signals in a time-division manner, and both signals that have been subjected to inverse fast Fourier transform Each of which is digitally demodulated, and includes two demodulator units that output received data and feedback data from the wireless terminal, and the predistorter includes data of a known pilot channel included in the transmission signal before digital modulation, A predistorter that updates the distortion compensation coefficient based on the difference from the feedback data It is.

  According to the present invention, the delay circuit that delays the transmission signal before amplification multiplied by the distortion compensation coefficient for a predetermined time, and the predistorter calculates the difference between the feedback signal and the signal delayed by the delay circuit as a fast Fourier transform. Since it is a radio base station device that detects the distortion included in the feedback signal by converting it, the amount of data subjected to fast Fourier transform can be greatly reduced to shorten the processing time, and the distortion compensation coefficient in the predistorter It is possible to speed up the process of updating the.

  According to the present invention, the second antenna receives a radio signal from the radio terminal, receives a transmission signal output from the first antenna as a feedback signal, and the A / D converter The received signal from the wireless terminal received by the antenna 2 and the feedback signal are sampled at the same sampling frequency as a frequency multiplex signal, and the digital quadrature detection unit digitally quadrature-detects the digitally converted frequency multiplex signal and performs fast Fourier transform. The conversion unit performs fast Fourier transform on the digital quadrature detected signal in a time division manner to separate the received signal from the wireless terminal and the feedback signal, and the inverse fast Fourier transform unit performs time division on both the separated signals. The inverse fast Fourier transform is performed, and the demodulator digitally demodulates both the signals subjected to the inverse fast Fourier transform and receives them from the wireless terminal. And a predistorter that updates the distortion compensation coefficient based on the difference between the known pilot channel data included in the transmission signal before digital modulation and the feedback data. Since it is a device, all the received signals and feedback signals from the terminals received by the same antenna are processed by a common circuit, so that no dedicated receiving circuit for feedback signals is required, and the circuit configuration is greatly simplified. The apparatus cost can be reduced and the apparatus can be reduced in size, and a signal having a high S / N ratio emitted at a close distance is used as a feedback signal, and a distortion compensation coefficient based on a difference from information of a known pilot channel By performing the optimization process, the distortion compensation coefficient optimization process in the predistorter is highly accurate. One there is an effect that can be fast to be carried out.

Embodiments of the present invention will be described with reference to the drawings.
The radio base station apparatus according to the embodiment of the present invention provides feedback from a distortion feedback transmission feedback signal after A / D conversion before the FFT (Fast Fourier Transform) unit that detects distortion components. A subtractor that subtracts the transmission signal delayed by the time is provided, and the reference table is updated by performing FFT on the difference between the transmission feedback signal and the transmission signal, reducing the amount of calculation in the FFT unit and speeding up the processing. It can be made.

  In addition, the radio base station apparatus according to the embodiment of the present invention receives a transmission feedback signal for distortion compensation at a reception-dedicated antenna among the two antennas of the diversity receiver, thereby providing a circuit dedicated for feedback. It is unnecessary and the apparatus can be simplified.

FIG. 1 is a configuration block diagram of a radio base station apparatus (first apparatus) according to the first embodiment of the present invention. Note that FIG. 1 shows the transmission system and feedback system of the first apparatus, and the reception system has a general configuration and operation, and therefore illustration and description thereof are omitted.
As shown in FIG. 1, the first apparatus includes a modulation unit 11, a digital quadrature modulation unit 12, a multiplier 13, a D / A (Digital-Analog) converter 14, a band pass filter BPF; Filter) 15, mixer 16, main amplifier (PA) 17, directional coupler 18, band pass filter (BPF) 19, antenna duplexer (Duplex) 20, and band pass filter (BPF) ) 21, mixer 22, frequency generation unit 23, band pass filter (BPF) 24, A / D (Analog-Digital) converter 25, delay unit 26, adder 27, and FFT unit 28 , An averaging unit 29, an LUT (LUT; Look-Up Table) convergence calculation unit 30, and a distortion compensation table 31.

Each component will be briefly described.
In the transmission system, the modulation unit 11 performs spreading processing, gain setting, and hybrid phase shift keying on the input data to generate, synthesize, and output the physical channel signals.
The digital quadrature modulation unit 12 upsamples the modulated data and performs digital quadrature modulation using a Hilbert filter or the like.
The multiplier 13 performs complex multiplication of the orthogonally modulated transmission data and the distortion compensation coefficient from the distortion compensation table 31.

The D / A converter 14 converts the digital signal into an analog signal.
The band pass filter 15 performs band limitation for extracting a desired IF frequency band.
The mixer 16 multiplies the IF transmission signal by the input frequency from the frequency generation unit 23 and up-converts the IF transmission signal to a radio frequency.

The main amplifier 17 amplifies the transmission signal with a predetermined amplification factor, and nonlinear distortion occurs during amplification.
The directional coupler 18 branches and extracts a feedback signal from the amplified transmission signal.
The band pass filter 19 limits the transmission signal to a desired frequency.
The antenna duplexer 20 is for using the antenna for both transmission and reception, and outputs a transmission signal to the antenna and outputs a reception signal from the antenna to the reception system.
Since the reception system has a general configuration for demodulating a reception signal and outputting reception data, description thereof is omitted here.

In the feedback system, the band-pass filter 19 band-limits the feedback signal branched by the directional coupler 18.
The mixer 22 multiplies the feedback signal by the frequency from the frequency generator 23 and down-converts it to the IF frequency.

The frequency generator 23 oscillates a specific frequency to be output to the mixer 16 and the mixer 22.
The band pass filter 24 further limits the band of the feedback signal of the IF frequency.
The A / D converter 24 converts an analog signal into a digital signal.

The delay unit 26 delays the transmission signal before D / A conversion by a feedback time.
The adder 27 subtracts the delayed transmission signal from the feedback signal from the A / D converter 25.
The FFT unit 28 performs fast Fourier transform to detect out-of-band components.
The averaging unit 29 integrates the power value of the out-of-band component for a certain time and averages it.

The distortion compensation table 31 stores a distortion compensation coefficient.
The LUT convergence calculation unit 30 optimizes the distortion compensation value of the distortion compensation table 31 so as to minimize the averaged power value, and updates the distortion compensation table 31 with the convergence value.

The operation of the transmission system and the feedback system in the first device having the above configuration will be described.
In the transmission system of the first device, transmission data is digitally modulated by the modulation unit 11 and orthogonally modulated by the digital quadrature modulation unit 12, and then the I and Q signals are respectively distorted from the distortion compensation table 31 by the multiplier 13. The complex multiplication with the compensation coefficient is performed to compensate for distortion, and the analog signal is converted by the D / A converter 14.
A part of the transmission data subjected to distortion compensation is input to the delay unit 26 and delayed.

  The D / A converted transmission signal is band-limited by the BPF 15, up-converted to a radio frequency by the mixer 16, and amplified by the main amplifier 17 at a predetermined amplification factor. Then, the band is limited by the BPF 19 via the directional coupler 18, and is output from the antenna duplexer 20 to the antenna as a transmission signal and transmitted.

  On the other hand, in the feedback system, the feedback signal branched by the directional coupler 18 is band-limited by the BPF 21, down-converted to the IF frequency by the mixer 22, band-limited by the BPF 24, and digital signal by the A / D converter 25. Is converted to

  The adder 27 extracts a difference by subtracting the transmission signal delayed by the feedback time in the delay unit 26 from the feedback signal converted into a digital signal, and the FFT unit 28 reduces the distortion caused by the amplification. The spectrum of the resulting difference (out-of-band component) is output.

  In this way, the first device does not perform FFT on the feedback signal itself, but performs FFT after taking the difference from the transmission signal. The amount of calculation can be greatly reduced, and the processing speed can be increased.

  Then, the power value of the out-of-band component extracted by the FFT unit 28 is averaged by the averaging unit 29, and the distortion compensation value is optimized by the LUT convergence calculation unit 30 so that the power value of the out-of-band component is minimized. The distortion compensation table 31 is updated.

  According to the radio base station apparatus according to the first embodiment of the present invention, the delay unit 26 that delays the transmission signal before amplification by the feedback time is provided, and the adder 27 performs feedback from the feedback signal of the transmission signal. The transmission unit delayed by the time is subtracted, the FFT unit 28 performs FFT on the difference, and the LUT convergence calculation unit 30 updates the distortion compensation table 31 so that the power value of the out-of-band component is minimized. As a result, the amount of data input to the FFT unit 28 can be reduced to greatly reduce the amount of computation of the FFT, and the speed of processing for updating the distortion compensation table 31 can be increased.

Next, a radio base station apparatus (second apparatus) according to the second embodiment of the present invention is described.
The second device is a diversity-structured reception unit having two reception systems, and receives its own transmission signal transmitted from the antenna connected to the antenna duplexer with another antenna dedicated to reception, This is used as a feedback signal. In particular, a signal received from a terminal and a feedback signal are A / D converted by the same A / D converter as a frequency multiplexed signal. As a result, the circuit configuration can be simplified, and a feedback signal with a high S / N ratio emitted at a close distance can be obtained. Based on this, the distortion compensation coefficient can be optimized and highly accurate distortion compensation can be performed. It is obtained.

The configuration of the second device will be described with reference to FIG. FIG. 2 is a configuration block diagram of the second device. The same parts as those in FIG. 1 will be described with the same reference numerals.
As shown in FIG. 2, the second device includes two antennas, and includes a 0-system antenna (Ant0) connected to the antenna duplexer 20 and a reception-only 1-system antenna (Ant1). The transmission diversity is performed by the 0-system antenna, and the reception is performed by both the 0-system and the 1-system.

The 0-system transmission system of the second device includes a modulation unit 11, a digital quadrature modulation unit 12, a D / A converter 14, a BPF 15, a frequency oscillator 56, a mixer 16, a main amplifier 17, It is composed of a bandpass filter 19, an antenna duplexer (Duplex) 20, and a 0-system antenna (Ant0). The 1-system dedicated to reception is a 1-system antenna (Ant1) and an LNA (Low Noise Amplifier). ) 40, BPF 41, mixer 42, BPF 44, A / D converter 45, digital quadrature detection unit 46, FFT unit 47, frequency transition unit 48, LPF (Low Pass Filter) 49, IFFT The unit 50, the demodulation unit 51, the distortion compensation correction unit 52, the distortion compensation table correlation value detection circuit 53, and a plurality of frequency oscillators 54 and 55.
Since the 0-system receiving system has a general configuration and operation, description thereof is omitted.

The 0-system transmission system is almost the same as the transmission system in the first apparatus shown in FIG. 1, and thus the description thereof will be omitted here. However, the modulation unit 11 refers to the distortion compensation table 53 before the digital quadrature modulation. Thus, the first device differs from the first device in that phase compensation and level compensation are performed.
In the second apparatus, a bit string preset in a pilot channel (CPICH in W-CDMA) is explicitly transmitted for updating the distortion compensation table.

The configuration of system 1 will be specifically described.
The 1-system antenna receives not only a normal reception signal from the terminal but also a signal transmitted from its own 0-system antenna as a feedback signal.
The LNA 40 amplifies the received signal.
The BPFs 41 and 44 limit the band of the input signal. Here, a filter that passes the reception frequency (1950 MHz) from the terminal and the band of its own transmission frequency (2014 MHz) is provided.
The mixer 42 multiplies the received signal by the frequency from the frequency oscillator 54 and down-converts it to the IF frequency.

The A / D converter 45 undersamples the received signal based on a clock from the frequency oscillator 55 (for example, 61.44 MHz which is 16 times the chip rate).
The digital quadrature detection unit 46 performs digital quadrature detection on the received digital signal at a predetermined frequency. As a feature of the second device, a received signal from a terminal and a feedback signal are simultaneously subjected to quadrature detection by a single digital quadrature detection unit by frequency multiplexing. In this example, it is assumed that the sample rate is halved with digital quadrature detection.

Similar to the first device, the FFT unit 47 performs an FFT operation, and also has a function as a BPF that separates a received signal and a feedback signal.
The frequency transition unit 48 performs frequency transition processing on the feedback signal.
The LPF 49 removes high frequency components by multiplying each frequency component by a filter coefficient.
The IFFT unit 50 performs an inverse FFT operation.
The demodulator 51 despreads and digitally demodulates the IFFT converted signal.

  As described above, in the second device, the configuration of the reception circuit of the 1-system antenna is all common to the reception signal and the feedback signal, and the device configuration can be greatly simplified. If the feedback is made with a simple directional coupler in order to simplify the device, the reproducibility of the coupling amount, the frequency characteristic, etc. is reduced, but the second device receives the signal via the regular route from the transmission system to the reception system. Therefore, no additional parts are required, and a stable feedback signal can be obtained. However, this does not prevent the feedback signal from being coupled to the reception system using a directional coupler as in the first device.

  The distortion compensation correction unit 52 compares the data demodulated by the demodulation unit 51 with the transmitted pre-modulation data of the pilot channel, compares the amplitude and phase, and updates the distortion compensation value of the distortion compensation table 53. It is. The distortion compensation table 53 stores distortion compensation values in association with transmission power. The distortion compensation correction unit 52 and the distortion compensation table 53 constitute a predistorter. Here, although the predistorter is configured to include a distortion compensation table, the distortion compensation correction unit 52 calculates a distortion compensation control value without providing the table, and directly transmits the control value to the modulation unit 11. You may make it output.

Next, adjustment of the reception level of the reception signal and the feedback signal will be described.
In the first system, since the reception signal from the terminal and the feedback signal are multiplexed and processed, it is necessary to make the reception levels of both signals the same level. Compared with the received signal from the terminal, the feedback signal is a signal emitted at a very close position, so that the reception level at the 1-system antenna becomes very high. Therefore, the second device is provided with a configuration for weakening the reception level of the feedback signal.

  As a configuration for weakening the reception level of the feedback signal, a new filter (not shown) may be provided in front of the LNA 40 to suppress the reception level of the feedback signal. FIG. 3 is a schematic explanatory diagram illustrating the frequency characteristics of a filter connected to the system 1 antenna of the second device.

  Since the frequency received from the terminal is different from the frequency of the feedback signal (that is, its own transmission frequency), as shown in FIG. 3, the filter connected to the system 1 antenna does not attenuate much the original received signal band. And the transmission signal band is greatly attenuated to reduce the level, whereby the levels of both the received signal from the terminal and the feedback signal can be made the same level.

As another configuration, it is also conceivable to provide a reflector for appropriately preventing the wraparound from the 0-system antenna to the 1-system antenna. FIG. 4 is an explanatory diagram when a reflector is provided on the antenna.
As shown in FIG. 4, in another configuration, a reflector is provided between the 0-system antenna and the 1-system antenna to make it difficult for radio waves to arrive directly from the 0-system antenna to the 1-system antenna. The size of the reflection plate is adjusted so that the reception level of the transmission signal band at becomes a desired level.

  Furthermore, the level of both signals can be finely adjusted by digital AGC (Auto Gain Control) by the FFT unit 47 and the IFFT unit 50, and the levels of both signals can be made comparable. In addition, you may use combining a filter and a reflecting plate.

Next, a method for receiving a frequency multiplexed signal in the first system of the second device will be described in detail.
In the example of FIG. 2, IF band undersampling is applied to the W-CDMA system based on the specification of 3GPP (3rd generation mobile communication system standard).
In the 2 GHz band frequency of 3GPP, the center frequency of the transmission frequency and the center frequency of the reception frequency are 190 MHz apart. For example, if the transmission frequency is 2140 MHz, the reception frequency is 1950 MHz.

The system 1 antenna receives a 1950 MHz reception signal (described as “Rx” in the figure) and a 2140 MHz transmission signal (“Tx” in the figure) as a feedback signal, and the BPF 41 receives a frequency multiplexed signal including the frequency band. Become.
Then, if the IF signal is obtained by down-conversion by the mixer 42 using the frequency from the local oscillator, and the A / D converter 45 performs undersampling at 61.44 MHz to convert it to a digital signal, the center frequency of the received signal is 7. The center frequency of 68 MHz and the feedback signal (“DPD” in the figure) is 13.36 MHz, and there is a deviation of 5.68 MHz.

When the digital quadrature detection unit 46 performs quadrature detection of the frequency multiplexed signal at the center frequency of the received signal of 7.68 MHz, the center frequency of the received signal becomes 0 MHz, but the center frequency of the feedback signal is shifted by 7.68 MHz. There is still a 5.68 MHz shift.
Therefore, in the second device, this shift is corrected by the frequency transition unit 48 subsequent to the FFT unit 47.

  For example, when a signal after quadrature detection is handled as 8 times oversampling of a W-CDMA spreading rate of 3.84 MHz (3.84 × 8 = 30.72 MHz), if a 4096-point FFT is used, an interval in the frequency domain Becomes 7.5 kHz. Even if the band of the feedback signal is shifted by 757 points (corresponding to 5.6775 MHz), a deviation of 2.5 kHz still remains. This shift is absorbed by performing a correction operation using the adjacent frequency. The correction may be a linear calculation or a higher-order correction calculation.

In the second apparatus, the digital quadrature detection unit 46 performs processing without separating the reception signal and the feedback signal, but the FFT unit 47 and subsequent units perform processing for each of the reception signal and the feedback signal by time division processing. It is like that. In FIG. 2, the lower arrow indicates the received signal, and the upper arrow indicates the feedback signal.
Alternatively, as shown by the dotted line in FIG. 2, if a second digital quadrature detection unit 46 ′ that performs quadrature detection centering around 13.36 MHz is additionally provided, the FFT unit 47 is also used in time division, but the frequency transition unit 48 Can be made unnecessary.

  Since the FFT unit 47 performs conversion from the time domain to the frequency domain, the output of the FFT unit 47 is separated into a received signal and a feedback signal. In each processing part in the subsequent stage, each signal is time-divided. Processing is performed. At that time, since FFT is an operation based on many samples, suppression of signals outside the band such as quantization noise can be expected. Further, since the dynamic range is expanded by increasing the effective digits, AGC (Auto Gain Control) can be performed during the FFT processing as in Japanese Patent Application No. 2006-010296.

Then, the LPF 49 performs appropriate root Nyquist filtering or the like for each received signal and feedback signal in the frequency domain, and removes high frequency components. At that time, the frequency characteristics of the reception system and digital quadrature detection may be corrected simultaneously. It is also possible to perform despreading in the frequency domain by multiplying the Fourier transform of the spreading code (long code). Alternatively, decimation can be performed at the same time by deleting points on the high-frequency side that are no longer necessary, reducing the number of points, and performing IFFT. By performing FFT processing in this way, various signal processing can be performed intensively, and as a result, the amount of calculation can be reduced.
The feedback signal is digitally demodulated by the demodulator 51a and the received signal is demodulated by the demodulator 51b, and the received data is extracted from the received signal and output to another circuit. The demodulated data of the feedback signal is output to the distortion compensation correction unit 52.

  Then, the distortion compensation correction unit 52 optimizes the distortion correction coefficient based on the pilot channel (CPICH) data before modulation of the transmission signal and the demodulated data of CPICH of the feedback signal input from the demodulation unit 51. The distortion compensation table 53 is updated and used for the next distortion compensation.

Next, the relationship between the IF band frequency, the sampling clock of the A / D converter 45 and the clock frequency of the digital quadrature detection 46 in the second device will be described.
In the second apparatus, both the received signal and the feedback signal are detected by undersampling in the A / D converter 45. Therefore, the sampling frequency and the detection frequency must be selected so that the IF band and aliasing band of each signal do not overlap. Don't be. In addition, as described above, in 3GPP, the center frequency difference between transmission and reception is fixed at 190 MHz, and it is necessary to secure 5 MHz (center frequency ± 2.5 MHz) for each frequency band.

The IF frequency is determined so as to satisfy the conditions regarding these frequencies.
An example of IF frequency and aliasing will be described with reference to FIG. FIG. 5 is an explanatory diagram showing the relationship (example 1) between the IF frequency and aliasing.
As shown in FIG. 5, there are aliasing that appears at sampling frequency intervals (a frequency that is an integral multiple of the sampling frequency) and that that appears after detection frequency separation before and after a frequency that is an integral multiple of the sampling frequency. One of these is selected as the IF band.

  For example, when the digital quadrature detection unit 46 extracts the received signal at 7.68 MHz, if the sampling clock of the A / D converter 45 is 61.44 MHz, the IF band is 69.12 MHz, 130.56 MHz, 192.00 MHz. It will be either ...

  If the IF band of the reception baseband signal is 253.44 MHz, the IF band frequency of the transmission baseband signal is 443.44 MHz, which is a difference of 190 MHz, and as shown in FIG. Therefore, the 5 MHz band can be secured.

  That is, when the sampling frequency is 61.44 MHz and digital quadrature detection is performed at 7.68 MHz, each combination of the reception IF band and the corresponding transmission IF band (the difference between the transmission and reception frequencies is 190 MHz) can be selected. This setting is appropriate because it satisfies the condition that the frequency band is 5 MHz and no aliasing overlaps with the selected IF band.

Next, another example will be described with reference to FIG. FIG. 6 is an explanatory diagram showing a relationship (example 2) between the IF frequency and aliasing.
When the digital quadrature detection unit 46 detects the received signal at 7.68 MHz and the sampling clock of the A / D converter 45 is 30.72 MHz as shown in FIG. 6, the IF band of the received signal is 30. The frequency is an integer multiple of .72 MHz, or a frequency separated from the integer multiple frequency by around 7.68 MHz.

  If the reception IF band is 238.08 MHz, the corresponding transmission IF band is 428.08 MHz. In this case, there is only a 2 MHz separation from 430.08 MHz, and a bandwidth of 5 MHz cannot be secured. . Therefore, this combination of detection frequency and sampling frequency is inappropriate.

Next, still another example will be described with reference to FIG. FIG. 7 is an explanatory diagram showing a relationship (example 3) between the IF frequency and aliasing.
As shown in FIG. 7, when the detection of the received signal is 9.52 MHz, the sampling clock of the A / D converter 45 is 30.72 MHz, and the reception IF band is 236.24 MHz, the transmission IF band is 426.24 MHz, The difference from 430.08 MHz is 3.84 MHz. That is, since the center frequency 426.24 MHz ± 2.5 MHz does not overlap with 430.08 MHz, a band of 5 MHz can be secured, and the combination of the detection frequency and the sampling frequency is an appropriate combination.

  As described above, in the second apparatus, the sampling clock of the A / D converter 45 and the detection frequency of the digital quadrature detection are set so that the IF band and aliasing of the received signal and the feedback signal do not overlap, and a highly accurate feedback signal is set. To be able to take out.

Next, the distortion compensation control of the second device will be described.
In the second apparatus, the distortion compensation correction unit 52 corrects the phase compensation amount and the level compensation amount based on the despread pilot channel (CPICH) data output from the demodulation unit 51, thereby correcting the distortion. Update the table.
Specifically, the distortion compensation correction unit 52 uses the CPICH containing a predetermined bit string in the feedback signal from the demodulation unit 51, and the CPICH information before transmission and the CPICH in the feedback signal. The information is compared to detect a level drop and a phase shift.

  Since the feedback signal is transmitted by the adjacent antenna, it can be treated as having no degradation due to propagation. If the reception level of the feedback signal is measured and the receiver gain GRX which is a fixed value is subtracted, The amount of amplitude distortion by the main amplifier 17 can be detected.

In addition, the phase rotation amount, that is, the phase compensation amount taking the complex conjugate can be immediately calculated from the known bit string of CPICH. Since a sufficient S / N ratio can be obtained for the feedback signal, the averaging time can be significantly shorter than that of the conventional DPD.
As described above, by using the information of the known pilot channel for optimization of the distortion compensation value, the processing amount in the distortion compensation correction unit can be reduced.

  Furthermore, in the second apparatus, since the S / N ratio of the feedback signal is good, the despreading time (average processing interval after multiplication of the spreading code) of the demodulator 51 can be shortened, and the processing speed is increased. Can be achieved. As a result, it is possible to immediately correct even an instantaneous peak that could not be followed in the past, and to perform highly accurate distortion compensation.

  Next, the processing in the distortion compensation correction unit 52 will be specifically described with reference to FIG. FIG. 8 is a flowchart showing processing of the distortion compensation correction unit 52 of the second device. FIG. 8 shows processing performed every period T, and a configuration in which the distortion compensation correction unit 52 directly corrects distortion compensation for the modulation unit 11 will be described.

As shown in FIG. 8, when receiving the demodulated data of the feedback signal from the demodulating unit 51, the distortion compensation correcting unit 52 obtains the despreading result c (nT) = a + jb of the pilot channel CPICH (100). Here, n is a natural number, and T corresponds to one symbol time (short code period).
Then, the distortion compensation correction unit 52 calculates the amplitude d (nT) of the feedback signal by d (nT) = a ² + b ² (102).

On the other hand, the distortion compensation correction unit 52 obtains CPICH complex conjugate information C (nT) = (A + jB) in the transmission signal from the modulation unit 11 of the transmission system (120), and the amplitude of the transmission CPICH at time (nT) D (nT) is calculated by D (nT) = A ² + B ² (122).

Then, the distortion compensation correction unit 52 calculates the phase compensation amount P (nT) based on the results of the processing 102 and the processing 120 by P (nT) = (C (nT) * c (nT) ^† ) / d. (104).
Furthermore, the distortion compensation correction unit 52 determines the amplitude distortion correction amount S (nT) based on the results of the processing 102 and the processing 122 as S (nT) = D ÷ d ÷ GRX = exp (logD−logd−logGRX). (106). GRX is a level reduction amount by the receiving system, and is a known constant.

Then, the distortion compensation correction unit 52 gives information on the pilot channel subjected to phase correction and amplitude correction to the modulation unit 11 (110). Specifically, the distortion compensation correction unit 52 performs phase correction on known data to obtain C ((n + 1) T) = C (nT) * P (nT), and further performs amplitude correction to obtain C (( n + 1) T) = C ((n + 1) T) × S (nT) is calculated and output to the modulation unit 11 as an updated CPICH. Then, the modulation unit 11 performs distortion correction on the other transmission data in the same manner as the pilot channel to obtain transmission data.
Specifically, the distortion correction information (phase compensation amount and amplitude correction amount) updated here is applied to the transmission signal at time (mt + nT + t ₀ ). However, m is a natural number that satisfies mt ≦ T, t corresponds to one sample time, and t ₀ corresponds to a processing delay from processing 100 to 110.
In this way, the processing of the distortion compensation correction unit 52 of the second device is performed. In this example, although the update cycle is T, it is updated instantaneously, and the channel power control cycle is 10 symbols, so that it can sufficiently follow rough power fluctuations.

Here, the processing in the case where the distortion compensation correction unit 52 directly corrects distortion compensation for the modulation unit 11 has been described, but the phase and amplitude of the distortion compensation table 53 are stored in the distortion compensation table 53 as shown in FIG. When the distortion compensation coefficient is stored, the distortion compensation table 53 corresponding to the power calculated in the process 122 based on the phase compensation amount calculated in the process 104 and the amplitude distortion calculated in the process 106 is stored. The distortion compensation coefficient may be updated, and the modulation unit 11 may perform the next distortion compensation with reference to the updated distortion compensation table 53.
Alternatively, the processing 100 to 110 is performed at a sample rate period higher than the chip rate without performing despreading, and the processing of averaging P (nT) and S (nT) in an IIR (Infinite Impulse Response) filter is performed. It may be performed after 106.

  The radio base station apparatus according to the second embodiment of the present invention includes a transmission / reception 0-system antenna (Ant0) including an antenna duplexer and a reception-only 1-system antenna (Ant1). A radio base station apparatus that receives data by a diversity method, receives a transmission signal emitted from a 0-system antenna as a feedback signal by a 1-system antenna, and converts a frequency-multiplexed signal of the original received signal and the feedback signal into an A / D The converter 45 undersamples, the digital quadrature detection unit 46 performs quadrature detection, and the FFT unit 47 separates the received signal and the feedback signal by changing the band detected in the time-division FFT processing, and the IFFT unit 50 The time-division inverse FFT conversion is performed, the demodulator 51 digitally demodulates and outputs received data and feedback data, and the distortion compensation correction unit 52 transmits the data. Based on the difference between the pilot channel (CPICH) in the data and the CPICH in the feedback data, the phase compensation amount and the amplitude distortion correction amount are calculated, and the distortion compensation table 53 is updated (or the phase compensation amount and the amplitude are updated in the modulator 11). The distortion correction amount is output, and the modulation unit 11 outputs the next transmission data reflecting the phase compensation amount and the amplitude distortion correction amount). The system can be greatly simplified, the apparatus cost can be reduced and the apparatus can be downsized, and a signal with a high S / N ratio emitted at a close distance is used as a feedback signal and information on a known pilot channel. As a result, the averaging time can be shortened, the processing of the distortion compensation correction unit 52 can be greatly reduced, and the distortion compensation coefficient can be optimized with high accuracy and high speed. There is an effect.

  Further, according to the second radio base station apparatus, in the first system, from the antenna to the digital quadrature detection unit 46, the original received signal and the feedback signal are processed as frequency multiplexed signals without being separated, and the FFT unit After separating both signals in 47, the time division processing is performed with a common circuit configuration, so that the configuration of the apparatus can be further simplified.

  Furthermore, according to the second radio base station apparatus, the sampling frequency in the A / D converter 45 is set to 8 times or more of the chip rate, the one with high resolution (for example, 16 bits or more) is adopted, and the FFT is used. By using such a large-scale filter means, it is possible to secure a band and a dynamic range that can simultaneously handle the reception signal IF band and the transmission signal IF band whose levels change without using analog AGC. Therefore, it is possible to obtain a feedback signal with a high S / N ratio with a simple configuration and to improve the distortion compensation accuracy.

  Furthermore, according to the second radio base station apparatus, the band of the transmission signal (feedback signal) transmitted from the 0-system antenna is greatly attenuated to the 1-system antenna, and the original received signal transmitted from the terminal is Since a filter having a characteristic that does not attenuate the band so much is provided, the levels of the feedback signal and the received signal can be made comparable, and there is an effect that the band can be handled without inconvenience as a frequency multiplexed signal.

Next, a radio base station apparatus according to the third embodiment of the present invention is described.
The third device is similar to the second device in that the signal transmitted by the 0-system antenna is received as a feedback signal by the 1-system antenna, but the processing with the analog signal is performed separately for the received signal and the feedback signal. It is performed by a circuit, and digital processing after digital quadrature detection is performed in a time-sharing manner with a common configuration, and the device configuration can be simplified compared to providing a reception system dedicated to feedback signals, and transmission signals The isolation characteristic from the received signal can be improved.

FIG. 9 is a configuration block diagram of a radio base station apparatus (third apparatus) according to the third embodiment of the present invention. In addition, the part which has the same structure as FIG.1 and FIG.2 is demonstrated using the same code | symbol.
As shown in FIG. 9, the third device includes a 0-system antenna (Ant0) having an antenna duplexer 20 and a reception-only 1-system antenna (Ant1). Since both are the same as the second apparatus shown in FIG.

The 1-system antenna receives an original received signal from the terminal and a feedback signal transmitted from its own 0-system antenna.
The difference from the second device is that the received signal reception circuit includes the same LNA 40, BPF 41, mixer 42, BPF 44, and A / D converter 45 as the second device, and feedback is provided separately. As a signal reception circuit, an attenuator (ATT) 60, a BPF 61, a mixer 62, a BPF 64, and an A / D converter 65 are provided. Further, digital quadrature detection is performed by switching a received signal and a feedback signal in a time division manner. A switch 66 for inputting to the unit 46 is provided.

Among the above components, the BPF 41 of the reception signal system is a filter that passes the frequency band of the reception signal, and is 1950 MHz here. Since the LNA 40, the mixer 42, and the BPF 44 are the same as those in the second device, description thereof is omitted.
The A / D converter 45 samples a received signal with a sampling clock of 61.44 MHz to obtain a received digital signal having a center frequency of 7.68 MHz.
The digital quadrature detection unit 46 performs digital quadrature detection by multiplying the received digital signal by a 7.68 MHz quadrature signal.

The feedback signal system attenuator 60 attenuates the amplitude of the feedback signal to obtain an appropriate reception level.
The BPF 61 is a filter that passes the frequency band 2140 MHz of the feedback signal.

The mixer 62 downconverts the feedback signal to the IF frequency at a frequency from a local oscillator common to the mixer 42.
The BPF 64 band-limits the feedback signal converted to the IF frequency, and here, the center frequency of the pass band is set to 259.12 MHz.

The A / D converter 65 undersamples with a sampling clock 61.44 MHz to obtain a digital signal with a center frequency of 13.36 MHz.
The digital quadrature detection unit 46 'is a quadrature signal of 13.36 MHz and performs digital quadrature detection on the input feedback digital signal.

The switch 66 switches the received digital signal from the digital quadrature detection unit 46 and the feedback digital signal from the digital quadrature detection unit 46 ′ in a time division manner and outputs it to the FFT unit 47.
Since the processing after the FFT unit 47 is the same as that of the second apparatus, description thereof is omitted.

The reception operation in the first system in the third device will be briefly described.
The received signal received by the system 1 antenna is amplified by the LNA 40, band-limited by the BPF 41, down-converted to an IF frequency by the mixer 42, further band-limited by the BPF 44, and down-sampled by the A / D converter 45. Are converted into digital signals, subjected to quadrature detection by the digital quadrature detection unit 46, and output to the FFT unit 47 by the switch 66 in a time-sharing manner.

  On the other hand, the level of the feedback signal received by the system 1 antenna is reduced by the attenuator 60, band-limited by the BPF 61, down-converted to an IF frequency by the mixer 62, and further band-limited by the BPF 64, and an A / D converter. The digital signal is converted into a digital signal by a sampling clock common to the received signal at 65, subjected to quadrature detection by the digital quadrature detection unit 46 ′, and output to the FFT unit 47 by time division by the switch 66.

  The input received signal and feedback signal are subjected to FFT processing in the FFT unit 47 to separate the received signal and the feedback signal, and the feedback signal is frequency converted by the frequency transition unit 48.

The received signal and the feedback signal are each band-limited by the LPF 49, subjected to inverse FFT conversion by the IFFT unit 50, despread by the demodulation unit 51, and output received data and feedback data.
As for the feedback data, the distortion compensation correction unit 52 obtains the difference from the modulation data of the pilot channel in the same manner as the second device, and based on this, the phase compensation amount and the amplitude distortion correction amount are calculated, and the distortion compensation table 53 is obtained. Is updated. The modulation unit 11 of the transmission system performs distortion compensation for the next transmission data based on the updated distortion compensation table 53.
Note that the operations of the 0-system transmission system and the 0-system reception system of the third device are the same as those of the second device, and thus description thereof is omitted.

  In the radio base station apparatus according to the third embodiment of the present invention, the configuration after the digital quadrature detection unit 46 is used in common for the reception signal and the feedback signal, thereby providing an independent reception system for the feedback signal. Compared to the second device, the device configuration can be simplified, and the analog signal portion is separated into two systems so that frequency multiplexing is not performed. There is an effect that can be improved.

  Further, the digital quadrature detection unit 46 is provided after the switch 66 so that the frequency can be adjusted, and is shared by the received signal and the feedback signal. The received signal is quadrature detected at 7.68 MHz, and the feedback signal is quadrature detected at 13.36 MHz. You may make it do.

  INDUSTRIAL APPLICABILITY The present invention is suitable for all radio base station apparatuses that can simplify the apparatus configuration and can compensate for nonlinear distortion of a power amplifier at high speed and with high accuracy. The use of coupling between antennas is not essential as long as it has a receiver and is shared with the feedback system, but in addition to the base station that performs reception divers, AAA (Adaptive Array Antenna) and MIMO (Multiple Input Multiple Output) It can be applied to a wireless LAN router using

FIG. 2 is a configuration block diagram of a radio base station apparatus (first apparatus) according to the first embodiment of the present invention. It is a block diagram of the configuration of a radio base station apparatus (second apparatus) according to the second embodiment of the present invention. It is a schematic explanatory drawing which shows the frequency characteristic of the filter connected to the 1st system antenna of a 2nd apparatus. It is explanatory drawing at the time of providing a reflecting plate in an antenna. It is explanatory drawing which shows the relationship (example 1) of IF frequency and aliasing. It is explanatory drawing which shows the relationship (example 2) of IF frequency and aliasing. It is explanatory drawing which shows the relationship (example 3) of IF frequency and aliasing. It is a flowchart figure which shows the process of the distortion compensation correction | amendment part 52 of a 2nd apparatus. It is a block diagram of a configuration of a radio base station apparatus (third apparatus) according to the third embodiment of the present invention.

Explanation of symbols

  DESCRIPTION OF SYMBOLS 11 ... Modulation part, 12 ... Digital quadrature modulation part, 13 ... Multiplier, 14 ... D / A conversion part, 15, 19, 21, 24, 41, 44, 61, 64 ... Band pass filter, 16, 22, 42 , 62 ... mixer, 17 ... main amplifier, 18 ... directional coupler, 20 ... antenna duplexer, 23 ... frequency oscillator, 25, 45, 65 ... A / D converter, 26 ... delay device, 27 ... adder, 28 ... FFT unit, 29 ... Averaging unit, 30 ... LUT convergence calculation unit, 31,53 ... Distortion compensation table, 40 ... LNA, 46 ... Digital quadrature detection unit, 47 ... FFT unit, 48 ... Frequency transition unit, 49 ... LPF, 50 ... IFFF section, 51 ... demodulation section, 52 ... distortion compensation correction section, 60 ... attenuator, 66 ... switch

Claims (2)

An amplifier for amplifying the transmission signal;
The transmission signal before amplification is multiplied by a distortion compensation coefficient that compensates for nonlinear distortion generated in the amplifier, and a feedback signal of the transmission signal after amplification is input, and distortion included in the feedback signal is fast Fourier transformed. A predistorter that detects and updates the distortion compensation coefficient so that the distortion is minimized,
A radio base station apparatus that transmits and receives radio signals to and from a radio terminal,
A delay circuit that delays the transmission signal before amplification multiplied by the distortion compensation coefficient for a predetermined time;
The radio base station apparatus, wherein the predistorter performs fast Fourier transform on a difference between the feedback signal and the signal delayed by the delay circuit. A modulation unit that digitally modulates transmission data; an amplifier that amplifies the modulated transmission signal; a first antenna dedicated to transmission / reception or transmission; a second antenna dedicated to reception;
Multiply the transmission signal before amplification by a distortion compensation coefficient that compensates for nonlinear distortion generated in the amplifier, and input the feedback signal of the transmission signal after amplification, and update the distortion compensation coefficient based on the feedback signal With a predistorter
A radio base station apparatus that transmits and receives radio signals to and from a radio terminal,
The second antenna is an antenna that receives a radio signal from a radio terminal and receives a transmission signal output from the first antenna as a feedback signal;
An A / D converter that samples the received signal from the wireless terminal received by the second antenna and the feedback signal at the same sampling frequency as a frequency multiplexed signal;
A digital quadrature detection unit for digital quadrature detection of the frequency multiplexed signal digitally converted by the A / D converter;
Fast Fourier transform of the digital quadrature-detected signal in a time division manner to separate a received signal from the wireless terminal and the feedback signal; and
An inverse fast Fourier transform unit that performs inverse fast Fourier transform on the separated signals in a time-sharing manner;
Two demodulating units for digitally demodulating both the signals subjected to the inverse fast Fourier transform and outputting received data and feedback data from the wireless terminal;
The predistorter is a predistorter that updates the distortion compensation coefficient based on a difference between known pilot channel data included in the transmission signal before digital modulation and the feedback data. A wireless base station device.

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