WO2008009633A2 - Procédé d'étalonnage radio - Google Patents

Procédé d'étalonnage radio Download PDF

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Publication number
WO2008009633A2
WO2008009633A2 PCT/EP2007/057240 EP2007057240W WO2008009633A2 WO 2008009633 A2 WO2008009633 A2 WO 2008009633A2 EP 2007057240 W EP2007057240 W EP 2007057240W WO 2008009633 A2 WO2008009633 A2 WO 2008009633A2
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WO
WIPO (PCT)
Prior art keywords
frequency
signal
local oscillator
signals
calibrating
Prior art date
Application number
PCT/EP2007/057240
Other languages
English (en)
Other versions
WO2008009633A3 (fr
Inventor
Frank Op 't Eynde
Original Assignee
Asic Ahead
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Asic Ahead filed Critical Asic Ahead
Publication of WO2008009633A2 publication Critical patent/WO2008009633A2/fr
Publication of WO2008009633A3 publication Critical patent/WO2008009633A3/fr

Links

Classifications

    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
    • H03D3/00—Demodulation of angle-, frequency- or phase- modulated oscillations
    • H03D3/007—Demodulation of angle-, frequency- or phase- modulated oscillations by converting the oscillations into two quadrature related signals
    • H03D3/009—Compensating quadrature phase or amplitude imbalances
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
    • H03D7/00—Transference of modulation from one carrier to another, e.g. frequency-changing
    • H03D7/18—Modifications of frequency-changers for eliminating image frequencies
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/0003—Software-defined radio [SDR] systems, i.e. systems wherein components typically implemented in hardware, e.g. filters or modulators/demodulators, are implented using software, e.g. by involving an AD or DA conversion stage such that at least part of the signal processing is performed in the digital domain
    • H04B1/0007—Software-defined radio [SDR] systems, i.e. systems wherein components typically implemented in hardware, e.g. filters or modulators/demodulators, are implented using software, e.g. by involving an AD or DA conversion stage such that at least part of the signal processing is performed in the digital domain wherein the AD/DA conversion occurs at radiofrequency or intermediate frequency stage
    • H04B1/0014—Software-defined radio [SDR] systems, i.e. systems wherein components typically implemented in hardware, e.g. filters or modulators/demodulators, are implented using software, e.g. by involving an AD or DA conversion stage such that at least part of the signal processing is performed in the digital domain wherein the AD/DA conversion occurs at radiofrequency or intermediate frequency stage using DSP [Digital Signal Processor] quadrature modulation and demodulation
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/0003—Software-defined radio [SDR] systems, i.e. systems wherein components typically implemented in hardware, e.g. filters or modulators/demodulators, are implented using software, e.g. by involving an AD or DA conversion stage such that at least part of the signal processing is performed in the digital domain
    • H04B1/0028—Software-defined radio [SDR] systems, i.e. systems wherein components typically implemented in hardware, e.g. filters or modulators/demodulators, are implented using software, e.g. by involving an AD or DA conversion stage such that at least part of the signal processing is performed in the digital domain wherein the AD/DA conversion occurs at baseband stage
    • H04B1/0032—Software-defined radio [SDR] systems, i.e. systems wherein components typically implemented in hardware, e.g. filters or modulators/demodulators, are implented using software, e.g. by involving an AD or DA conversion stage such that at least part of the signal processing is performed in the digital domain wherein the AD/DA conversion occurs at baseband stage with analogue quadrature frequency conversion to and from the baseband
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/06—Receivers
    • H04B1/16—Circuits
    • H04B1/26—Circuits for superheterodyne receivers
    • H04B1/28—Circuits for superheterodyne receivers the receiver comprising at least one semiconductor device having three or more electrodes

Definitions

  • the present invention relates to the field of calibration methods for radio signals.
  • Receivers of electromagnetic radio signals are used in a wide variety of applications. Examples are standard (music) radio broadcasting, TV broadcasting, digital voice communications such as GSM or DECT, digital data communications such as WiFi or Bluetooth and many others .
  • Modern radio receivers often incorporate a quadrature mixer to convert high-frequency signals into signals with another frequency (typically a lower frequency) .
  • the principle schematic of such a radio receiver is shown in Fig.l.
  • An antenna signal is applied to a first signal conditioning block called high-frequency section (1) .
  • two derived signals are generated, namely an in-phase signal I (2) and a quadrature signal Q (3) .
  • These two signals are further conditioned by two signal conditioning blocks called I-path (4) and Q-path (5) .
  • the resulting signals (6,7) are then combined in a signal demodulator block (8) .
  • the high-frequency section comprises a Low-Noise Amplifier (LNA) and two identical mixers (i.e., signal multipliers) for frequency conversion.
  • the mixers are driven by two local oscillator (LO) signals (10,11) with a phase difference of 90 degrees.
  • LO local oscillator
  • the frequency converted mixer output signals (2,3) are applied to an I-path and Q-path, respectively.
  • These paths each comprise low-pass filters (LPF) , variable gain blocks (VGA) and Analogue-to-Digital Converters (ADC) .
  • LPF low-pass filters
  • VGA variable gain blocks
  • ADC Analogue-to-Digital Converters
  • the resulting signals (6,7) are then further demodulated by dedicated digital circuits .
  • V 9 (t) A 9 .sm( ⁇ g .t) (1)
  • equation (3a) represents a sinusoidal signal that is leading 90° with respect to the signal expressed by (3b) .
  • These two signals are customarily- represented as arrows (14,15) in an I-Q plane, as shown in Fig.3a, whereby the length of the arrows indicates signal magnitude, while their angles with respect to the horizontal axis represent their phases .
  • the two arrows have equal lengths (hence, equal signal amplitudes) while they are placed perpendicularly to each other (indicating a 90° phase difference) .
  • the I-arrow is leading 90° with respect to the Q-arrow.
  • a set of I and Q signals with equal amplitudes and with a phase difference of 90° is denoted as a perfect quadrature signal.
  • the antenna input frequency (16) is now smaller than the Local Oscillator frequency ⁇ co 9 ⁇ co w ) .
  • (3a) represents a sinusoidal signal that is lagging 90° with respect to the signal expressed by (3b) .
  • This can again be represented by two arrows in the I-Q plane, whereby the I-arrow (17) is lagging 90° with respect to the Q-arrow (18) .
  • the demodulator block (8) can only distinguish between antenna signal frequencies larger than ⁇ L0 and frequencies smaller than ⁇ I0 by means of the phases difference between I and Q signal .
  • FIG.3c A more complex situation is shown in Fig.3c where an antenna signal with signal components at both sides of the local oscillator frequency is depicted.
  • the I- signal (19) and the Q-signal (20) have different amplitudes and/or the phase difference is not 90°. They are not a perfect quadrature signal. Again, this can be represented in the I-Q plane.
  • Such a non-perfect quadrature signal can be decomposed in two perfect quadrature signals, represented by sets of arrows 21-22 and 23-24, respectively. In one set, the I-arrow (21) is leading with respect of the Q-arrow (22) while in the other set the I- arrow (23) is lagging.
  • the non-perfect quadrature signal can be decomposed into two components, one originating from the left antenna signal component and one originating from the right antenna signal component.
  • Fig.4 a situation is depicted where the signal gains and/or phases of the I-path and Q-path are not identical.
  • a perfect quadrature signal at the inputs of these signals paths i.e., signals (2) and (3) have the same amplitude and a phase difference of exactly 90°
  • output signals 6 and 7 with unequal amplitude and/or with a phase difference that differs from 90°. This is shown in Fig.4c.
  • Such a non-perfect quadrature signal can be decomposed into a quadrature signal with I leading Q plus another quadrature signal with I lagging Q.
  • the deformed signals 6 and 7 are interpreted by the signal demodulator (8) as if the antenna signal is composed of two signal components situated at both sides of the local oscillator frequency (see Fig.4e) .
  • image signal generation Due to image signal generation, an input signal at a frequency (25) above (or below) the local oscillator frequency is interpreted as if there was a second small signal component at the image frequency (26) , which is smaller (or larger) than the local oscillator frequency.
  • the magnitude of the image frequency signal component is function of the gain error and/or the phase error.
  • the phase error is known, it can be corrected by postprocessing the signals 6 and 7.
  • the next question is to determine how much compensation or correction needs to be applied.
  • the problem of autocorrecting the image signal generation mainly reduces to the problem of measuring the gain and phase errors and determining the appropriate correction coefficients.
  • Fig.5. the radio receiver is coupled to a radio transmitter. Transmitted signals are converted back to LF signals by the receiver. By applying proper test signals to the transmitter inputs (27,28), the generated image signals can be measured by observing signals 6 and 7. With a suitable search algorithm, calibration coefficients can be determined that minimise the image signals after postprocessing .
  • this method assumes that no image signals are generated by the transmitter of Fig.5.
  • this transmitter can suffer from an image signal generation problem, comparable to the image signal generation problem in the receiver. It is then impossible to distinguish between image signals generated by errors in the receiver or by errors in the transmitter. If the receiver would have been calibrated upfront, this method could be used to calibrate the transmitter, or vice versa.
  • the set-up of Fig.5 does not allow determining separate correction coefficients for receiver and transmitter.
  • Fig.6 An alternative approach is depicted in Fig.6.
  • the receiver is calibrated by means of test signals obtained from a reference transmitter.
  • a base station in a cellular network can be assumed to contain a transmitter of superior quality, with virtually no image signal generation.
  • the receiver output signals (6 and 7 in Fig.6) are as shown in Fig.4c-e.
  • the receiver circuitry can now attempt to search for correction coefficients such that after signal postprocessing, the amplitude of the image signal (26) in Fig.4e is minimised. Once these coefficients are known, they can be applied to other received signals.
  • Image tones (30) generated by receiver gain errors or phase errors coincide with the signal tones at the opposite side of the centre frequency. Hence, these image signals are very difficult to detect. In other words, some existing communications standards do not provide adequate test signals for autocalibration of image signal generation. Introducing such test signals would require modifications to existing communications standards. This is a very difficult process.
  • the present invention aims to provide a calibration method that overcomes the problems encountered in the prior art solutions .
  • the present invention relates to a method for calibrating a radio receiver system provided with a frequency conversion circuit comprising local oscillator means operable at a first frequency.
  • the method comprises the steps of - shifting the local oscillator means' frequency to a second frequency offset from the first frequency,
  • said signal is a test signal from a reference transmitter.
  • the reference transmitter is advantageously a base station transmitter from a wireless communications network.
  • the test signal preferably contains the pre-amble of a transmission burst.
  • test signal is a multi-carrier signal.
  • the second frequency is then preferably offset from said first frequency by an amount equal to an integer multiple of half of the carrier frequency spacing.
  • the invention relates to a method for correcting a received signal converted in frequency in a frequency conversion circuit of a radio receiver system, whereby the frequency converting circuit comprises local oscillator means operating at a first frequency.
  • the method comprises the steps of - determining correction information by applying the method as previously described, shifting the local oscillator means' frequency back to the first frequency,
  • Fig . 1 represents a quadrature radio receiver scheme .
  • Fig. 2 represents an implementation of a quadrature radio receiver.
  • Fig. 3 represents some input signals and their corresponding I-Q signals.
  • Fig. 4 represents deformed quadrature components .
  • Fig. 5 represents autocalibration with a loopback.
  • Fig. 6 represents autocalibration with a reference antenna signal.
  • Fig. 7 represents some autocalibration signal spectra.
  • Fig. 8 represents the LO frequency shifting during autocalibration.
  • Fig. 9 represents a flow chart of the image signal reduction autocalibration algorithm according to the invention.
  • Fig.8a shows an antenna signal spectrum (31) that contains energy at frequencies on both sides of the local oscillator frequency.
  • the image signal spectrum (32) coincides (entirely or partially) with the original antenna signal spectrum.
  • the local oscillator frequency is changed, the frequencies of the image signals are changed (see expression (4)).
  • the local oscillator frequency can be changed in such a way that the image signal spectrum is moved away from the original antenna signals.
  • FIG.8a shows the signal spectrum of an IEEE802.16 (WiMax) signal (33), during the pre-amble.
  • WiMax is a wireless communications standard that is using multi- carrier signals, i.e. signals that contain a number of well-defined tones.
  • Image signals (34) generated due to gain or phase imbalance between the I and Q paths coincide with existing tones. They are very difficult to measure.
  • the local oscillator frequency is shifted by an amount, equal to the frequency difference ( ⁇ in Fig.8b) between two tones
  • the image signals are shifted in frequency by an amount of 2 ⁇ (see Fig.8b, right) .
  • the term 'frequency' is used or the pulsation ⁇ , which actually equals, as generally known, frequency multiplied by a factor 2 ⁇ .
  • they fall on frequencies that are allocated to signal tones with no energy during the preamble. In this way, the image signals can be measured easily.
  • the proper gain corrections and phase corrections can be determined to minimise the image signals after signal postprocessing.
  • the local oscillator frequency can be set back to the original value.
  • the calibration coefficients determined during the test phase i.e., the calibration method
  • the algorithm representing the autocalibration method and the method for correcting a received signal as described above is depicted in Fig.9.

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  • Engineering & Computer Science (AREA)
  • Computer Networks & Wireless Communication (AREA)
  • Signal Processing (AREA)
  • Power Engineering (AREA)
  • Superheterodyne Receivers (AREA)
  • Digital Transmission Methods That Use Modulated Carrier Waves (AREA)

Abstract

La présente invention se rapporte à un procédé d'étalonnage d'un système de récepteur radio muni d'un circuit de conversion de fréquence comprenant un moyen d'oscillateur local pouvant fonctionner à une première fréquence. Le procédé comprend les étapes consistant à : - décaler la fréquence du moyen d'oscillateur local vers une seconde fréquence décalée de cette première fréquence, - recevoir un signal grâce au moyen d'oscillateur local pouvant fonctionner à la seconde fréquence, - déterminer par l'intermédiaire du signal reçu des coefficients d'étalonnage permettant de compenser des signaux d'image du signal reçu introduits dans le système de récepteur radio.
PCT/EP2007/057240 2006-07-18 2007-07-13 Procédé d'étalonnage radio WO2008009633A2 (fr)

Applications Claiming Priority (2)

Application Number Priority Date Filing Date Title
EP06447093.3 2006-07-18
EP06447093 2006-07-18

Publications (2)

Publication Number Publication Date
WO2008009633A2 true WO2008009633A2 (fr) 2008-01-24
WO2008009633A3 WO2008009633A3 (fr) 2008-07-17

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PCT/EP2007/057240 WO2008009633A2 (fr) 2006-07-18 2007-07-13 Procédé d'étalonnage radio

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Cited By (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
WO2011106633A1 (fr) * 2010-02-25 2011-09-01 Qualcomm Incorporated Procédés et appareil pour mesurer et/ou utiliser les informations de déséquilibre iq et/ou de décalage en continu d'émetteur et/ou de récepteur

Citations (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US20030139167A1 (en) * 2002-01-24 2003-07-24 Ciccarelli Steven C. System and method for I-Q mismatch compensation in a low IF or zero IF receiver
US20040152436A1 (en) * 2003-01-31 2004-08-05 Ditrans Corporation Systems and methods for coherent adaptive calibration in a receiver
US20050070239A1 (en) * 2003-09-29 2005-03-31 Silicon Laboratories, Inc. Apparatus and method for digital image correction in a receiver

Patent Citations (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US20030139167A1 (en) * 2002-01-24 2003-07-24 Ciccarelli Steven C. System and method for I-Q mismatch compensation in a low IF or zero IF receiver
US20040152436A1 (en) * 2003-01-31 2004-08-05 Ditrans Corporation Systems and methods for coherent adaptive calibration in a receiver
US20050070239A1 (en) * 2003-09-29 2005-03-31 Silicon Laboratories, Inc. Apparatus and method for digital image correction in a receiver

Cited By (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
WO2011106633A1 (fr) * 2010-02-25 2011-09-01 Qualcomm Incorporated Procédés et appareil pour mesurer et/ou utiliser les informations de déséquilibre iq et/ou de décalage en continu d'émetteur et/ou de récepteur
US8630598B2 (en) 2010-02-25 2014-01-14 Qualcomm Incorporated Methods and apparatus for measuring and/or using transmitter and/or receiver IQ imbalance information and/or DC offset information

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WO2008009633A3 (fr) 2008-07-17

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