WO2006043533A1

WO2006043533A1 - Receiver

Info

Publication number: WO2006043533A1
Application number: PCT/JP2005/019093
Authority: WO
Inventors: Yasumi Imagawa; Toshihiro Nagayama; Yasunori Miyahara; Kazunori Yamada
Original assignee: Matsushita Electric Industrial Co., Ltd.
Priority date: 2004-10-19
Filing date: 2005-10-18
Publication date: 2006-04-27

Abstract

A receiver enabled to match a plurality of communication modes by eliminating a DC offset voltage with a simple circuit structure. In this receiver, a first mixer unit (104) converts the output signal of a high-frequency unit (102) into I and Q signals of a first IF frequency by multiplying the output signal of a first local oscillation unit (103). A first channel selection filter (105) limits the band of the output signal of the first mixer unit (104). A second mixer unit (106) converts the output signal of the first channel selection filter (105) into I and Q signals of a second IF frequency by multiplying the output signal of a second local oscillation unit (107). A second channel selection filter (108) limits the band of the output signal of the second mixer unit (106). In accordance with the communication mode, a receiver mode setting unit (111) sets the first local oscillation unit (103), the first channel selection filter (105), the second local oscillation unit (107) and the second channel selection filter (108).

Description

Specification

Receiving machine

Technical field

[0001] The present invention relates to a receiver mounted on a communication terminal of a wireless communication system.

Background art

In recent years, ZIF (Zero Intermediate Frequency) receivers such as direct conversion receivers and LIF (Low Intermediate Frequency) receivers in which the intermediate frequency is set to a low frequency have been actively developed. For such ZIF receivers and LIF receivers, a channel selection filter composed of discrete elements such as SAW (Surface Acoustic Wave) filters and ceramic filters is formed on the IC chip. Therefore, it is expected that the mobile terminal can be downsized and low cost. In addition, UMTS (Universal System for Mobile: one of the wireless communication systems used in digital cellular phones) and W-CDMA (Wideband-Code Division Multiple Access) (Universal / Code Division Multiple Access) The development of multi-mode type receivers (hereinafter referred to as multi-mode receivers) that can receive different communication methods (such as Mobile Telecommunication System: the third generation mobile communication system of Tsubaki Tsubame) on the same terminal has also been carried out. Yes.

[0003] Hereinafter, a conventional receiver will be described. In general, in TSM (Time Division Multiple Access) wireless systems such as GSM, PDC (Personal Digital Cellular) and PHS (Personal Handy-phone System) The receiver performs an intermittent reception operation in order to reduce current consumption. On the other hand, CDMA radio systems such as UMTS (W—CDMA) perform continuous reception during communication.

FIG. 1 is a configuration diagram showing an example of a conventional ZIF receiver. First, a conventional ZIF receiver will be described with reference to FIG. The high frequency filter 2 attenuates a signal outside the reception frequency band among the radio signals received by the antenna 1. The high frequency amplifier 3 amplifies the output signal of the high frequency filter 2. Quadrature mixer 4 has almost the same frequency as the reception frequency. The baseband I and Q signals that are orthogonal to each other are generated by mixing the output signal of the local oscillator 5 that outputs a pair of orthogonal local oscillation signals and the signal amplified by the high-frequency amplifier 3 . The channel selection filter 7 removes unnecessary waves by limiting the bands of the I and Q signals output from the multipliers 4a and 4b of the orthogonal mixer 4. The variable gain amplifier 8 amplifies the I and Q signals output from the channel selection filter 7 to a desired level. The decoding unit 10 converts the I and Q signals amplified by the variable gain amplifier 8 into digital signals by the ADCs 10a and 10b, and decodes them by the decoder. In such a ZIF receiver, the reception sensitivity is degraded by the baseband offset voltage. Such an offset voltage is generated as a result of self-mixing in the quadrature mixer 4 between the mismatch of the circuit components and the local oscillation signal leaked to the high-frequency signal input of the quadrature mixer 4 and the output of the local oscillator 5. The ZIF receiver shown in Fig. 1 eliminates the offset voltage generated in the baseband by providing a high pass filter (HPF) with capacitive coupling 6 and capacitive coupling 9 between each circuit block.

[0005] In the TDMA system, an intermittent reception operation for receiving only a slot for time division is performed. Therefore, the receiver must be started at a high speed to shift to a reception operation. However, as described above, the offset voltage is increased. When capacitive coupling is used to eliminate the HPF cutoff frequency, the DC bias voltage fluctuates at startup. FIG. 2 is an explanatory diagram showing how the bias voltage fluctuates when the ZIF receiver of FIG. 1 is started. As shown in Fig. 2, the DC bias voltage fluctuation 21 occurs at the start-up of the HPF cutoff frequency, so the time constant of the DC bias voltage stabilization time 22 until the receiver stabilizes significantly increases the start-up time. In addition, the reception characteristics may be degraded by the attenuation of the low frequency components of the I and Q signals or the fluctuation of the group delay time due to HPF.

[0006] Means for removing such an offset voltage is disclosed, for example, in Patent Document 1 below. FIG. 3 is a configuration diagram illustrating an example of the offset voltage removal circuit disclosed in Patent Document 1. In FIG. Since the basic configuration of the receiver in FIG. 3 is the same as that in FIG. 1, the description thereof is omitted, and only the portion from which the offset voltage is removed will be described. In FIG. 3, the ADC (Analog to Digital Converter: AD converter) 10a, 10b included in the decoder 10 and the offset voltage detector lOd detect the offset voltages of the I and Q components. The offset voltage is removed by applying negative feedback to the adders 10e, lOf and the adders 11a, l ib. However, such a configuration requires a circuit that removes the offset voltage, and furthermore, it is necessary to adjust the offset voltage to a sufficiently small value with respect to the amplitude of the weak received signal. difficult. Furthermore, the offset voltage removal circuit must be optimized for each wireless system such as GSM and UMTS. In addition, since the offset voltage is also changed by the gain control operation by AGC (Automatic Gain Control), it is necessary to control offset voltage removal to cope with it.

Next, a conventional LIF receiver will be described. Fig. 4 is a block diagram showing an example of a conventional LIF receiver. In FIG. 4, the same components as those in FIG. 1 are denoted by the same reference numerals. In FIG. 4, the high frequency filter 2 attenuates signals other than the reception frequency band among the radio signals received by the antenna 1. The high frequency amplifier 3 amplifies the output signal of the high frequency filter 2. The quadrature mixer 4 has a quadrature relationship by mixing the output signal of the local oscillation unit 5 that outputs a pair of orthogonal local oscillation signals offset by the radio signal power frequency and the signal amplified by the high frequency amplifier 3. Converts to an intermediate frequency (IF) signal of I and Q components. The channel selection filter 7 removes unnecessary waves by limiting the bandwidth of the I and Q signals output from the multipliers 4a and 4b of the orthogonal mixer 4. The variable gain amplifier 8 amplifies the I and Q signals output from the channel selection filter 7 to a desired level. In this LIF receiver, since it is converted to a low IF frequency, the offset voltage generated by the channel selection filter 7 and the variable gain amplifier 8 can be removed by the HPF by capacitive coupling. That is, the channel selection filter 7 may be a BPF (Band Pass Filter). The ADCs 10a and 10b convert the I and Q signals amplified by the variable gain amplifier 8 into digital signals. The second quadrature mixer 12 composed of digital circuits converts the output signals of ADC10a and 10b into digital baseband I and Q signals that are in a quadrature relationship. The decoding unit 10 decodes the output signal of the second orthogonal mixer 12.

FIG. 5 is an explanatory diagram for explaining an example of setting an intermediate frequency in a conventional LIF receiver. That is, FIG. 5 shows an example of setting the intermediate frequency in the conventional LIF receiver described in FIG. In FIG. 5, reference numeral 31 denotes a signal arrangement in the high frequency band, and reference numeral 33 denotes a signal arrangement in the intermediate frequency band (IF band). To receive signal 31a When there are adjacent signal waves 31b, 31c, 31d, 31e, and 31f that are disturbing waves, the first local oscillation is performed so that the output frequency of the first local oscillation unit 5 shown in Fig. 4 is about 1Z2 channel interval. When the frequency 32 is set, in the IF band signal arrangement 33, a reception desired wave 33a and adjacent disturbing waves 33b, 33c, 33d, 33e, 33f are arranged. The adjacent interference wave 33c is an image signal.

[0009] It is well known that an image signal always exists in a reception method using an intermediate frequency as shown in FIG. In the case of a cellular phone using the TDMA system, for example, the frequency used in PDC and PHS is planned so that the adjacent interfering wave that becomes a disturbance is separated by more than the next adjacent channel frequency. In addition, in GSM, standards that use adjacent channel frequencies but are resistant to interference have been relaxed. Therefore, as shown in Fig. 5, it is desirable to set the frequency at which the channel spacing is about 1Z2 as the intermediate frequency. Specifically, IF = 12.5KHz for PDC, IF = 150kHz for PHS, and IF = 100kHz for GSM. Therefore, the signal of the adjacent channel band corresponding to the image frequency is usually suppressed using an image removal mixer. As described above, in the cellular phone using the TDMA system, the adjacent channel frequency is not used or the standard for interference is relaxed. Therefore, in the image removal mixer, if the image signal is about 30 dB, Since it can be easily removed, sufficient anti-jamming characteristics can be secured. On the other hand, because the CDMA system uses adjacent channel frequencies, it is necessary to suppress more than 60 dB using only the image rejection mixer. At this time, the amount of image rejection is determined by the phase orthogonality between I and Q and the coincidence of amplitude. To obtain an image rejection characteristic of 60 dB or more, the quadrature phase error between I and Q is Less than 0.1 degree, the amplitude error between I and Q should be less than 0.1 dB. A technique for image removal (that is, video exclusion) in a LIF receiver is also disclosed in, for example, Patent Document 2 below.

Patent Document 1: Japanese Patent Laid-Open No. 7-111471

Patent Document 2: Japanese Translation of Special Publication 2002-523958

Disclosure of the invention

Problems to be solved by the invention

[0010] However, in the above-described prior art, a plurality of communication methods and communication modes are supported. When configuring a multi-mode receiver, there are various problems as described below. First, if a ZIF receiver is used for TDMA schemes such as GSM, PDC, and PHS, it is difficult to start up the terminal at high speed because the receiver must perform capacitive coupling between the blocks to remove the offset voltage. . In other words, the terminal rises slowly. There is also a problem that it is difficult to realize the function of removing the offset voltage with a simple circuit configuration. On the other hand, when a LIF receiver is used for a CDMA system such as U MTS (W-CDMA), it is difficult to easily realize an image removal mixer to ensure adjacent channel interference characteristics. In this way, if a receiver that supports both TDMA and CDMA is either a ZIF receiver or a LIF receiver, the desired reception characteristics cannot be obtained! There is no selfishness.

An object of the present invention is to provide a receiver capable of removing a DC offset voltage with a simple circuit configuration and supporting a plurality of communication methods without deteriorating a desired reception characteristic. is there.

Means for solving the problem

[0012] A receiver of the present invention includes a first local oscillator that outputs a pair of orthogonal local signals, and a received signal that is orthogonalized by multiplying the output signal of the first local oscillator. A first mixer that obtains I and Q signals of the intermediate frequency, a first filter that band-limits the I and Q signals of the first intermediate frequency, and a second filter that outputs a pair of orthogonal local signals. By multiplying the output signal of the local oscillator and the first filter by the output signal of the second local oscillator, the second intermediate frequency of the frequency is higher than the first intermediate frequency. A second mixer for obtaining I and Q signals, a second filter for band-limiting the I and Q signals of the second intermediate frequency, and a decoding controller for decoding the output signal of the second filter. And controlling the first local transmitter based on a communication method It adopts a configuration comprising a receiver mode setting means for selecting one of the intermediate frequency, a.

The invention's effect

[0013] According to the present invention, it is possible to remove the DC offset voltage with a simple circuit configuration, and it is possible to cope with a plurality of communication methods and communication modes.

Brief Description of Drawings [0014] [FIG. 1] Configuration diagram showing an example of a conventional ZIF receiver

[Fig. 2] An explanatory diagram showing how the bias voltage fluctuates when the ZIF receiver of Fig. 1 is started. [Fig. 3] A configuration diagram showing an example of a conventional offset voltage elimination circuit.

圆 4] Configuration diagram showing an example of a conventional LIF receiver

FIG. 5 is an explanatory diagram for explaining an example of setting an intermediate frequency in a conventional LIF receiver. FIG. 6 is a configuration diagram showing a receiver according to Embodiment 1 of the present invention.

FIG. 7 is an explanatory diagram for explaining an outline of a GSM frame configuration taking the TDMA scheme as an example in the receiver of the present invention.

FIG. 8 is an explanatory diagram for explaining the outline of the frame structure of UMTS using the W-CDMA system as an example in the receiver of the present invention.

FIG. 9 is a configuration diagram of an offset voltage removal circuit according to the second embodiment of the present invention.

FIG. 10 is a block diagram showing a basic configuration of an amplifying apparatus according to Embodiment 4 of the present invention.

FIG. 11A is a first explanatory diagram illustrating an input signal amplification method according to Embodiment 4 of the present invention. FIG. 11B is a first explanatory diagram illustrating an input signal amplification method according to Embodiment 4 of the present invention. FIG. 12A is a second explanatory diagram illustrating an input signal amplification method according to Embodiment 4 of the present invention. FIG. 12B is a second explanatory diagram illustrating an input signal amplification method according to Embodiment 4 of the present invention. FIG. 13 is a block diagram showing a basic configuration of an amplifying apparatus according to Embodiment 5 of the present invention.

FIG. 14 is a diagram showing a level diagram of the receiver in the embodiment 5 shown in FIG. 13. FIG. 15 is a circuit diagram showing a detailed configuration of the band eliminate filter in FIG.

BEST MODE FOR CARRYING OUT THE INVENTION

[0015] Hereinafter, several embodiments of the receiver of the present invention will be described in detail with reference to the drawings. In the drawings used in the respective embodiments described below, the same constituent elements are denoted by the same reference numerals, and redundant description is omitted as much as possible.

[0016] <Embodiment 1>

FIG. 6 is a block diagram showing a receiver according to Embodiment 1 of the present invention. In FIG. 6, the receiver includes an antenna unit 101, a high frequency unit 102, a first local oscillation unit 103, a first mixer unit 104, a first channel selection filter 105, a second mixer unit 106, and a second local unit. Oscillator 107, second channel selection filter 108, variable gain amplifier 109, decoder 110, receiver module The mode setting unit 111 and the third local oscillation unit 112 are provided.

First, the function of each component of the receiver shown in FIG. 6 will be described. Each component is composed of one or more integrated circuits. The antenna unit 101 has a function of receiving radio signals in a plurality of frequency bands by the plurality of antennas 101a and 101b. The high frequency unit 102 includes a high frequency filter corresponding to a plurality of frequency bands and a plurality of high frequency amplifiers, and the radio signal force received by the antenna unit 101 also attenuates frequencies other than the necessary band and amplifies the necessary band frequency. It has a function. The first local oscillation unit 103 has a function of outputting a pair of orthogonal local signals. The first mixer section 104 has a function of converting the output signal of the high-frequency section 102 into the I and Q signals of the orthogonal first IF frequency f12 by multiplying the output signal of the first local oscillation section 103. Has

[0018] The first channel selection filter 105 has a function of selecting a desired signal by band-limiting the I and Q signals that are output signals of the first mixer unit 104. Note that the first channel selection filter 105 is an LPF (Low Pass Filter) and includes an offset voltage that hinders signal processing. This offset voltage is removed by the HPF (High Pass Filter). Therefore, the first channel selection filter 105 results in a BPF (Band Pass Filter). The first channel selection filter 105 may be a filter specially designed as a BPF.

[0019] The second local oscillator 107 has a function of outputting a pair of orthogonal local signals. The second mixer unit 106 multiplies the I and Q signals output from the first channel selection filter 105 by the output signal of the second local oscillation unit 107, thereby obtaining the second IF frequency fl3, It has a function to convert to Q signal. However, the relationship between the two frequencies is the first IF frequency f 12 and the second IF frequency f 13. The second channel selection filter 108 has a function of selecting a desired signal by band-limiting the output signal of the second mixer unit 106. The second channel selection filter 108 is a plurality of filters having different passbands or a channel selection filter having a filtering power capable of variably setting the passband, and also has a function of setting a plurality of passband widths and center frequencies. ing.

[0020] The variable gain amplifier 109 is configured to output the signal output from the second channel selection filter 108. A function of adjusting the amplitude to a preset amplitude by control from the decoding unit 110 is provided. The variable gain amplifier 109 uses the intermediate frequency band higher than the baseband or the low IF band, so that the decoding unit 110 can be capacitively coupled to the second channel selection filter 108 and the subsequent stage in the previous stage. Therefore, it is not necessary to remove the offset necessary for the ZIF receiver.

[0021] The decoding unit 110 includes an AD converter unit 110a, a digital mixer unit 110b, a CDMA decoding unit 110c, a TDMA decoding unit 11 Od, and an AGC (Automatic Gain Controller) lOe. The AD converter unit 110a has a function of converting the output signal of the variable gain amplifier unit 109 into digital analog power. The digital mixer section 110b has a function of converting the output of the AD converter 110a into digital I and Q signals. Furthermore, AGClOe has a function of controlling the gain of the variable gain amplifier unit 109 according to the signal amplitude. The CDMA decoding unit 110c has a function of decoding, for example, a UMTS (W-CDMA) signal. Further, the TDMA decoding unit 110d has a function of decoding, for example, a GSM signal.

[0022] The receiver mode setting unit 111 includes a first local oscillation unit 103, a first mixer unit 104, a first channel selection filter 105, and a second local unit according to a CDMA or TDMA communication method. A function for setting the oscillation unit 107, the second channel selection filter 108, and the decoding unit 110 is provided.

The third local oscillating unit 112 has a function of generating a reference clock signal for the decoding unit 110. The third local oscillating unit 112 may be shared if the frequency of the third local oscillating unit 112 is double or divided with the second local oscillating unit 107.

[0024] With such a configuration, the receiver switches the receiver mode between the ZIF receiver mode and the LIF receiver mode according to the communication method (that is, the CDMA method or the TDMA method). Specifically, the receiver sets the frequencies of the first local oscillator 103 and the second local oscillator 107 based on the receiver mode, and selects the first channel selection filter 105 and the second channel selection. The filter 108 is configured as a filter having a pass characteristic optimum for each communication method.

[0025] When setting to the ZIF receiver mode, the receiver sets the first local oscillation unit 103 to a frequency substantially the same as the reception frequency. As a result, the I and Q signals of the first IF frequency fl2 are Source signal. Further, the I and Q signals are quadrature modulated by the second local oscillation unit 107 and the second mixer unit 106, and are upmixed to the second IF frequency fl3.

[0026] On the other hand, when setting to the LIF receiver mode, the receiver sets the first local oscillation unit 103 to a frequency slightly offset from the reception frequency by the frequency of the first IF frequency fl2. . Further, the I and Q signals are orthogonally modulated by the second local oscillator 107 and the second mixer 106, and are upmixed to the second IF frequency f13.

[0027] Next, the use of each receiver mode of the ZIF receiver mode and the LIF receiver mode will be specifically described by taking a digital communication system as an example. Currently, TDMA and CDMA are the typical digital communication systems used in mobile phones. As described in the background art, the ZIF receiver needs to remove the offset voltage generated in the baseband part. As shown in Fig. 1, the easiest way to remove this offset voltage is to capacitively couple between circuit blocks. In this case, however, it has an HPF characteristic and is cut off so as not to affect the reception sensitivity characteristic. Setting the frequency makes the time constant very long.

FIG. 7 is an explanatory diagram for explaining an outline of a GSM frame configuration using the TDMA scheme as an example in the receiver of the present invention. As shown in FIG. 7, since GSM is a TDMAZFDD method, the timing of reception slot 200, monitor slot 201, and intermittent reception operation 202 shown in FIG. 7 will be described here. For example, the monitor slot 201 receives the monitor slot 201a, and the reception slot 200 receives the reception slot 20 Oa designated as the communication slot. In GSM, one frame is 4.615 ms, one slot power is 77 seconds, and one frame is composed of 8 slots. Receive operation will be performed. The intermittent reception operation 202 is an indispensable method for reducing current consumption. However, in the configuration shown in Fig. 1, the time required until the bias voltage is stable when the receiver is activated and becomes receivable has the relationship shown in Fig. 2, which causes an obstacle to the intermittent reception operation. Therefore, the ZIF receiver mode is suitable for the TD MA system! /!

[0029] However, in the LIF receiver mode, it is converted to an intermediate frequency equivalent to about 1Z2 channel spacing. Therefore, even if an HPF that removes the offset voltage is placed, ZIF Compared to the receiver mode, the cutoff frequency of the HPF can be set higher, and the start-up time can be easily increased. Therefore, LIF receiver mode is suitable for TDMA system.

[0030] Next, as an example of the CDMA system, a frame structure of UMTS (W-CDMA) will be described. FIG. 8 is an explanatory diagram for explaining the outline of the frame structure of UMTS, taking the W-CDMA system as an example in the receiver of the present invention. Since UMTS (W-CDMA) is a CDM AZFDD system, the received frame 300 shown in FIG. 8 will be described here. In UMTS (W-CDMA), one frame is 10 ms, and since continuous reception is basically performed at the specified transmission rate from the base station during communication, intermittent reception is not performed as shown in intermittent reception operation 301. And the signal band is wide. Therefore, even if capacitive coupling HPF is used to remove the offset between circuit blocks, the cut-off frequency can be set high to some extent.Therefore, as shown in FIG. It can be shortened. Therefore, ZIF receiver mode is suitable for CDMA.

[0031] As described above, it is appropriate to set the ZIF receiver mode in the CDMA system and the LIF receiver mode in the TDMA system.

Next, referring to FIG. 6, when switching between the ZIF receiver mode and the LIF receiver mode, the first local oscillator 103, the first channel selection filter 105, and the second local oscillator 107 Each setting of second channel selection filter 108 and decoding section 110 will be described.

[0033] As a ZIF receiver mode, a UMTS (W-CDMA) receiver will be described as an example, and as a LIF receiver mode, a GSM receiver will be described as an example. First, the case of switching to the ZIF receiver mode, that is, reception of a UMTS (W-CDMA) signal will be described with reference to FIG. The output frequency of the first local oscillator 103 is set to be almost the same as the reception frequency, and the overall characteristics of the first channel selection filter 105 and the second channel selection filter 1 08 are UMTS (W-CDMA) transmission. 3. The speed should be 3.84 Mcps baseband signal, and adjacent channel signal band should be filtered. A specific example of the low-frequency pass characteristics of the first channel selection filter 105 has been published in Reference 1 below, and the HPF has a cutoff frequency of about 20 KHz or less that does not affect the reception characteristics. Appropriate.

[0034] Next, the case where the reference clock of UMTS (W—CDMA) is 19.2 MHz is explained. Light up. Taking the second local oscillator 107 as an example in which the reference clock is 19.2 MHz and the output frequency of 1Z4 is 3.84 MHz, the second IF frequency fl3 is also 3.84 MHz. Assuming that the sampling frequency of the AD converter 110a is 15.36 MHz, the second IF frequency f13 is a 3.84 MHz digitized signal. The digital mixer 咅 110b converts the digital I and Q signals with 15.36MHz 1Z4 orthogonal clocks, and the CDMA decoder 110c performs the decoded signal processing.

[0035] Next, the case of switching to the LIF receiver mode, that is, reception of a GSM signal will be described. The output frequency of the first local oscillating unit 5 is a frequency that is detuned (that is, offset) from the reception frequency by 1Z2 channel interval or more. A specific example of the first IF frequency f12 is presented in Reference 2 below. Further, it is described that the low frequency of the first channel selection filter 105 may be about 10 kHz.

Here, a case where the GSM reference clock is 13 MHz will be described. Now, if the first IF frequency fl2 is 135.4167 KHz, which is 1Z96 with a reference clock of 13 MHz, and the output frequency of the second local oscillator 107 is 1Z4 of the reference clock, 3.25 MHz, then the second IF frequency The wave number fl3 is 3.385417MHz. It is assumed that the second channel selection filter 108 secures an attenuation amount that is insufficient in the first channel selection filter 105.

[0037] If the sampling clock of the AD converter 110a is 3.25 MHz, the second IF frequency f13 is undersampled, and the digitized 135.4167KHz is obtained at the output of the AD converter 110a. The digital I and Q signals are obtained from the 1Z96 orthogonal clock of 13 MHz clock and the digital mixer 110b, and the TDMA decoder 110d performs the decoding process.

[0038] Next, a specific example of characteristic switching in each block for making both the LIF receiver and the ZIF receiver compatible will be described. In the ZIF receiver, the first channel selection filter 105 is a low-pass filter (LPF) with a cut-off frequency of about 2 MHz. In order to remove the DC offset voltage generated in the circuit, the capacitive coupling is performed as described above. A band pass filter (BPF) can be realized by combining a high pass filter (HPF). The second channel selection filter 108 can be realized by configuring a band pass filter (BPF) having a center frequency of 3.84 MHz and a pass band of about 4 MHz. In the LIF receiver, the first channel selection filter 105 is an LPF with a cutoff frequency of about 200 kHz. The second channel selection filter 108 can be realized by configuring a band pass filter (BPF) having a center frequency of 3. 385 417 MHz and a pass band of 400 kHz. The image frequency of the second IF frequency f 13 of the LIF receiver is + 6. 771 MHz or 6.771 MHz away from the desired signal in the high frequency band. Cut of the first channel selection filter If the off-frequency is 200 kHz, the image signal with the second IF frequency f 13 can be removed. If image signal removal is insufficient, it may be used as a notch filter.

[0040] The first channel selection filter 105 and the second channel selection filter 108 of the ZIF receiver and the LIF receiver can be configured by being individually arranged and switched appropriately. In addition, if the structure of the ZIF receiver filter is configured by switching the constants of the LIF receiver, the receiving circuit can be greatly reduced in size.

[0041] Thus, according to the present embodiment, it is possible to provide a receiver capable of supporting a plurality of communication systems (for example, TDMA system and CDMA system) without providing a special offset voltage removal circuit. it can.

Next, a receiver according to Embodiment 2 of the present invention will be described. As described in Embodiment 1 above, it is not necessary to provide a special offset voltage removal circuit to construct a receiver that can handle a plurality of communication methods. However, the DC offset voltage caused by the mismatch of the components in the second mixer section 106 in Fig. 6 appears as carrier leak at the second IF frequency f 13 and consequently affects the reception sensitivity. It is also possible to do. In such a case, the offset voltage may be canceled as necessary. Therefore, in the second embodiment of the present invention, a configuration example for canceling the offset voltage will be described.

FIG. 9 is a configuration diagram of the offset voltage removal circuit according to the second embodiment of the present invention. Therefore, the second embodiment will be described with reference to FIGS. The first local oscillating unit 103 is set in a state where the high frequency signal is not frequency-converted by the first mixer unit 104 in the OFF state. As a result of this operation, it is possible to set no signal input even in areas where strong received signals exist. At this time, the gain for setting the variable gain amplifier section 109 is In general, it is desirable to set the gain approximately 10 to 20 dB higher than the gain at the time of sensitivity point signal. By this operation, carrier leak can be detected efficiently and adjustments can be made so as not to affect the reception sensitivity. In addition, since the noise band can be narrowed by setting the filter after the second intermediate frequency to the GSM setting, the carrier leak can be detected more efficiently.

[0044] In FIG. 9, the carrier leak occurring at the second IF frequency f13 is read by the carrier detector l lOg based on the RS SI output of AGCl lOe, and the adjustment voltage is output from DAC l lOi and added. Adjust to vessel 114 to minimize carrier leakage. This adjustment operation is performed individually for the I channel or Q channel. At this time, the second mixer section 106 on the I channel or Q channel side that has not been adjusted should not perform the frequency conversion operation. Although not specifically explained with reference to the figure, it can be realized, for example, by turning off the power of the mixer or not inputting the second local oscillation signal.

[0045] Next, the adjusted DAC control values of the I and Q channels are stored in the setting information holding unit lOh, and are output to DAClOi at the time of reception. Such adjustment and storage may be performed at every reception, or may be performed every preset time or every factory shipment adjustment of the mobile terminal device. In other words, such adjustment and storage can be performed at any time. By adding a few functions in this way, stable offset voltage removal can be realized.

Next, a receiver according to Embodiment 3 of the present invention will be described. The power described in the first embodiment for UMTS (W-CDMA) and GSM receivers is not limited to these wireless communication systems. AMPS (Advanced Mobile Phone Service: analog / cellular mobile) It can also be applied to OFDM (Orthogonal Frequency Division Multiplexing) and UWB (Ultra Wideband) for digital communications. In addition, even in the case of the TDMA scheme, any communication system that satisfies the radio standard in the ZIF receiver mode may be used in those ZIF receiver modes. Furthermore, even if it is a CDMA system, it may be used in the LIF receiver mode as long as the communication system satisfies the radio standard in the LIF receiver mode. For example, in a wireless LAN such as IEEE802.llg, frequency hopping, C Since it is appropriately used in the DMA and OFDM communication modes, it can be used in an optimal receiver state.

[0047] Also, the ability to modularize the CDMA decoding unit 110c and the TDMA decoding unit l lOd of the decoding unit 110 shown in FIG. 6, a card that can change the wireless communication system that can be decoded by software, or By using a combination of both, it is possible to change from UMTS (W-CDMA) to IS95, or from GSM to PHS. By storing the software in, for example, a removable memory, it is possible to realize a receiver that supports many communication methods.

Furthermore, the frequency characteristics and Q of the first channel selection filter 105, the second channel selection filter 108, and the digital filter mounted in the decoding unit 110 can be varied as shown in FIG. With a circuit configuration, it is possible to set the frequency characteristics and Q value of the filter for any communication method. With such a configuration, it is possible to configure a receiver that supports any communication method within the receivable frequency band.

[0049] It should be noted that the power described in the first and third embodiments for the multimode receiver is not limited to this. In other words, for single-mode receivers that use TDMA or CDMA, the fixed integration in LIF receiver mode or ZIF receiver mode has been used so far for each wireless communication system. Although the receiver is configured by a circuit, receivers of various wireless communication systems can be configured by the same integrated circuit. Further, the receivers of Embodiments 1 to 3 can be mounted on a communication terminal in combination with a transmitter.

[0050] <Embodiment 4>

FIG. 10 is a block diagram showing a basic configuration of the amplifying apparatus according to Embodiment 4 of the present invention. First, the configuration of the amplification device shown in FIG. 10 will be described. The amplifying apparatus modulates the first I channel signal and the first Q channel signal, which are orthogonal baseband signals, and band-limits the output signal of the orthogonal modulation unit 1000, and the first Band-eliminate filter 1001 that removes the carrier leakage of the orthogonal modulation output that occurs according to the DC offset voltage of the I-channel signal and the first Q-channel signal, and the amplification unit 1002 that amplifies the output signal of the band-eliminate filter 1001 And the output of the amplifier 1002 The configuration includes a second I channel signal, which is a baseband signal orthogonal to the signal, and an orthogonal demodulation unit 1003 that performs orthogonal demodulation to the second Q channel signal.

Next, the operation of the amplifying device shown in FIG. 10 will be described. As shown in FIG. 10, a band-eliminating filter 1001 is provided on the output side of the quadrature modulation unit 1000, and is generated according to the DC offset voltages of the first I channel signal and the first Q channel signal that are baseband signals. The carrier leak of the output signal of the quadrature modulator 1000 is removed. As a result, the SZN ratio of the input signal of the quadrature modulation unit 1000 can be prevented from being lowered, and the amplification unit 1002 can amplify the baseband signal.

[0052] Here, a method for amplifying a signal while preventing a decrease in the SZN ratio of the input signal due to the DC offset voltage of the first I channel signal and the second Q channel signal will be described. FIG. 11 is a first explanatory diagram showing the method of amplifying the input signal related to the amplifying device shown in FIG. 10. FIG. 11A shows the output waveform of the quadrature modulator 1000, and FIG. 11B shows the output waveform of the band-eliminate filter 1001. ing. 12 is a second explanatory diagram illustrating an amplification method of the input signal related to the amplification device shown in FIG. 10. FIG. 12A is an input waveform of the quadrature demodulation unit 1003, and FIG. 12B is an output waveform of the quadrature demodulation unit 1003. Is shown. Therefore, the method for amplifying the input signal will be described with reference to FIG. 10, FIG. 11, and FIG.

[0053] On the output side of quadrature modulation section 1000, the DC offset voltages of the first I channel signal and the first Q channel signal are converted into carrier leak 1101 shown in FIG. 11A, and the first I channel signal And the first Q channel signal is converted to an input signal 1102. At this time, the carrier leak 1101 appears superimposed on the input signal 1102. The magnitude of the carrier leak occurs according to the magnitude of the DC offset voltage of the first I channel signal and the first Q channel signal, and the SZN ratio of the input signal 1102 is the ratio of the input signal 1102 to the carrier leak 1101. Become

As shown in FIG. 11A, carrier leak 1101 on the output side of quadrature modulation section 1000 is suppressed by a band eliminate filter 1001 having a band eliminate filter cutoff frequency 1103 shown in FIG. 11B. As a result, the carrier leak 1104 is suppressed, and the signal ratio between the input signal 1105 and the carrier leak 1104 is increased. Therefore, the SZN ratio of the input signal is improved and good reception characteristics can be secured. [0055] After that, the amplification unit 1002 amplifies this signal (that is, an input signal with an improved SZN ratio), and then the quadrature demodulation unit 1003 performs the second I-channel signal and the second I-channel signal as the baseband signals. By converting (demodulating) the signal to a Q channel signal, the baseband signal can be amplified to a desired level while suppressing an increase in the baseband DC offset voltage. Note that a single pass filter or a band pass filter that performs band limitation may be provided between the orthogonal modulation unit 1000 and the orthogonal demodulation unit 1003, and band limitation may be performed at the IF frequency after the orthogonal modulation unit 1000. As a result, band limitation can be performed while suppressing an increase in the baseband DC offset voltage.

[0056] Here, a state in which band limitation is performed at the IF frequency by a low-pass filter or a band-pass filter provided between the orthogonal modulation unit 1000 and the orthogonal demodulation unit 1003 will be described with reference to FIG. This figure shows how the band is limited by the band pass filter. First, as shown in FIG. 12A, the carrier leak 1203 is removed by the band elimination filter cutoff frequency 1201 by the band elimination filter 1001, and the vicinity of the input signal 1204 is band-limited by the bandpass filter cutoff frequency 1202 by the bandpass filter. After that, the quadrature demodulator 1003 converts the baseband signal into the second I channel signal and the second Q channel signal, thereby blocking the band elimination filter 1001 and the bandpass filter as shown in FIG. 12B. The band is limited by the frequency band, and a desired input signal 1206 can be obtained. As a result, it is possible to limit the band of the baseband signal while avoiding an increase in the DC offset voltage of the baseband.

Note that the amplifying unit 1002 shown in FIG. 10 may be configured as a variable amplifier. By using such a variable amplifier, it is possible to variably amplify the input signal while avoiding a change in the baseband DC offset voltage.

[0058] Hereinafter, a method for variably amplifying an input signal while avoiding a change in the direct current offset voltage of the first I channel signal and the first Q channel signal will be described with reference to FIGS. 10 and 11. Explain. Also in this case, the SZN ratio of the input signal 1105 is improved by the same operation as that described in the case of the amplification unit 1002 in which the amplification degree is not variable, and the reception characteristic can be ensured satisfactorily. In addition, since the flow of operation during this time is exactly the same as described above, avoid duplication. Therefore, the description is omitted. After that, variable amplification of the signal is performed by the variable amplifier corresponding to the amplification unit 1002 at the IF frequency, and the signal is converted into the second I channel signal and the second Q channel signal by the orthogonal demodulation unit 1003. As a result, even when the gain of the variable amplifier is changed, variable amplification can be performed while avoiding a change in the DC offset voltage of the baseband signal.

FIG. 13 is a block diagram showing a basic configuration of the amplifying apparatus according to Embodiment 5 of the present invention. In this figure, the configuration of a receiver using an amplification device is shown. Hereinafter, the receiver according to the fifth embodiment will be described with reference to FIG.

The receiver shown in FIG. 13 receives an antenna 1301 that receives a radio signal, a bandpass filter 1302 that limits the band of the signal received by the antenna 1301, and a signal that is band-limited by the bandpass filter 1302. A high-frequency amplifier 1303 to amplify, a first quadrature demodulator 1304 comprising mixers 1305 and 1306 that quadrature-demodulates the signal amplified by the high-frequency amplifier 1303, and a first quadrature demodulator 1304 having a phase difference of 2 The first 90 degree phase shifter 1307 that supplies two signals, the first local oscillator 1308 that is the signal source of the first 90 degree phase shifter 1307, and the output of the first quadrature demodulator 1304 are orthogonal The baseband amplifiers 1309 and 1310 that amplify the I channel signal and the Q channel signal, the low pass filters 1311 and 1312 that limit the output signals of the baseband amplifiers 1309 and 1310, respectively, and the low pass filters 1311 and 1312 Output signal A quadrature modulator 1313 comprising mixers 1315 and 1316, and a second 90-degree phase shifter for supplying two signals having a phase difference of 90 degrees to the quadrature modulator 1313 and a second quadrature demodulator 1322 described later. 1317, a second local oscillator 1318 that is a signal source of the second 90-degree phase shifter 1317, a band-eliminate filter 1319 that band-limits an IF signal that is an output signal of the quadrature modulator 1313, and a band-eliminate filter 13 19 Low-pass filter 1320 that further band-limits the IF signal band-limited by IF, variable amplifier 1 321 that variably amplifies the IF signal band-limited by low-pass filter 1320, and quadrature demodulation of the IF signal that is variably amplified by variable amplifier 1321 The second quadrature demodulator 1322 including the mixers 1323 and 1324 and the demodulator 1325 that demodulates the output signal of the second quadrature demodulation 1322 are provided. Note that the amplifier 1326 includes a quadrature modulator 1313, a second 90-degree phase shifter 1317, a second local oscillator 1318, a non-eliminate finalizer 1319, a Ronos finalizer 1320, and a variable amplifier 1321 and a second quadrature demodulator 1322.

FIG. 14 is a diagram showing a level diagram of the receiver in the fifth embodiment shown in FIG. In this figure, the horizontal axis represents the items of each component shown in FIG. 13, and the vertical axis represents the signal level. Therefore, the level diagram of the receiver shown in FIG. 13 will be described with reference to FIG. Note that the same reference numerals are given to the reference numerals of the constituent elements in FIG. 13 and the reference numerals of the constituent elements shown on the horizontal axis of the level diagram of FIG.

[0063] Conventionally, since the baseband amplifiers 1309 and 1310 and the low-pass filters 1311 and 1312 obtain a large gain, the DC offset voltages of the I-channel signal and the Q-channel signal, which are baseband signals, are amplified. There was a problem that the SZN ratio of the battery deteriorated. Therefore, in the receiver of the present invention, as shown in the level diagram of FIG. 14, the signal whose level is attenuated by the bandpass filter 1302 is slightly amplified by the high-frequency amplifier 1303 and the first quadrature demodulator 1304, and further amplified. In other words, the baseband amplifiers 1309 and 1310 amplify slightly as the minimum gain setting to obtain the required reception sensitivity, and suppress the generation of DC offset voltage in the baseband part.

[0064] Furthermore, the Rhonos-Finolators 1311 and 1312, the quadrature modulator 1313, the node eliminator filter 1319, and the low-pass filter 1320 reduce the carrier leak on the output side of the quadrature modulator 1313 while keeping the signal level constant. By eliminating the band elimination filter 1319, the reduction of the SZN ratio of the received signal is prevented.

[0065] Then, for amplification of the received signal, the variable amplifier 1321 provided in the IF frequency band of the output signal of the quadrature modulator 1313 is amplified with a large gain to increase the signal level, and the second quadrature demodulator The signal is demodulated by 1322. As a result, even when the gain of the variable amplifier 1321 is changed, variable amplification can be performed while avoiding a change in the DC offset voltage of the baseband signal. This also eliminates the need for a DC offset voltage elimination device that was conventionally required for each of the I-channel signal and the Q-channel signal, so that the circuit scale of the receiver can be reduced. Furthermore, since the conventional DC offset voltage elimination device is composed of an analog negative feedback circuit, Convergence time indicating the time to complete the removal of the offset voltage has become a problem. However, the present invention does not require a DC offset voltage elimination device for the analog negative feedback circuit, so the DC offset voltage elimination is performed at high speed. This eliminates the need for complicated control for removing the DC offset voltage and simplifies the receiver design.

FIG. 15 is a circuit diagram showing a detailed configuration of the band-eliminated filter 1319 of FIG. Therefore, the configuration of the band-eliminated filter 1319 in FIG. 13 will be described in more detail with reference to FIG. The band-eliminated filter 1319 can be configured using a generally known Twin-T type band-eliminating filter. The band elimination filter 1319 is connected to the output stage of the Twin-T type band elimination filter 1500 at the output stage of the buffer amplifier 1501, the variable amplifier 1502 for changing the feedback amount of the output signal of the buffer amplifier 1501, and the output side of the variable amplifier 1502. And buffer amplifier 1503.

[0067] By setting the gain of variable amplifier 1502, the Q value of band-eliminated filter 1319 can be set. For example, as the gain of the buffer amplifier 1501, the variable amplifier 1502, and the buffer amplifier 1503 approaches 1 time (that is, the loop gain becomes 1 time), the Q value increases to an infinitely large value. Therefore, by setting the gain of the variable amplifier 1502, a band-eliminated filter 1319 having a steep cut-off frequency characteristic can be realized. Therefore, if the received signal is in the vicinity of a direct current or direct current, the above-mentioned Twin-T type band elimination filter 1319 can be used to suppress the missing of the received signal component and remove the carrier leak. can do. If the received signal component does not exist in the vicinity of the direct current or direct current, the Q value of the band elimination filter may be reduced. In that case, you can use a band elimination filter other than the Twin-T type mentioned above!

[0068] The second local oscillator can be obtained by using a band elimination filter with a variable cutoff frequency as shown in FIG. 13 as a non-eliminating filter 1319, f array, and Japanese Unexamined Patent Publication No. 2001323652. The cut-off frequency of the band-eliminated filter 1319 can be automatically matched to the output frequency of 1318 (that is, the carrier leak frequency).

[0069] Thus, when the cutoff frequency of the band eliminate filter 1319 fluctuates due to element variations and temperature characteristics when the band eliminate filter 1319 is integrated. However, the carrier leak can be appropriately removed by the band eliminate filter 1319. Therefore, the configuration as shown in FIG. 13 eliminates the need for individual adjustment for each receiver, thereby further improving the productivity of the amplification device and the receiver.

[0070] It should be noted that Embodiment 5 described above applies the amplification device of the present invention to a receiver. The amplification device of the present invention may be used for a transmitter. Further, a communication device may be configured by using the amplification device of the present invention in combination with a receiver and a transmitter. By doing so, it is possible to construct a receiver, a transmitter, and a communication device that avoid the influence of the DC offset voltage.

[0071] (Reference 1)

2000 IEEE Radio Frequency Integrated Circuit symposium MOM3B-2 Analog bas ebandIC for use in direct conversion WCDMA receiver.

(Reference 2)

2000 IEEE Radio Frequency Integrated Circuit Symposium MOM3B— 3 A LOW IF POLYPHASE Receiver for GSM using log domain signal processing

[0072] This specification is based on Japanese Patent Application No. 2004-304051 filed on October 19, 2004 and November 9, 2004.

Based on Japanese Patent Application No. 2004-325228. All these contents are included here. Industrial applicability

[0073] The receiver of the present invention can cancel the offset voltage with a simple circuit configuration and cope with a plurality of communication methods and communication modes, so that it can be used for communication-related infrastructures such as the broadcasting field and the communication field. be able to.

Claims

The scope of the claims

[1] a first local oscillator that outputs a pair of orthogonal local signals;

A first mixer that multiplies the received signal by the output signal of the first local oscillator to obtain orthogonal I and Q signals of the first intermediate frequency;

A first filter for band-limiting the I and Q signals of the first intermediate frequency;

A second local oscillator that outputs a pair of orthogonal local signals;

By multiplying the output signal of the first filter by the output signal of the second local oscillator, a second intermediate frequency I, Q signal having a frequency higher than the first intermediate frequency is obtained. 2 mixers,

A second filter for band-limiting the I and Q signals of the second intermediate frequency;

Decoding means for decoding the output signal of the second filter;

Receiver mode setting means for selecting the first intermediate frequency by controlling the first local transmitter based on a communication method;

A receiver comprising:

[2] The second filter is a plurality of filters having different pass bands or a channel selection filter having a filter force capable of variably setting the pass bands,

The receiver according to claim 1, wherein the receiver mode setting means switches a pass band of the second filter according to a value of the second intermediate frequency.

[3] The receiver mode setting means selects, as the first intermediate frequency, LIF (Low Intermediate Frequency) in the TDMA communication system and ZIF (Zero Inte ermediate Frequency) in the CDMA communication system. The receiver according to 1.

4. The receiver according to claim 1, wherein the second mixer detects carrier leak caused by quadrature modulation and removes a DC offset voltage by applying negative feedback to its input DC bias.

[5] A communication terminal comprising the receiver according to claim 1.