WO2004093315A1

WO2004093315A1 - Method of, and circuit for, broadening the frequency band of frequency dependent loads

Info

Publication number: WO2004093315A1
Application number: PCT/IB2004/001202
Authority: WO
Inventors: Kevin R. Boyle; Robert F. Milsom
Original assignee: Koninklijke Philips Electronics N.V.
Priority date: 2003-04-17
Filing date: 2004-04-08
Publication date: 2004-10-28
Also published as: GB0308913D0

Abstract

A method of, and circuit for, broadening the frequency band of a frequency dependent load such as an antenna (10) comprises coupling at least one bulk acoustic wave (BAW) resonator (12) to the frequency dependent load. The area of the at least one BAW resonator (12) is selected appropriately for the impedance bandwidth of the frequency dependent load (10). An inductive tuning means (14) tunes the at least one BAW resonator for a reactance slope and resistance symmetry about a centre frequency of the frequency dependent load. The inductive tuning means may be a series connected or parallel connected inductance.

Description

DESCRIPTION

METHOD OF, !D CIRCUIT FOR, BROADEMIMG THE FREQUENCY BAND OF FREQUENCY DEPENDENT LOADS

The present invention relates to a method of, and circuit for, broadening the frequency band of frequency dependent loads, for example, antennas operating in the frequency range of 500 MHz to 10 GHz.

Techniques for bandwidth broadening based on the use of resonant circuits are well known. However the Q of the resonant circuits must be sufficiently high that the total efficiency is not degraded.

"Electromechanical Transducers and Wave Filters " by W. P. Mason, published by van Nostrand 1948, discloses on pages 263 and 264, section 8.4, that it is possible to make wide band crystal filters by using inductors as well as crystals and capacitors. Attention is given to the effect of dissipation because the ratio of reactance to resistance of the best inductors mounted in what was at the time a reasonable space does not exceed 400. The effect of dissipation is twofold in that it may add a constant loss to the insertion loss characteristic of the filter and it may cause a loss varying with frequency in the transmitting band of the filter. Hence if the dissipation in the inductors needed to widen the band of the filter produces only an additive loss which is independent of the frequency, a result which may be satisfactory in many applications is obtained. Such a result is obtained if all the dissipation is concentrated on the ends of the filter section, for example by altering the terminating resistance to incorporate these resistances in the termination so that a resistance-less filter section terminated in its correct impedances is obtained.

US Patent Specification 5,604,507 discloses a broad-banded antenna having an antenna matching unit situated within a protective housing in the form of a metal shield. The matching network includes a toroidal inductor serially connected with the antenna such as to also create a parasitic capacitance with the metal shield. The resulting network is tuned. An antenna compensating network comprising a parallel resonance network increases the bandwidth of the antenna. The parallel resonance inductor is orientated so that the fields it generates are perpendicular to those of the antenna and the matching toroidal inductor to prevent coupling between the inductors. An optional series resonant network may enhance the compensating network with a capacitor and inductor connected in series to the antenna and each other. The fields of the series resonant inductor are perpendicular to those of the parallel resonance inductor. A series capacitor can be used with antennas operating at typical frequencies of 26 to 50 MHz. Shunt capacitors only are used at frequencies in the ranges 132 to 174 MHZ, 144 to 162 MHZ and 450 to 512 MHz.

A drawback of these prior proposals is that they are not appropriate for use with frequency dependent loads at higher frequencies in the range 500 MHz to 10GHz.

An object of the present invention is to widen the frequency band of a frequency dependent load in an effective manner.

According to one aspect of the present invention there is provided a method of widening the frequency band of a frequency dependent load, comprising selecting the area of at least one bulk acoustic wave (BAW) resonator to match the impedance bandwidth of the frequency dependent load, and inductively tuning the at least one BAW resonator for reactance slope and resistance symmetry about a centre frequency of the frequency dependent load.

According to a second aspect of the present invention there is provided a circuit comprising a frequency dependent load, at least one bulk acoustic wave (BAW) resonator coupled to the frequency dependent load, and inductive tuning means for inductively tuning the at least one BAW resonator. The present invention is based on the fact that BAW resonators retain their high Q characteristics so that their total efficiency is not degraded. Also it has been shown that using thin-film technology they can be made physically small but still capable of handling relatively high powers and high frequencies up to 10 GHz and their electrode area can be selected to provide the appropriate reactance for the required tuning. By way of comparison component technologies currently available in wireless communications devices have only a moderate Q. For example, discrete inductors and capacitors have maximum Qs of approximately 50 and 200, respectively. For a given physical size these values degrade with increases in inductance or capacitance making it difficult to design narrowband, high Q resonators. Although transmission lines do not have this limitation, they too have only moderate Q (with the substrates typically used) and occupy a relatively large area. Ceramic structures can have high Qs but have the disadvantage of occupying a considerable volume.

The BAW resonator may be coupled to the load either in parallel or in series. The BAW resonator and the load may be mounted on separate carriers and a well controlled interface may comprise an electrically conductive connection between a melted tiny solder ball having a cross sectional diameter of the order of 100μm on one of the carriers and a landing pad on a second carrier. This type of electrical connection is referred to in the art as "flip chip". The BAW resonator is preferably manufactured using aluminium nitride

(AIN) technology which is currently the piezoelectric material of choice for RF BAW resonators but there are alternatives, for example zinc oxide (ZnO).

The present invention will now be described, by way of example, with reference to the accompanying drawings, wherein:

Figure 1 is a block schematic diagram of an RF front end including an embodiment of the present invention,

Figure 2 is a simplified diagram of a flip-chip connection between a planar antenna and a BAW resonator, Figure 3 is a simplified cross sectional view through one implementation of a BAW resonator, Figure 4 is an equivalent circuit diagram of a BAW resonator and added reactive components,

Figure 5 is a Smith chart of a BAW series resonance,

Figure 6 is a Smith chart for one arrangement of a BAW resonator with shunt inductance,

Figure 7 is a Smith chart for a second arrangement of a BAW resonator with shunt inductance,

Figure 8 is a Smith chart of a PCS (Personal Digital Cellular System) PIFA (Planar Inverted-F Antenna) response, Figure 9 is a Smith chart of a PCS PIFA plus shunt connected, parallel

BAW resonator, and

Figure 10 illustrates the transmission characteristic of a PIFA plus a BAW resonator.

In the drawings the same reference numbers have been used to indicate corresponding features.

Referring to Figure 1 , a frequency dependent load in the form of an antenna 10, which may be a PIFA, is coupled to one terminal of a BAW resonator 12. An inductive tuning stage 14 is connected to the BAW resonator 12. A second terminal of the BAW resonator 12 is coupled additional RF circuitry 16. By way of example the additional RF circuitry 16 comprises a diplexer 18 having a first terminal coupled to another BAW resonator 12' with an inductive tuning stage 14'. A low noise amplifier (LNA) 19 is coupled to an output of the another BAW resonator 12'. A power amplifier 20 is coupled to a further BAW resonator 12" with inductive tuning stage 14". An output of the further BAW resonator 14" is coupled to the diplexer 18.

For convenience of description, the present invention has been described with reference to a circuit having one BAW resonator but it is to be understood that more than one BAW resonator is likely to be used. The BAW resonator 12 may be formed as part of an integrated RF module 22 which is connected to the antenna 10 by way of a controlled interface. In an alternative embodiment the antenna 10 is mounted on or carried by the same module 24 as the BAW resonator 12.

At the frequencies of interest, namely between 500 MHz and 10 GHz it is important that the antenna 10 and the BAW resonator 12 are designed together because any connections are an integral part of the design and must be well characterised. Ideally the connections are so small that their inductances at the frequencies of interest can be ignored.

Figure 2 illustrates diagrammatically a so-called "flip-chip" arrangement for providing a well controlled interface between the antenna 10 and the BAW resonator 12. The antenna 10 is a PIFA formed as a metallised layer 26 on one side of a substrate 28. On the other side of the substrate 28 there is provided an electrically conductive landing pad 30 which is connected through the substrate 28 to the PIFA. One or more small solder balls 32 having a size of the order of 100 microns (μm) is or are provided on the BAW resonator 12. The landing pad 30 is juxtaposed with the or at least one of the balls 32 and the solder is melted to provide a substantially non-inductive connection between the antenna 10 and the BAW resonator 12.

Figure 3 illustrates schematically a cross-sectional view through an implementation of a BAW resonator 12. The BAW resonator 12 comprises a substrate 34 on to which a reflection element 36 is typically sputtered. A first electrode 38 is provided on the reflection element 36. A piezoelectric layer 40 of AIN (aluminium nitride) is provided on the first electrode 38 and a second electrode 42 is provided on the layer 40. The BAW resonator 12 is formed by the area of overlap of the first and second electrodes 38, 42 so that the desired performance can be obtained by altering the area of the second electrode 42.

Figure 4 shows the equivalent circuit of the BAW resonator 12 which has been shown to accurately model its behaviour in the frequency range concerned. The equivalent circuit comprises first and second terminals 44, 46. A series resistance R_s representing the resistance of contacts plus electrodes (typically between 0.1 and 0.5Ω) has one end connected to the terminal 44 and three shunt arranged branches connected between its second end and the second terminal 46. The first shunt branch 48 comprises the series connection of a resonator effective motional inductance LR, given by

J_jR = , a resonator effective motional capacitance CR, given by

C_R ⁼ K²C_O ' ^{anc a rθSθnator} effective motional resistance RR, given by

, where ω_s is the angular frequency of the series resonance of ω_sC_RQ_R the resonator, K is the coupling coefficient of the resonator, typically about 10% lower than the coupling of the piezoelectric material (0.22 for AIN based resonator), and Q_R is the resonator quality factor (typically 500 to 1000). The second shunt branch 50 comprises the resonator static capacitance Co, related to the characteristic impedance by

^~ = ₁ - where Z₀ is the resonator characteristic ω_sz κ) impedance. The third shunt branch 52 comprises the substrate loss resistance R_p (typically 10 KΩ).

It can be seen that the values of these equivalent circuit components can be varied by way of the characteristic impedance, Z₀, which is, in turn, inversely proportional to the resonator area. In this way, the reactance slope, or bandwidth, of the resonator can be chosen as required to match a wide range of load impedances (with a wide range of bandwidths). However, the parallel resonance of a BAW resonator always occurs at a slightly higher frequency ω_p than the series resonance, given by Qy = o_s +K² ■ '^z°^r AIN, this means that typically the parallel resonance occurs at a frequency that is fractionally just 2.4% higher than the series resonance as shown by the asymmetry in Figure 5. Figure 5 is a Smith chart for a resonator designed for series resonance at 1950MHz, shown by the marker S2, which is the UMTS transmit centre frequency. Markers S1 and S3 show the transmit band edges of 1920 and 1980 MHz, respectively. The reactance slope increases with frequency within the band, as does the resistance, and therefore the loss in the band broadening circuit. The asymmetry shown in Figure 5 can be removed by the addition of a shunt inductor, L_p, across the terminals 44, 46 (Figure 4) to produce symmetry as shown in Figure 6 in which the markers S1 , S2, S3 have the same values as in Figure 5. In Figure 6, both the reactance slope and the resistance are substantially constant over the band of interest. The Q is greater than 350 over the band(Q_R = 500, Z₀ = 35Ω, f?_s-=0.1Ω, R_p = 10KΩ, /< = 0.22, L_p = 3.3nH with Q = 50). More importantly, the series resistance is approximately 2Ω over the band. This equates to a loss of less than 4% ( ~ 0.2dB) in a 50Ω system. Figure 7 shows the response with Z₀ = 10Ω, L_p = 1.2nH (all other parameters being the same as described with reference to Figure 6). In Figure 7 the markers S1 and S3, at frequencies of 1788 and 2009 MHz, respectively, correspond with the points at which the resistance becomes 2Ω. Based on this, the useable fraction of the bandwidth of the resonator is approximately 9%. This is of the order required to broaden the bandwidth of antennas for cellular radio bands such as GSM and PCS.

It is anticipated that the useful bandwidth of AIN based resonators will be in the range of approximately 0 to 15%.

Similar results may also be obtained for parallel resonance by using a inductor L_s (Figure 4) in series with the BAW resonator 12 (Figure 1). In this way, circuits with variable, high Q series and parallel resonators can be designed for bandwidth broadening purposes.

Referring to Figure 8 which shows an Sn plot for a PIFA designed for the PCS 1900MHz band. The markers sal, sa2 and sa3 correspond to the band edge and centre frequencies as follows: sal = 1850MHz, sa2 = 1920MHz and sa3 = 1990MHz.

The band-edge efficiencies and the return loss are:

All of the loss of efficiency is due to mismatch.

Referring to Figure 9, the chart relates to a PCS PIFA and a shunt connected, parallel BAW resonator (QR = 500, Z₀ = 100Ω, Rs = 0.5Ω, R_p = 10KΩ, K = 0.22, L_s = 7.5 nH with Q = 50). The markers sal , sa2, sa3 correspond to the frequencies described with reference to Figure 8.

The band-edge efficiencies and return loss are:

Comparing Figures 8 and 9 it is evident that the return loss is significantly improved. Also the overall efficiency is improved, indicating that the return loss is not simply improved at the expense of transmission loss.

Figure 10, which is a graph of efficiency in dB plotted against frequency in GHz, shows the transmission characteristics of an antenna and a BAW resonator. The marker m12 corresponds to a frequency of 1.920 GHz and an efficiency of -0.090dB. In the case of Figure 10 the antenna and the BAW resonator also acts as a filter that may provide sufficient selectivity for some applications.

By way of comparison Figure 10 shows in broken lines a type of characteristic which may be expected when bandwidth broadening is performed using typical inductances and capacitances. In the present specification and claims the word "a" or "an" preceding an element does not exclude the presence of a plurality of such elements. Further, the word "comprising" does not exclude the presence of other elements or steps than those listed.

From reading the present disclosure, other modifications will be apparent to persons skilled in the art. Such modifications may involve other features which are already known in the design, manufacture and use of radio transceivers and component parts therefor and which may be used instead of or in addition to features already described herein. Although claims have been formulated in this application to particular combinations of features, it should be understood that the scope of the disclosure of the present application also includes any novel feature or any novel combination of features disclosed herein either explicitly or implicitly or any generalisation thereof, whether or not it relates to the same invention as presently claimed in any claim and whether or not it mitigates any or all of the same technical problems as does the present invention. The applicants hereby give notice that new claims may be formulated to such features and/or combinations of such features during the prosecution of the present application or of any further application derived therefrom.

Claims

1. A method of widening the frequency band of a frequency dependent load, comprising matching the area of at least one bulk acoustic wave (BAW) resonator (12) to the impedance bandwidth of the frequency dependent load (10), and inductively tuning the at least one BAW resonator for reactance slope and resistance symmetry about a centre frequency of the frequency dependent load.

2. A method as claimed in claim 1 , characterised by balancing the impedance characteristic around the series resonance of the at least one BAW resonator using a shunt inductance (L_p).

3. A method as claimed in claim 1 , characterised by balancing the impedance characteristic around the parallel resonance of the at least one

BAW resonator using a series inductance (L_S).

4. A method as claimed in claim 1 , 2 or 3, characterised by controlling an interface between the at least one BAW resonator (12) and the frequency dependent load (10).

5. A circuit comprising a frequency dependent load (10), at least one bulk acoustic wave (BAW) resonator (12) coupled to the frequency dependent load, and inductive tuning means (14) for inductively tuning the at least one BAW resonator.

6. A circuit as claimed in claim 5, characterised in that the at least one BAW resonator is coupled in parallel with the frequency dependent load.

7. A circuit as claimed in claim 6, characterised in that the inductive tuning means comprises an inductance (L_s) connected in series with the at least one BAW resonator.

8. A circuit as claimed in claim 5, characterised in that the at least one BAW resonator is coupled in series with the frequency dependent load.

9. A circuit as claimed in claim 8, characterised in that the inductive tuning means comprises an inductance ( _p) connected in parallel with the at least one BAW resonator.

10. A circuit as claimed in any one of claims 5 to 9, characterised in that the at least one BAW resonator has aluminium nitride (AIN) as its piezoelectric material.

11. A circuit as claimed in any one of claims 5 to 10, characterised in that the frequency dependent load comprises an antenna.

12. An integrated module comprising a circuit as claimed in any one of claims 5 to 11.

13. A receiver including a circuit as claimed in any one of claims 5 to