WO2002071429A1

WO2002071429A1 - Electrical circuit for preventing an arc across an electrical contact

Info

Publication number: WO2002071429A1
Application number: PCT/EP2002/001071
Authority: WO
Inventors: Jürgen Bruck; Jörg SCHULTHEISS
Original assignee: Tyco Electronics Amp Gmbh
Priority date: 2001-03-01
Filing date: 2002-02-01
Publication date: 2002-09-12
Also published as: EP1366502A1; ATE342575T1; DE50208415D1; DE10290874D2; EP1366502B1

Abstract

The invention relates to an electrical circuit for preventing an arc across an electrical contact on opening the contact, whereby the electrical circuit comprises a timer which forces a time-delayed increase in the contact voltage in comparison with an unconnected contact. According to the invention, an electrical circuit for preventing an arc across an electrical contact, which very probably restricts the formation of an arc on opening said circuit contact and is furthermore economically produced and may be miniaturised may be achieved, whereby said electrical circuit comprises a transistor, wired in parallel to the circuit contact. Said transistor may be, for example, a power MOSFET, operated in the source circuit, or a Darlington transistor, formed by a bipolar transistor and a field effect transistor.

Description

Electrical circuit to avoid arcing over

an electrical contact

The present invention relates to an electrical circuit for avoiding an arc over an electrical contact when the contact is opened according to the preamble of patent claim 1.

If the current flowing in a circuit, in particular a circuit with an inductive load component, is interrupted by means of a mechanically moved contact, an arc can occur over the opening switch contact as a result of a current impression by the inductive component, via which the current flow is maintained at least for a short time. Such an arc can greatly reduce the service life of the switch contact or - at higher voltages than a stationary arc

- lead to the destruction of the contact. Depending on the contact distance and contact material, a certain minimum voltage is required to form such an arc. If this minimum voltage is undershot, no arc forms. Since the mechanical switch contact that opens only increases the contact distance comparatively slowly due to its finite movement speed, the voltage across the contact must fall below this minimum voltage at all times.

The following switches are particularly suitable for switching inductive loads:

- mechanical switches or relays with double or multiple contacts in series,

- mechanical switches or relays with magnetic arc extinguishing (e.g. blow magnet or arcing chamber),

- mechanical switches or relays with magnets and voltage-limiting electronic components,

- electronic switches.

When switching small loads with the help of a mechanical switch contact according to the state of the art, by inserting an RC element, the shutdown voltage voltage above the switching contact for the period of the contact opening below the voltage required for the ignition of an arc. After the contact has completely opened, the breakdown voltage required is approximately 600 V at a contact spacing of 0.2 mm, so that in general an arc can no longer be ignited. Such a conventional RC circuit is used for. B. in the monograph "Relay technology: basis and latest developments", Verlag Moderne Industrie, 1998, on page 57 under the chapter "Reduction of switching arcs" and is shown in Figure 18.

An alternative to the conventional RC circuit shown in FIG. 18 is the additional use of a voltage-limiting element (for example a Zener diode or a varistor) shown in FIG. 19, as a result of which the RC element is only a very small part of the Shutdown energy must take up and the main part is implemented by the Zener diode after reaching the Z voltage (80 V in the present example). The capacitance C can be designed correspondingly lower: in the present example, 1 μF is sufficient instead of the 1000 μF used in FIG. All other components shown in FIG. 19 have the same reference numerals and have the same value as the corresponding components in FIG. 18.

However, these circuits according to the prior art have the disadvantage that they are no longer practical at higher loads (such as occur, for example, in automotive applications) due to the necessary size of the capacitor and the high costs involved. In addition, when the contact is switched on again due to the discharge of the high capacitance via the low-resistance resistor, a high inrush current spike results, which in connection with the switch-on contact bouncing can lead to welding of the contact pieces.

It is therefore an object of the present invention to provide an electrical circuit for avoiding an arc over a switch contact, which effectively prevents the formation of an arc when the switch contact is opened and, moreover, is inexpensive to manufacture and largely miniaturizable. This object is achieved by an electrical circuit with the features of claim 1. Advantageous developments of the invention are the subject of several dependent claims.

The invention is based on the knowledge that the electronic device provided according to the invention in parallel with the electrical contact, even when switching off higher inductive loads, for example in direct current circuits with working voltages above 20 V, causes the voltage across the contact to rise suddenly to supercritical values after the opening of the Contact is prevented and thus an opening arc can be reliably avoided. The electrical circuit according to the invention also does not require its own power supply and is only connected to a switch or relay via two electrical connections. In contrast to other electronic circuits connected in parallel to the contact, the electric circuit according to the invention can also be used to prevent the arc on a changeover switch for braking an electric motor from being braked or in the event of a polarity reversal. A particular advantage of the electrical circuit according to the invention is the possibility of a very compact structure, which is of great advantage in automotive engineering, among other things. The service life of the mechanical contacts can be increased significantly, since only a relatively low mechanical wear of the contacts occurs. Finally, switching contacts with the electrical circuit according to the invention represent a particularly cost-effective solution, since single contacts with small contact thicknesses can be used, the dynamic properties of the switching contact do not have to be subjected to great demands, and other complex arc-quenching devices can be dispensed with entirely.

In order to be able to discharge the capacitance C in the circuit according to FIG. 1 as quickly as possible when the switch is closed again, and thereby to ensure good dynamic properties of the electronic device, according to an advantageous development, a discharge diode V1 is connected in parallel with the resistor R2 Cathode is connected to the capacitance C. The diode V1 simultaneously protects the gate of the amplifier V3 against negative gate-source voltages. Another resistor R1, which is connected between the capacitor C and the switch contact, limits the discharge current of the capacitor C and improves the dynamic properties of the circuit.

According to an advantageous embodiment, a voltage-limiting component V2 can be connected in parallel with the capacitor C to protect the amplifier.

A particularly simple and inexpensive solution for such a voltage-limiting component is a zener diode.

According to an alternative embodiment according to FIG. 2, a voltage-limiting component V4 can also be connected in parallel with the amplifier V3 as overvoltage protection for the amplifier when an inductive load is switched off.

Here, too, the use of a Zener diode is advantageous because it is a very inexpensive component.

According to an advantageous embodiment, a power MOSFET is used as amplifier V3. The advantage of this solution is that MOS field effect transistors can be controlled with little power and the overall structure can be greatly miniaturized. The MOSFET is operated, for example, in a source circuit, i. H. the source connection is connected to the connection of the resistor R102, which is connected to the switch contact, and the gate connection is connected to the common connection of capacitance C and resistor R102. The capacitance C is in the feedback branch between the drain and gate connection. This embodiment offers the advantage that the capacitance C can have a comparatively low value and nevertheless has the effect of a significantly larger capacitance (Miller effect).

If a Darlington transistor (FIG. 7) is used as the amplifier, which has a first and a second transistor, the voltage across the opening contact can advantageously be kept below the minimum voltage for arcing for a certain time, in order then to be sufficient large contact distance to quickly rise to the value appropriate for demagnetizing the load circuit. The time for which the voltage across the contact is at a constant Value kept below the minimum voltage is determined by the series connection of a resistor and a capacitor.

According to an advantageous embodiment, the Darlington transistor comprises two bipolar transistors. As a result, the voltage across the opening contact can be kept at the value of the base-emitter voltage of the Darlington transistor for a certain adjustable time and then quickly increased to the voltage required to demagnetize the load inductance. The adjustable time is determined by the charging of the capacity. During this time, the Darlington transistor conducts the current of the direct current circuit, first at the lower voltage level, then at a higher voltage level. The charging of the capacitance is essentially determined by the base-emitter voltage of the first transistor.

According to an alternative advantageous embodiment, the Darlington transistor comprises a field effect transistor as the first transistor and a bipolar transistor as the second transistor. As a result, the voltage of the opening contact can advantageously be kept at the value of the gate-source voltage, for example of the logic power MOSFET, and this can be increased again quickly after the adjustable time has elapsed. In this embodiment, the MOSFET carries the current of the DC circuit and the charging of the capacitance is determined by the gate-source voltage of the MOSFET. Another advantage of this solution is that the substrate diode of the MOSFET can take over the task of the freewheeling diode D1 according to FIG.

A voltage-limiting component can be connected in parallel with the output of the amplifier as overvoltage protection for the amplifier when an inductive load is switched off. A particularly simple and inexpensive solution for such a voltage-limiting component is a Zener diode. Their breakdown voltage should be far above the operating voltage of the direct current circuit in order to enable the inductive load circuit to commute quickly. Advantageously, the required Z-voltage can also be set by connecting a plurality of Z-diodes with a lower Z-voltage in series, so that the voltage-limiting element as a whole has a smaller differential resistance and also a lower temperature. receives the coefficient and can better dissipate a possible power loss through the division.

The base of the Darlington transistor is controlled in a particularly effective manner by a second amplifier T9, which has a line type complementary to the Darlington transistor.

In order to keep amplifier T9 saturated during the adjustable time, it is controlled by a third amplifier, transistor T10, which has the same conductivity type as the Darlington transistor.

In order to prevent large cross currents from occurring when a closing normally closed contact hits the circuit according to the invention in the reset state in the case of a relay application, a thyristor structure can be provided parallel to the input of the Darlington transistor.

The advantageous properties of the electrical circuit according to the invention can be used particularly effectively in an electromagnetic relay, the electrical circuit being connected in parallel with a normally open contact of the relay.

Likewise, the electrical circuit according to the invention can advantageously be used in an electrical connector to avoid an arc when the plug connection is released.

The invention is explained in more detail below with the aid of the embodiments shown in the accompanying drawings. Similar or corresponding details of the electrical circuit according to the invention are provided with the same reference symbols in the figures. Show it:

1 shows a switch contact with a DC voltage supply and an inductive load, to which an electrical circuit for avoiding an arc is connected in parallel in accordance with a first advantageous embodiment; Figure 2 shows the switch contact of Figure 1 with an electrical circuit to avoid an arc according to a second advantageous embodiment;

FIG. 3 shows the typical time profile of contact voltage and load current for a switching contact equipped with the electrical circuit according to the invention;

Figure 4 shows the application of the electrical circuit according to the invention for the braked shutdown of an electric motor;

Figure 5 shows the application of the electrical circuit according to the invention for motor pole circuits;

Figure 6 shows the application of the electrical circuit for connectors according to the invention;

FIG. 7 shows a circuit diagram of an electrical circuit for avoiding an arc according to a third embodiment;

Figure 8 is a circuit diagram of an electrical circuit to avoid an arc according to a fourth embodiment;

FIG. 9 shows an equivalent circuit diagram of a relay with an electrical circuit for avoiding an arc according to a fifth embodiment;

FIG. 10 shows a circuit diagram of a thyristor structure from an electrical circuit for avoiding an arc according to a sixth embodiment;

FIG. 11 shows a time course of the voltage across a make contact, the current through a load inductance and the collector current of a first transistor during a switching process in the circuit according to the invention in accordance with the third embodiment; FIG. 12 shows an enlarged detail from the time diagram in FIG. 11;

FIG. 13 shows a time course of a switching process when using an electrical circuit according to a fourth embodiment;

FIG. 14 shows a time diagram for the switching process in a circuit according to the fifth embodiment for an operating voltage of 60 V;

FIG. 15 shows a time diagram of a switching process in a circuit according to the fifth embodiment at an operating voltage of 42 V;

FIG. 16 shows a time diagram of a switching process in a circuit according to the fifth embodiment at an operating voltage of 24 V;

FIG. 17 shows a time diagram to explain the effect of a thyristor structure according to FIG. 9;

FIG. 18 shows a conventional RC contact circuit according to the prior art;

Figure 19 shows another conventional RC contact circuit according to the prior art.

Switching off inductive DC loads at voltages above 20 V may occur. a. in automotive engineering (24 V and 42 V on-board electrical systems).

Figure 1 shows schematically an equivalent circuit of such a circuit. L_Last denotes the load inductance, R_Last the ohmic component of the load resistance and U_L the DC voltage supply. The load circuit is to be interrupted by opening the switch contact 101. Without a suitable measure, a mechanical single contact would be destroyed in the existing loads by a very intense and possibly even permanently burning arc after a short operating time or immediately. Therefore, according to the invention, an electrical circuit for avoiding an arc over the contact, which is referred to below as circuit 100, is connected in parallel with the switching contact 101. According to a first and a second advantageous embodiment, which are shown in FIGS. 1 and 2, a sudden voltage rise at the switch contact 101 is prevented in order to avoid the arc and instead a linear voltage rise is forced.

In addition to a series circuit comprising a capacitor C and a resistor R2, which is connected in parallel with the switch contact 101, the circuit 100 according to the invention has an N-channel enhancement MOSFET V3, which is operated in the source circuit. The source and bulk connections are connected to one another and, like the resistor R2, are at the potential of the switch contact connection 1. The drain connection of the MOSFET V3 is connected to the switch contact connection 2. The gate connection is led to the connection point between the capacitance C and the resistor R2. When the switching contact 101 is opened, the voltage across the contacts increases abruptly; without wiring, an arc could ignite after reaching a material-dependent minimum voltage, which is typically in the range from 12 V to 16 V. By using the active circuit 100 in the first or second embodiment, it is achieved that this sudden increase from a certain threshold voltage changes to an approximately linear increase due to the activation of the MOSFET V3. The threshold voltage is given by the value of the resistor R1, which is connected between the capacitance C and the terminal 2 of the switching contact, and the gate-source threshold voltage of the MOSFET. The rate of rise of the voltage across contact 101 essentially depends on the values of capacitance C, resistor R2 and gate-source threshold voltage MOSFET.

By suitable dimensioning of the circuit 100, it can be achieved that the linear increase in the voltage across the contact 101 takes place so slowly during the opening phase of the contact 101 that the voltage is at all times safely below the minimum value required for the ignition of the arc. On the other hand, the circuit must be dimensioned in such a way that the contact voltage increases quickly enough to ensure that the maximum voltage and thus the highest possible resistance of the drain is within the contact changeover time, which is in the region of approximately half a millisecond. Source path at MOSFET V3 is reached. As a result, a shutdown with as little loss of energy as possible can be ensured. On the other hand, in the case of a changeover contact, a short circuit between see + U_L and -U_L prevented. The capacitance C therefore preferably has a value of 10 nF to 30 nF, the resistor R2 a value of approximately 1 kΩ and the resistor R1 a value of 2 Ω.

In the case of an inductive load caused by L_Last, a voltage surge can occur at contact 100, i.e. the voltage across the contact 100 rises to values above the operating voltage. By inserting a Zener diode V2, a limitation to the maximum voltage determined by the diode V2 and the gate-source threshold voltage of the MOSFET is achieved, avalanche operation of the MOSFET V3 is thus prevented. The loss of heat generated in the MOSFET when switching off can preferably be dissipated by thermal coupling with the mechanical structure of a switch or relay. If the switch contact 101 is closed, the capacitance C must very quickly, i. H. can be discharged within milliseconds so that rapid switching changes are possible. This is done via the resistor R1, which has a very small value in the range of approximately 2Ω, and the diode V1, which is polarized in the direction of flow during discharge. Resistor R1 also dampens the tendency of the circuit to oscillate.

According to a second advantageous embodiment, as shown in FIG. 2, instead of the Zener diode V2 as overvoltage protection for the MOSFET, a Zener diode V4 can also be connected between the drain and the source connection of the MOSFET. This additional suppressor diode is mainly used for particularly high inductive loads and, after reaching the maximum voltage, absorbs the main part of the remaining switch-off energy. The resulting heat loss should also be dissipated via a direct thermal coupling with the mechanical structure of the switch or relay.

FIG. 3 shows the typical time profile of contact voltage and load current when a switching contact is switched off, which is equipped with a circuit 100 according to the invention in accordance with the first or second embodiment. The load inductance L_Last has a value of 10 mH, the maximum load current is 10 A, the load voltage is 42 V. The maximum contact voltage is limited to 60 V, the time until the switch-off energy is fully implemented is approx. 3 milliseconds. An arc does not ignite. The circuitry 100 according to the invention can also be used if the movable center contact 1 does not remain open after the make contact has opened, but is switched to ground by subsequent contact with the break contact, as occurs, for example, when an electric motor 118 is switched off when braking. The corresponding structure is outlined in Figure 4.

FIG. 5 shows a polarity reversal circuit for an electric motor 118 as a further possibility of using the circuit 100 according to the invention. Such polarity reversal circuits are always required when the direction of rotation of an electric motor 118 is to be changed, such as in the motor vehicle sector with window regulators or the like. In the embodiment shown, two changeover relays 101A, 101B, each with a circuit 100A, 100B according to the invention, are connected between the center contact 1 and the make contact 2. The two normally open contacts 2 of the switching contacts 101A and 101B are connected to one another and are at + U_L. The break contacts 3 are also connected to one another and connected to ground. With the arrangement shown, a compact, arcing-free polarity reversal switching for the electric motor 118 can be implemented by means of a single contact.

The circuit 100 according to the present invention can, however, also be used for the arc-free disconnection of circuits in plug connectors. FIG. 6 shows such a plug connector 120, which comprises a plug part 122 and a socket part 124. To open a circuit, the contact pin 126 and the contact socket 128 must be separated from one another. In order to prevent the occurrence of an arc when the circuit is opened, an auxiliary contact 130 is provided in the socket part 124, and the circuit 100 according to the invention is connected between the auxiliary contact 130 and the contact socket 128. When opening, the load circuit between contact socket 128 and contact pin 126 is first interrupted. However, contact pin 126 still remains electrically connected to auxiliary contact 130. This closes the circuit between the contact pin 126, the auxiliary contact 130, the circuit 100 and the contact socket 128. The circuit 100 becomes active and takes over the controlled switching off of the load circuit. After the load circuit has been finally disconnected by the circuitry 100, the electrical connection between the contact pin 126 and the auxiliary contact 130 can be disconnected. The in Load circuit contained inductive energy is converted into heat. It is necessary to ensure that the time difference between the disconnection of the contact pin 126 from the contact socket 128 and the disconnection of the contact pin 126 from the auxiliary contact 130 is large enough to ensure that the load circuit is switched off by the circuitry 100. Depending on the pulling speed of the plug part 122, the distance between the contact socket 128 and the auxiliary contact 130 must be chosen to be sufficiently large.

FIG. 7 schematically shows a further equivalent circuit diagram of a circuit with a switch contact 101 and an inductive DC load. The load circuit is in turn to be interrupted by opening the switching contact 101. Without a suitable measure, as in the circuit diagram shown in FIG. 1, an arc would ignite when the switching contact 101 is opened. Therefore, in order to avoid an arc, an electrical circuit 100 according to a third advantageous embodiment is connected in parallel with the switch contact 101. In this and the embodiments shown in FIGS. 8 to 10, the formation of an arc is prevented in that the voltage across the contact 101 is initially kept constant at a low level and only increases to its final value when the contact is opened so far is that no more arc ignites.

According to this third embodiment, the electrical circuit 100 according to the invention has a Darlington transistor which is formed by the transistors T1 and T2. The base of this Darlington transistor is controlled by a transistor T9 with a conduction type complementary to the Darlington transistor. The transistor T9 in turn is controlled by a transistor T10 with the same conductivity type as the Darlington transistor in such a way that the transistor T9 is kept in saturation for an adjustable time. This adjustable time is determined by a timing element R10, C1, which is located in the emitter branch of the transistor T10. During this time, the Darlington transistor is fully conductive at the low voltage level. The capacitor C1 is charged only by the difference in the base voltages of the Darlington transistor on the one hand and the transistor T10 on the other. When the charging of the capacitor C1 has progressed so far that the collector current Ic of the transistor T10 drops and the transistor T9 can no longer be kept saturated, the Darlington transistor is increasingly blocked and the voltage above it is driven through the load inductance. The contact spacing should now be widened so that the subsequent rapid voltage rise can no longer lead to the ignition of an arc.

The phase of the almost constant low voltage across the Darlington transistor is now followed by a rapid rise in voltage until the breakdown voltage of the Z diode Z1 again controls the transistor T1 and a second phase of almost constant high voltage follows during which the inductive part of the Load circuit can quickly commutate. The voltage rise between the two phases is slightly delayed due to the parasitic collector base capacities of all transistors (Miller effect).

As is clear from the associated time profiles, which are shown in FIGS. 11 and 12, the voltage across the switch contact 101 rises to approx. 2 immediately after the switch contact 101 opens (assumed in the diagrams at t = 50 ms) to 5 V in order to bring the electrical circuit according to the invention, which in principle has no energy, before opening the switch contact 101 into the operating point in less than 1 μs (due to the Miller effect of the transistors T2, T9 and T10). This is followed by a voltage plateau, which is approx. 1.4 V. The voltage across the switch contact 101 remains at this level until the capacitor C1 is charged to the extent that the decaying collector current of the transistor T10 due to the difference voltage from the base-emitter voltage of T1 / 2 and the base-emitter voltage of T10 Transistor T9 can no longer maintain saturation. This is followed by a relatively steep rise in voltage, only slowed down by the Miller capacitors, until it is accepted by the Zener diode Z1. Since the respective collector base capacities decrease with increasing voltage, the voltage increase also takes place with increasing speed. At the high level determined by the Zener diode, the load inductance LL commutates and a damped oscillation follows, during which the parasitic energies are dissipated.

FIG. 8 shows a fourth embodiment in which the transistor T1 of the Darlington transistor is formed by a logic power MOSFET instead of a bipolar transistor, as shown in FIG. 7. The characteristic base-emitter voltage of the Darlington transistor (approx. 1.5 V) essentially goes into the gate-source Voltage at the operating point of the MOSFET above (approx. 3.5 V). Again, the transistor T9 is initially saturated and thereby essentially connects the drain connection and the gate connection of the MOSFET via the base-emitter path of the transistor T2 until the capacitance C1 is charged. This is followed by the voltage increase, which is slowed down by the Miller effect, up to the Z voltage. The breakdown voltage of the Zener diode should be far above the operating voltage of the DC circuit in order to enable the inductive load circuit to commutate quickly. As shown in FIG. 9, the required Z voltage can be set by connecting a plurality of Z diodes of smaller Z voltages in series, these having a smaller differential resistance and also a smaller temperature coefficient, and a better dissipation due to the division can dissipate.

In the left part of the circuit diagram in FIG. 9, an operating case is reproduced, as is often found in relay applications: a complex load is switched on via contact 101, the normally open contact or make contact. With the switches 102 to 108 a three times bouncing break contact (break contact) of a changeover switch (changeover) is simulated, which the load, for. B. shorts when braking an engine.

In this case, it is possible in principle that the closing normally closed contact hits a discharged circuit arrangement according to FIG. 7 or 8. This would result in very large cross currents via the normally closed contact and the circuit arrangement, since this normally tries to prevent an immediate rapid rise in voltage, also across the normally open contact 101 in this case. However, a larger current through the Darlington transistor or the MOSFET requires a higher control voltage at the base or at the gate. This higher control voltage is used to ignite a thyristor structure, which is formed from the transistors T3 and T4 in connection with the transistor T8, so that the current supplied by the transistor T9 due to the lower operating voltage of the thyristor (<1 V) compared to the normal control voltage of the Darlington transistor (> 1, 2 V) or the MOSFET (> 3.5 V) is derived. Therefore, the current drops in the output stage T1, T2 and the voltage, only slowed down by the Miller capacitors, can rise to a maximum of the Z voltage. A special design of the thyristor structure is shown in FIG. The ignition of this thyristor is made via the sum of the voltages of the transistors T6 and T7 connected as diodes as reference voltage in conjunction with the resistance R7 to the normal control Darlington transistor voltage adjusted. If a MOSFET is used instead of a bipolar transistor for the transistor T1, a higher reference voltage and therefore a Z-diode may also be required due to the higher control voltage.

In the example shown in FIG. 9, it is assumed that the switching contact 101 opens at t = 50 ms, that the switching contact 102 closes at t = 50.35 ms and that the switching contact 103 opens at t = 50.40 ms, that the switching contact 104 closes at t = 50.45 ms and switch contact 105 opens at t = 50.50 ms, switch contact 106 closes at t = 50.55 ms and switch contact 107 opens at t = 50.60 ms, and that switch contact 108 closes at t = 50.65 ms.

Another possible embodiment of the thyristor is shown in FIG. After the thyristor has been ignited, the current to be dissipated is conducted via the transistor T3, the Schottky diode DS and the transistor T4. The capacitors Cv1 and Cv2 serve to equalize the charge in the starting phase of the circuit arrangement immediately after the normally open contact 101 is opened. The diodes T6 and T7 again form the reference voltage and control the transistor T4 here.

FIG. 11 shows a complete demagnetization process: in the left part, the voltage above the make contact (curve 110) only remains at a low level, namely after the circuit arrangement has started, the base-emitter voltage of the Darlington transistor is less than 2 V. Im Time range after 50.05 ms you can see the rapid voltage rise to approx. 75 V, this value being essentially determined by the Z voltage. A voltage level then follows over time, during which the load inductance commutates. In the area of the commutation of the load inductance, the current I at the inductance (curve 112) decreases linearly to zero. This area is followed by a decaying oscillation in the course of the voltage across the normally open contact while the parasitic energy is being reduced.

A partial enlargement of the left part of the illustration from FIG. 11 is shown in FIG. Immediately after opening contact 101 (at time T = 50 ms), the collector current of transistor T1 increases (curve 114) while the voltage via the contact 101 after a slight voltage spike at the low level below 1.5 V, the base-emitter voltage of the Darlington transistor remains (see curve 110).

If a MOSFET is used instead of a bipolar transistor for transistor T1, as shown in FIG. 8, the time profile shown in FIG. 13 results. Curve 110 in turn means the voltage across switch contact 101, curve 112 the current through the load inductance and curve 114 the current that flows into the electrical circuit according to the invention. The plateau voltage in the low range here is approximately 5 V due to the use of the MOSFET. In the later course of time, the demagnetization curve is identical to the course shown in FIG.

FIGS. 14, 15 and 16 show the demagnetization curves for different operating voltages, namely 60 V, 42 V and 24 V. After the flight time of an armature of a relay, the normally closed contact of a changeover switch (changeover contact) closes and forces the circuit to a voltage which is below the Z voltage. The load inductance is also demagnetized during the bounce time of the normally closed contact (as can be seen from the curve 112), but the circuit arrangement follows the voltage changes, only delayed by the Miller effect of the semiconductors. The circuit arrangement is not reset until the capacitor C1 is discharged again. This takes place via the resistor R10 and the diode D2 when, for example, the normally open contact 101 closes. After reopening the normally open contact 101, the circuit arrangement reacts again in the manner shown above and delays the voltage rise.

FIG. 17 shows excerpts of the effect of the thyristor structure under changed load conditions. After the load inductance has been demagnetized for the first time, capacitor C1 is reset due to overshoot and freewheeling via diode D1. A subsequent closing of the normally closed contact (here at t = 50.35 ms) forces a steep current rise in transistor T1 (curve 114 in the left part of the illustration, to the left of the auxiliary line) until the thyristor ignites at a value of approx. 16 A. , This value can be set with the reference voltage. With a rapid current rise to approximately 2.5 A, curve 116 shows a takeover of the control current supplied by transistor T9 and, associated therewith, a rapid voltage rise at the emitter of the Transistor T1 (curve 110 to the right of the auxiliary line). The further increase in the current in the Darlington transistor up to approx. 20 A is due to the Miller effect of the rapidly increasing voltage now starting. At the same time, however, transistor T10 is also actively blocked, so that only the Miller effect of transistor T9 is effective. This can be seen from the decreasing current in the thyristor (see curve 116 falling below 1 A). In the following, a positive feedback becomes effective due to the decreasing operating voltage of the thyristor structure.

Of course, the embodiments of the electrical circuit 100 shown in FIGS. 7 to 10 can also be used for the applications shown in FIGS. 4 to 6.

Claims

claims

1. Electrical circuit for avoiding an arc over an electrical contact when the contact is opened, the electrical circuit (100) being connected in parallel with the electrical contact (101),

characterized,

that the electrical circuit (100) comprises at least one transistor (T1, T2; V3) and at least one timing element (C1; C, R102).

2. Electrical circuit according to claim 1, characterized in that a discharge diode (V1) is connected in parallel with the resistor (R102), the cathode of which is connected to the capacitance (C).

3. Electrical circuit according to claim 1 or 2, characterized in that the capacitance (C) is connected to the contact (101) via a further resistor (R101).

4. Electrical circuit according to one of claims 1 to 3, characterized in that a voltage-limiting component (V2) is connected in parallel with the capacitor (C).

5. Electrical circuit according to claim 4, characterized in that the voltage-limiting component (V2) is a Zener diode.

6. Electrical circuit according to one of claims 1 to 3, characterized in that a voltage-limiting component (V4) is connected in parallel with the transistor (V3).

7. Electrical circuit according to claim 6, characterized in that the voltage-limiting component (V4) is a Zener diode or suppressor diode.

8. Electrical circuit according to one of claims 1 to 7, characterized in that the transistor (V3) is a power MOSFET.

9. Electrical circuit according to claim 8, characterized in that the resistor (R102) and the capacitance (C) each have two connections and the Po- who-MOSFET (V3) is connected to a source connection to the connection of the resistor (R102) connected to the contact (101), to a drain connection to the connection of the capacitance (C) connected to the contact (100) is connected, and is connected to a gate connection with the common connection of the capacitance (C) and the resistor (R102).

10. Electrical circuit according to claim 1, characterized in that the transistor is a Darlington transistor having a first (T1) and a second transistor (T2).

11. Electrical circuit according to claim 10, characterized in that the second transistor (T2) is a bipolar transistor, the collector of the second transistor with the first terminal of the contact (101) and the emitter of the second transistor (T2) with the input of the first transistor (T1) is connected.

12. Electrical circuit according to claim 11, characterized in that the first transistor (T1) is a bipolar transistor, the collector of the first transistor (T1) with the collector of the second transistor (T2) and with the first terminal of the contact (101 ) is connected, the input of the first transistor (T1) is the base of the bipolar transistor and the emitter of the first transistor (T1) is connected to the second terminal of the contact (101).

13. Electrical circuit according to claim 11, characterized in that the first transistor (T1) is a field effect transistor (T1), the drain connection of the field effect transistor (T1) with the collector of the second transistor (T2) and with the first connection of Contact (101) is connected, the input of the first transistor is the gate terminal of the field effect transistor (T1) and the source terminal of the first transistor (T1) is connected to the second terminal of the contact (101).

14. Electrical circuit according to one of claims 10 to 13, characterized in that a voltage-limiting element (Z1) is connected in parallel with the second transistor (T2).

15. Electrical circuit according to claim 14, characterized in that the voltage-limiting element (Z1) is at least one Zener diode.

16. Electrical circuit according to one of claims 10 to 15, characterized in that the input of the amplifier (T1, T2) is connected via a second amplifier (T9) to the first connection of the contact (101).

17. Electrical circuit according to claim 16, characterized in that a third amplifier (T10) controls the second amplifier (T9).

18. Electrical circuit according to one of claims 1 to 17, characterized in that the electrical circuit further comprises a thyristor structure (T3, T4, T8) which is connected to the input of the Darlington arrangement (T1, T2).

19. Electromagnetic relay or switching contact with an electrical circuit according to one of claims 1 to 18, characterized in that the electrical circuit is connected in parallel to a normally open contact of the relay.

20. Electrical connector with an electrical circuit according to one of claims 1 to 19.