WO2002033842A2 - Architecture hybride d'interface à ligne directe dans un modem - Google Patents

Architecture hybride d'interface à ligne directe dans un modem Download PDF

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Publication number
WO2002033842A2
WO2002033842A2 PCT/US2001/031938 US0131938W WO0233842A2 WO 2002033842 A2 WO2002033842 A2 WO 2002033842A2 US 0131938 W US0131938 W US 0131938W WO 0233842 A2 WO0233842 A2 WO 0233842A2
Authority
WO
WIPO (PCT)
Prior art keywords
modem
common mode
mode interference
transmission line
balance network
Prior art date
Application number
PCT/US2001/031938
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English (en)
Other versions
WO2002033842A3 (fr
Inventor
Dongtai Liu
Original Assignee
Centillium Communications, Inc.
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Centillium Communications, Inc. filed Critical Centillium Communications, Inc.
Priority to AU2002211684A priority Critical patent/AU2002211684A1/en
Publication of WO2002033842A2 publication Critical patent/WO2002033842A2/fr
Publication of WO2002033842A3 publication Critical patent/WO2002033842A3/fr

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Classifications

    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/0264—Arrangements for coupling to transmission lines
    • H04L25/0272—Arrangements for coupling to multiple lines, e.g. for differential transmission
    • H04L25/0274—Arrangements for ensuring balanced coupling
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B3/00—Line transmission systems
    • H04B3/02—Details
    • H04B3/30—Reducing interference caused by unbalanced currents in a normally balanced line
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/10—Compensating for variations in line balance
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/0002—Modulated-carrier systems analog front ends; means for connecting modulators, demodulators or transceivers to a transmission line

Definitions

  • This invention relates generally to a hybrid transceiver architecture, and, in particular to a Direct-Line Interface (DLI) hybrid transceiver architecture for digital subscriber line (DSL) applications and other wire-based communication technologies.
  • DLI Direct-Line Interface
  • DSL Digital subscriber line
  • ADSL digital subscriber line
  • xDSL refers to a number of DSL-based technologies, including DSL, ADSL, symmetric DSL (SDSL), high bit rate DSL (HDSL), very high bit rate DSL (VDSL), and rate adaptive DSL (RADSL).
  • DSL technology requires a transceiver that can operate despite various types of natural interference (e.g. lightening, capacitive and inductive coupling of parallel lines, and so forth) and man-made interference (e.g., amplitude modulated radio-wave emissions, power lines, industrial equipment emissions, and so forth) on telephone lines. Interference is easily picked up and propagated by long telephone lines connecting a central office (CO) location (e.g., a telephone branch office) to a remote office (RO) location (e.g., a customer).
  • CO central office
  • RO remote office
  • CMI common mode interference
  • BALUN balanced-to-unbalanced
  • Transceiver design challenges include CMI rejection and BALUN conversion.
  • the DSL line pair usually presents a balanced appearance to the modem. Therefore, there is not a great need for a transformer to provide a BALUN conversion function.
  • the CMI rejection function of the transformer needs to be provided by any practical hybrid transceiver.
  • a DSL transformer typically provides 30-50 decibels (dB) of CMI rejection.
  • CMI is a significant problem in a DLI central office xDSL modern.
  • a telephone line picks up CMI signals as it runs parallel with power lines that contain high harmonics of 60 Hertz (e.g., as the result of non-continuous power control, by silicon-controlled- rectifier devices). Amplitude modulated radio and other communication signals are also within the ADSL signal frequency band. Thus, it is not uncommon for a telephone cable to pick up many volts of CMI.
  • FIG. 1 illustrates a circuit diagram for a conventional DSL modem.
  • Drivers 102 and 104 receive input signals 136 and 138, and induce output signals in the 1 :1 transformer 106.
  • Receivers 108 and 110 receive input signals from the coupling transformer 106, and produce received signals 140 and 142.
  • Resistances 1 12 and 1 14 provide the appropriate feedback gain for driver 102.
  • Resistances 116 and 1 18 provide the appropriate feedback gain for driver 104.
  • Resistances 120 and 122 each provide a 50 ohm resistance for each differential output to the coupling transformer 106.
  • Resistances 124, 126, and 128 maintain a balanced network for local echo cancellation, while providing appropriate gain for receiver 108.
  • Resistances 130, 132, and 134 provide a similar function for receiver 1 10.
  • Coupling transformer 106 interfaces the modem with the telephone line, and provides a satisfactory rejection of CMI by insulating any signal that flows in both input signals 136 and 138. What is needed is a transceiver having high common mode interference rejection without the use of a coupling transformer.
  • One embodiment of the present invention provides a method for suppressing common mode interference received by a modem interfaced to a transmission line without a coupling transformer.
  • the method includes providing a canceling signal that neutralizes the common mode interference across reactive components included in the modem.
  • Another embodiment of the present invention includes a modem having no coupling transformer to interface with a transmission line, wherein the modem is configured to suppress common mode interference received from the transmission line.
  • the modem includes a suppression means for neutralizing common mode interference across reactive components included in the modem.
  • FIG. 1 illustrates a circuit diagram of a conventional DSL modem.
  • FIG. 2 illustrates a circuit diagram of a transceiver architecture in accordance with one embodiment of the present invention.
  • FIG. 3 illustrates a circuit diagram of a transceiver architecture in accordance with another embodiment of the present invention.
  • FIG. 4 illustrates a variation of the transceiver architecture shown in FIG. 3 in accordance with another embodiment of the present invention.
  • FIG. 5 illustrates a circuit diagram of a transceiver architecture in accordance with another embodiment of the present invention.
  • FIG. 6 illustrates a system employing transceivers having architectures in accordance with one embodiment of the present invention.
  • FIG. 7 illustrates a method for suppressing common mode interference received by a modem interfaced to a transmission line without a coupling transformer in accordance with one embodiment of the present invention.
  • a Direct-Line Interface (DLI) hybrid transceiver in accordance with the present invention offers several advantages over the traditional, transformer-based hybrid design.
  • the transceiver CMI rejection is improved compared to that which is achieved with a conventional transceiver that includes a transformer;
  • the hybrid transceiver balance is superior to conventional transceivers, resulting in less CMI to differential mode interference (DMI) conversion, where a portion of CMI is converted into differential signal in the transceiver; the input signal distortion introduced from the non- linearity of the transformer is eliminated;
  • the space and weight of a modem transceiver having no coupling transformer is significantly reduced (e.g., 70% less weight and 25% less space);
  • each port of the modem line card can be implemented at lower cost (e.g., approximately one dollar for each port); and (6) there is about a reduction (e.g., 5%) in line driving power consumption.
  • both the analog front-end (AFE) input and the hybrid-balance network (HBN) are exposed to the CMI.
  • the CMI should not exceed a small fraction of the input signal range to prevent voltage shifting the input signal beyond the dynamic sampling range of the A/D converter.
  • a typical A/D converter has a dynamic sampling range that can accept an input signal range (swing) of +/- 0.8 volt centered at 1.6 volt.
  • the AFE's input signal range is nearly one volt, and that the CMI should not exceed 5% of the input signal range, then the CMI should be suppressed so that it does not exceed 50 millivolts (mV) at the AFE inputs.
  • the AFE's CMI-to-DMI rejection is very high compared to that of the HBN.
  • the HBN usually contains capacitors or inductors or a combination of such reactive components. Commercially available capacitors and inductors commonly have relatively large tolerances (e.g., +/-5% or 10%). At upstream (RO to CO) DSL frequencies, the reactive impedance supplies a significant portion of the total impedance of the HBN.
  • a HBN with an effective impedance variation of +1-2 % or higher is typical, due to the contribution from the tolerance (e.g., impedance variation) of the reactive components.
  • Such a HBN impedance variation produces about a significant CMI-to-DMI conversion factor (e.g., 4% or higher CMI-to-DMI conversion factor).
  • CMI-to-DMI conversion factor e.g., 4% or higher CMI-to-DMI conversion factor.
  • a 20 mA CMI current through 50 ohms would produce 40 V in DMI on the receiver inputs of the AFE.
  • 40 mV in DMI would degrade the signal-to-noise (S/N) ratio to approximately 0.1 dB for a CO to RO line separation of 15,000 feet.
  • S/N signal-to-noise
  • a practical S/N ratio should be at least 10 dB, and preferably should be higher. Thus, minimizing such DMI improves the S/N ratio of the transmission line.
  • the CMI on the AFE receiver inputs should be suppressed by at least 26 dB (e.g., the CMI voltage should be suppressed by a factor of 20), so that the AFE receiver inputs receive no more than 50 millivolts of CMI;
  • the CMI across the HBN's complex arm should be suppressed by at least 60 dB (e.g., the CMI voltage should be suppressed by a factor of 1000), so that S/N ratio due to converted CMI is greater than or equal to 100 dB.
  • One embodiment of the present invention provides a hybrid transceiver architecture that satisfies the above design goals.
  • This embodiment utilizes two CMI reduction schemes: one scheme employs a closed-loop suppression technique, and the second scheme employs an open-loop suppression method.
  • the hybrid transceiver architecture and associated performance parameters can vary depending on factors such as given design goals and desired system performance.
  • the present invention is not intended to be limited to any one particular embodiment or set of design goals. Rather, the techniques described herein can be employed in a number of applications and architectures having varying design goals.
  • FIG. 2 illustrates a circuit diagram of a transceiver architecture in accordance with one embodiment of the present invention.
  • Resistor 212 is a positive feedback resistor d for an active termination resistance.
  • Resistor 216 is connected to feedback resistor R ic 218 and ground 220.
  • Line driver 204 receives input signal 202 on the positive input terminal and a negative feedback signal on the negative input terminal from feedback resistor R fc 218.
  • the output of line driver 204 is coupled to te ⁇ nination resistor R D 214 and the negative input te ⁇ ninals of trans-conductance amplifiers 208 and 210.
  • the hybrid balance network includes impedances A 230 and 232 and impedances B 234 and 236, which are coupled to the input terminals of A/D converter 240.
  • the outputs of trans- conductance amplifiers 208 and 210, which comprise an open-loop CMI suppression circuit, are also coupled to the input terminals of A/D converter 240.
  • Trans-conductance amplifier 206 comprises the closed-loop CMI suppression circuit in conjunction with the line driver 204.
  • Trans-conductance amplifier 206 receives CMI 242 on the negative input terminal, and drives an output connected to resistors 218 and 216.
  • Trans-conductance 5 amplifiers 206, 208, and 210 also have the positive input terminal connected to ground 220.
  • Trans-conductance amplifier 206 forms a closed-loop suppression circuit in s conjunction with line driver 204 thereby suppressing the received CMI 242 to a lower level to prevent saturation of componentry included in the modem, such as the A/D converter included in the AFE.
  • Trans-conductance amplifier 206 receives CMI 242 and outputs a suppression signal that is injected into the feedback loop of the line driver. The common mode interference on the transmission line is accordingly reduced.
  • the desired o suppression ratio can be obtained by selecting the trans-conductance gain g m of trans- conductance amplifier 206.
  • the suppression can be computed by 20 Log [R fc (g m +1/R d )].
  • Step 1 may alternatively be skipped if CMI 242 associated with the transmission line is low enough that it will not cause undesirable conditions such as saturation.
  • Step2 5 Trans-conductance amplifiers 208 and 210 form an open-loop suppression circuit to reduce CMT-to-DMI conversion by injecting a canceling signal into a hybrid balance network (e.g., at the receiver inputs).
  • This canceling signal (actually comprised of two signals in this embodiment: one from amplifier 208 and one from 210) neutralizes CMI across the reactive impedances A 230 and 232. More specifically, the injected canceling 0 signal causes the CMI level at the receiver inputs to match the level of CMI 242 associated with the transmission line.
  • the trans-conductance gain g is selected such that the HBN impedances A 230 and 232 have no CMI potential across them. As such, the impedances A 230 and 232 are removed from the CMI output equation. Accordingly, the large CMI-to-DMI ratio due to component variation in impedances A 230 and 232 is eliminated.
  • the tolerances of g m , g,, and the resistor values largely determine how well these steps reduce CMI and DMI.
  • the impedances B 234 and 236 are resistors.
  • g m is a real trans-conductance.
  • the cancellation accuracy is proportional to the resistor impedance tolerance, which can be, for example, +/-1% or +/-0.1%.
  • the CMI suppression achieved can be in the same order as the resistor variation less several decibels (e.g., 30 to 50 dB), while the CMI-to-DMI conversion ratio above 60 dB.
  • FIG. 3 illustrates a circuit diagram of a transceiver architecture in accordance with another embodiment of the present invention.
  • Resistor 212 is a positive feedback resistor R d for an active termination resistance.
  • Resistor 216 is connected to feedback resistor R[ C 218 and ground 220.
  • Line driver 204 receives input signal 202 on the positive input terminal and a negative feedback signal on the negative input terminal from feedback resistor R fC 218.
  • the output of line driver 204 is coupled to termination resistor R 0 214 and the negative input terminals of trans-conductance amplifiers 208 and 210.
  • the hybrid balance network includes impedances A 230 and 232 and impedances B 234 and 236, which are coupled to the negative input terminals of operational amplifiers 224 and 226.
  • the operational amplifiers 224 and 226 have negative feedback resistors 222 and 228, respectively, connected to the corresponding negative input terminals.
  • the output signals of the operational amplifiers 224 and 226 are received by the input terminals of an A/D converter 240 (shown in FIG. 2).
  • trans-conductance amplifiers 208 and 210 which comprise the open-loop CMI suppression circuit, are also received by the input terminals of an A/D converter 240.
  • Trans-conductance amplifier 206 which comprises the closed-loop CMI suppression circuit in conjunction with line driver 204, receives CMI 242 on the negative input terminal.
  • the output of the trans-conductance amplifier 206 is coupled to resistors 218 and 216.
  • Trans-conductance amplifiers 206, 208, and 210 also have their positive input terminals connected to ground 220.
  • Operational amplifiers 224 and 226 each have their positive input terminal tied to CMI 242. The operational amplifiers 224 and 226 serve a number of purposes.
  • the operational amplifiers 224 and 226 can be used to provide a third step in CMI suppression.
  • the CMI level at the inputs (e.g., Vrxo) of the AFE is the same as the level of CMI 242 after it is suppressed in step 1.
  • additional CMI canceling signals are injected into the negative inputs of operational amplifiers 224 and 226 to further reduce or eliminate CMI at their outputs. These canceling signals are provided by the likes of amplifiers 208 and 210 thereby further reducing the common mode interference applied to an analog front-end of the modem.
  • another 20 dB of CMI reduction can be achieved at the inputs of the AFE, thereby further reducing the common mode interference applied to an analog front- end of the modem (e.g., CMI voltage less than ImV at inputs of AFE for a 20 mA CMI 242).
  • the gain stage provided by operational amplifiers 224 and 226 can be useful, for example, in helping to restore the S/N degradation sometimes associated with an active termination.
  • the closed-loop suppression circuit e.g., trans-conductance amplifier 206 and line driver 204
  • the open-loop suppression circuit e.g., trans- conductance amplifiers 208 and 210
  • the gain stage e.g., operation amplifiers 224 and 226
  • the closed-loop suppression circuit e.g., trans-conductance amplifier 206 and line driver 204
  • the open-loop suppression circuit e.g., trans- conductance amplifiers 208 and 210
  • the gain stage e.g., operation amplifiers 224 and 226
  • the closed-loop suppression circuit e.g., trans-conductance amplifier 206 and line driver 204
  • the open-loop suppression circuit e.g., trans- conductance amplifiers 208 and 210
  • the gain stage e.g., operation amplifiers 224 and 226
  • the closed-loop suppression circuit e.g., trans-conductance amplifiers 208 and 210
  • the gain stage e.g.
  • FIG. 4 illustrates a variation of the transceiver architecture shown in FIG. 3 in accordance with another embodiment of the present invention.
  • This particular embodiment allows the use of external discrete components and operational amplifiers.
  • Drivers 408 and 436 receive input signals 402 and 404, respectively, and have negative feedback resistors 410 and 434, respectively.
  • Operational amplifier 420 is connected to a positive feedback network of resistors 414, 416, and 418, and is also connected to a negative feedback network of resistors 422, 424, and 426.
  • Resistors 412 and 438 are termination resistors (e.g., 15 ohm).
  • the HBN impedance A is implemented by capacitors 450 and 454 and resistors 452, 456, and 458.
  • the HBN impedance B is implemented by capacitors 440 and 444 and resistors 442, 446, and 448.
  • Operational amplifier 466 has feedback resistor 464, and provides an input signal to A/D converter 240.
  • Operational amplifier 468 has feedback resistor 472, and provides a second input signal to A/D converter 240.
  • Resistors 462 and 470 connect the operational amplifier 466 and 468 feedback signals together.
  • the TIP input signal 478 is coupled to capacitor 480, and the RING input signal 476 is coupled to capacitor 474.
  • FIG. 5 illustrates a circuit diagram of a transceiver architecture in accordance with another embodiment of the present invention.
  • Resistor 212 is a positive feedback resistor R d for an active termination resistance.
  • Resistor 216 is connected to feedback resistor R fC 218 and ground 220.
  • Line driver 204 receives input signal 202 and a negative feedback signal from feedback resistor R t - C 218.
  • An output signal of line driver 204 is coupled to termination resistor R 0 214.
  • the HBN includes impedances A 230 and 232 and impedances B 234 and 236, which are coupled to the inputs of operational amplifiers 224 and 226.
  • the negative input terminals of operational amplifiers 224 and 226 have negative feedback resistors 222 and 228, respectively.
  • the operational amplifiers 224 and 226 produce output signals received by A/D converter 240 (shown in FIG. 2). Likewise, the output signals of trans-conductance amplifiers 208 and 210, which comprise the open- loop CMI suppression circuit, are received by A/D converter 240.
  • Trans-conductance amplifier 206 receives CMI 242, and drives an output connected to the negative input terminals of trans-conductance amplifiers 208 and 210.
  • Trans-conductance amplifiers 206, 208, and 210 also have one positive input terminal connected to ground 220.
  • Operational amplifiers 224 and 226 each have the positive input terminal tied to CMI 242. Unlike the previous architectures, the CMI suppression scheme does not include the line driver 204 thereby simplifying the complexity of implementation.
  • the first level of CMI suppression (reducing the level of CMI on the transmission line) is not perfo ⁇ ned.
  • the first level of suppression can be skipped.
  • CMI suppression is perfo ⁇ ned as described above in steps 2 and 3 at the inputs of the A/D converter 240 (sometimes referred to as the V RX O input terminals of the A/D converter). This is usually feasible in an active termination hybrid architecture having a low value for the termination resistor R 0 (e.g., 5 to 10 ohms).
  • this embodiment can alternative employ operational amplifiers 224 and 226 on the inputs of the A/D converter 240 to provide further CMI suppression.
  • the operational amplifiers 224 and 226 serve similar purposes that were previously discussed in reference to FIG. 3 (e.g., boosting the transceiver gain and providing an additional step in CMI suppression).
  • architectures I, II, and III include an open-loop CMI suppression circuit that reduces CMI across reactive components of the HBN to reduce the CMI-to-DMI conversion due to the HBN component variations (e.g., variations in the HBN capacitor and inductor values).
  • Architectures I and II further include a closed-loop suppression circuit that reduces the CMI on the transmission line (e.g., for applications having a high level of CMI on the transmission line).
  • Architectures II and III have an additional CMI suppression circuit that reduces CMI on the A/D converter input terminals.
  • Other architectures will be apparent in light of this disclosure.
  • another architecture might include an open-loop suppression circuit without any further suppression componentry (e.g., no closed-loop suppression circuit or gain stage for providing additional suppression).
  • FIG. 6 illustrates a system employing transceivers having architectures in accordance with one embodiment of the present invention.
  • the system includes a central office transceiver 602 connected via a transmission line to a remote office transceiver 604.
  • the system can employ, for example, DSL-based technology to facilitate communication between the two offices.
  • the central office typically routes data from the customer DSL-based modem to a DSL access multiplexer (DSLAM).
  • DSL access multiplexer DSL access multiplexer
  • Any number of transceiver architectures including those shown in FIG. 2, FIG. 3, FIG. 4, or FIG. 5, can be used to effect one or both of the transceivers 602 and 604.
  • the present invention is operationally compatible with conventional modems.
  • one of two communicating modems might have an architecture in accordance with an embodiment of the present invention while the other modem may have a conventional architecture.
  • FIG. 7 illustrates a method for suppressing common mode interference received by a transceiver interfaced to a transmission line without a coupling transformer in accordance with one embodiment of the present invention.
  • the steps of this method may be carried out together to effect a comprehensive CMI reduction scheme.
  • a step may be carried out individually without perfo ⁇ ning the other steps to effect a needed CMI reduction scheme.
  • any combination of the steps may be carried out as well. What steps are perfo ⁇ ned depends on factors such as the CMI level of the transmission line, the sensitivity of the transceiver componentry, and the desired level of CMI at the input of the transceiver's AFE.
  • the steps may be implemented, for example, by a combination of software, hardware, or fi ⁇ nware as illustrated in Figures 2-5.
  • the method begins injecting 705 a suppression signal into a feedback loop of a line driver of the modem thereby reducing common mode interference on the transmission line.
  • This step may be necessary where the CMI received from the transmission line is strong enough to cause modem componentry to saturate or otherwise malfunction.
  • This step can be carried out, for example, by the likes trans-conductance amplifier 206 and line driver 204 as illustrated in Figures 2 and 3.
  • the method includes injecting 710 a canceling signal into a hybrid balance network of the modem thereby neutralizing common mode interference across reactive components included in the hybrid balance network.
  • this step is employed individually without the other steps, and can be carried out, for example, by the likes trans-conductance amplifiers 208 and 210 as illustrated in Figures 2, 3, and 5.
  • the method may also include injecting 715 the canceling signal into the gain stage operatively coupled to the hybrid balance network and the transmission line thereby further reducing the common mode interference applied to an analog front-end of the modem.
  • this step is employed along with a combination of steps 705 and 710 to reduce the CMI at the AFE inputs of the transceiver to be less than 1 mV.
  • the gain stage of such an embodiment might be effected, for example, by the likes of operational amplifiers 224 and 228 as illustrated in Figures 3 and 5.
  • the loop configuration and associated parameters, and the noise level can be detected to help characterize the transmission line condition.
  • This is referred to as "line probing.”
  • Low frequency line probing signals may be used to detect the loop configuration for various lengths of telephone line loops. Lower frequency signals are especially helpful for long telephone line loops, as higher frequency signals transmitted on such telephone line loops are heavily attenuated. Therefore, it is not uncommon for line probing signal frequencies as low as several hundred Hertz to be used in modems.

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  • Engineering & Computer Science (AREA)
  • Computer Networks & Wireless Communication (AREA)
  • Signal Processing (AREA)
  • Power Engineering (AREA)
  • Cable Transmission Systems, Equalization Of Radio And Reduction Of Echo (AREA)
  • Noise Elimination (AREA)

Abstract

Dans le cadre de cette invention, un modem, qui se couple à une ligne de transmission sans transformateur de couplage, est capable de réduire le brouillage de mode commun émanant de la ligne de transmission. Les circuits de suppression permettent d'éliminer, de façon satisfaisante, ce brouillage de mode commun. L'invention porte également sur des procédés permettant de réduire et d'éliminer le brouillage de mode commun.
PCT/US2001/031938 2000-10-17 2001-10-12 Architecture hybride d'interface à ligne directe dans un modem WO2002033842A2 (fr)

Priority Applications (1)

Application Number Priority Date Filing Date Title
AU2002211684A AU2002211684A1 (en) 2000-10-17 2001-10-12 Transformerless interface for modem

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US24128100P 2000-10-17 2000-10-17
US60/241,281 2000-10-17
US77802801A 2001-02-05 2001-02-05
US09/778,028 2001-02-05

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Citations (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
GB2093314A (en) * 1981-02-17 1982-08-25 Western Electric Co Battery Feed Circuit
WO1998028886A2 (fr) * 1996-12-23 1998-07-02 Telefonaktiebolaget Lm Ericsson (Publ) Circuit d'extremite de ligne destine a reguler le niveau de tension de mode commun sur une ligne de transmission

Patent Citations (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
GB2093314A (en) * 1981-02-17 1982-08-25 Western Electric Co Battery Feed Circuit
WO1998028886A2 (fr) * 1996-12-23 1998-07-02 Telefonaktiebolaget Lm Ericsson (Publ) Circuit d'extremite de ligne destine a reguler le niveau de tension de mode commun sur une ligne de transmission

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AU2002211684A1 (en) 2002-04-29

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