WO2001047260A1 - Amelioration d'un recepteur augmentant la capacite d'information de systemes existants de transmission de communications - Google Patents

Amelioration d'un recepteur augmentant la capacite d'information de systemes existants de transmission de communications Download PDF

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Publication number
WO2001047260A1
WO2001047260A1 PCT/US2000/033479 US0033479W WO0147260A1 WO 2001047260 A1 WO2001047260 A1 WO 2001047260A1 US 0033479 W US0033479 W US 0033479W WO 0147260 A1 WO0147260 A1 WO 0147260A1
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WIPO (PCT)
Prior art keywords
spectrum
signal
phase
data
mixer
Prior art date
Application number
PCT/US2000/033479
Other languages
English (en)
Inventor
Walter S. Ciciora
Ted E. Hartson
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Encamera Sciences Corporation
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Filing date
Publication date
Application filed by Encamera Sciences Corporation filed Critical Encamera Sciences Corporation
Priority to JP2001547862A priority Critical patent/JP2003518839A/ja
Priority to CA002395810A priority patent/CA2395810A1/fr
Priority to EP00984146A priority patent/EP1249125A1/fr
Priority to AU20818/01A priority patent/AU2081801A/en
Priority to MXPA02006191A priority patent/MXPA02006191A/es
Publication of WO2001047260A1 publication Critical patent/WO2001047260A1/fr

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Classifications

    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L5/00—Arrangements affording multiple use of the transmission path
    • H04L5/02—Channels characterised by the type of signal
    • H04L5/12—Channels characterised by the type of signal the signals being represented by different phase modulations of a single carrier
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04N—PICTORIAL COMMUNICATION, e.g. TELEVISION
    • H04N11/00—Colour television systems
    • H04N11/24—High-definition television systems
    • H04N11/30—High-definition television systems with transmission of the extra information by means of quadrature modulation
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04N—PICTORIAL COMMUNICATION, e.g. TELEVISION
    • H04N21/00—Selective content distribution, e.g. interactive television or video on demand [VOD]
    • H04N21/40—Client devices specifically adapted for the reception of or interaction with content, e.g. set-top-box [STB]; Operations thereof
    • H04N21/43—Processing of content or additional data, e.g. demultiplexing additional data from a digital video stream; Elementary client operations, e.g. monitoring of home network or synchronising decoder's clock; Client middleware
    • H04N21/438—Interfacing the downstream path of the transmission network originating from a server, e.g. retrieving encoded video stream packets from an IP network
    • H04N21/4382—Demodulation or channel decoding, e.g. QPSK demodulation
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04N—PICTORIAL COMMUNICATION, e.g. TELEVISION
    • H04N5/00—Details of television systems
    • H04N5/44—Receiver circuitry for the reception of television signals according to analogue transmission standards
    • H04N5/455—Demodulation-circuits
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04N—PICTORIAL COMMUNICATION, e.g. TELEVISION
    • H04N7/00—Television systems
    • H04N7/08—Systems for the simultaneous or sequential transmission of more than one television signal, e.g. additional information signals, the signals occupying wholly or partially the same frequency band, e.g. by time division
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04N—PICTORIAL COMMUNICATION, e.g. TELEVISION
    • H04N5/00—Details of television systems
    • H04N5/44—Receiver circuitry for the reception of television signals according to analogue transmission standards
    • H04N5/4446—IF amplifier circuits specially adapted for B&W TV

Definitions

  • the present invention relates to communications systems and methods for transmitting additional information via television transmissions and other transmissions.
  • the invention applies to a wide range of communication transmissions, this disclosure focuses, although not in a limiting way, largely on the applicability to television.
  • NTSC The standard method for over-the-air transmission of television signals in the United States in called NTSC.
  • NTSC The standard method for over-the-air transmission of television signals in the United States in called NTSC.
  • This is an analog system in which the picture is transmitted in a vestigial sideband modulation format on the visual carrier and the sound component transmitted as frequency modulation on a separate sound carrier.
  • HDTV High Definition Television
  • Great progress has been made in the area of digital TV bandwidth compression so that one or more HDTV signals can be conveyed in the standard television bandwidth of 6 MHz.
  • These HDTV developments have, by a combination of techniques, substantially reduced the bandwidth required for fully digital transmission of the video information.
  • the standard NTSC format allocates 6 MHz of spectrum to the transmission of the combined video and audio signals.
  • the visual carrier is placed 1.25 MHz above the lower band edge and the aural carrier 5.75 MHz above the lower band edge.
  • the visual information is impressed on the visual carrier using a vestigial sideband amplitude modulation (AM) technique so that the frequency components below the visual carrier occupy no more than the 1.25 MHz of the available spectral assignment, while the frequency range allocated to the visual information extends for approximately 4.2 MHz above the visual carrier.
  • AM vestigial sideband amplitude modulation
  • the color information is carried on the color subcarrier of the main visual carrier at approximately 3.58 MHz above the visual carrier (4.83 MHz above the lower band edge).
  • the modulation of the color information is both in-phase and quadrature and contains more lower sideband components than upper.
  • the aural information is carried on the separate aural carrier at 4.5 MHz above the visual carrier (5.75 MHz above the lower band edge) and is frequency modulated (FM) with a peak deviation of 25 KHz over the range of audio frequencies extending to somewhat above 15 KHz.
  • the amplitude modulation of the visual luminance information is solely an in-phase variation with no quadrature component prior to the vestigial filter while, 2) the aural subcarrier is solely frequency modulated. Subcarriers are sometimes added to the aural carrier but they too are frequency modulated on the main aural carrier. It can be seen then that not all of the information carrying capacity in the 6 MHz analog channel is occupied. There are no quadrature components in the region close to the visual carrier and no amplitude modulation components in the aural carrier region.
  • WavePhore, Inc. utilizes a signal which puts single sideband phase shift data in the area of approximately 3.9 to 4.2 MHz above the visual carrier (5.15 MHz to 5.45 MHz above the lower band edge) and is capable of transmitting in the order of 500 Kbits / second while causing only minor interference to the analog television signal. This system is covered under U.S. patent(s).
  • D-Channel Another approach of transmitting data within the NTSC broadcast format has been developed by Digideck, Inc. This technique is called the D-Channel and operates at a reduced level in the lower frequency portion of the video vestigial sideband is capable of transmission of something in the order of 750 Kbits per second with only minor interference to the analog television signal and is covered under U.S. patent(s).
  • Patent application 09/062,225 filed April 17, 1998.
  • the improvements of the present invention relax the need for higher performance filters in the data receiver of the invention of U.S. Patent application 09/062.225 filed April 17, 1998.
  • the present invention improves the television receiver and more specifically the data receiver of the visual portion of the signal of U.S. Patent application 09/062,225 filed April 17, 1998. While the reception of the aural portion of the signal of U.S. Patent application 09/062,225 filed April 17, 1998 is not further discussed here, it remains an important part of Ihe total system according to a preferred embodiment. It is not further discussed here because the present invention does not change its implementations.
  • the invention of U.S. Patent application 09/062,225 filed April 17, 1998 includes a Compensator Subsystem which adjusts the transmitted data spectrum so that when passed through the Nyquist filter of an existing television receiver, the data spectrum will become symmetrical about the television signal's visual carrier.
  • the data signal will be in quadrature with the visual signal to the extent that the Compensator Subsystem accurately compensates for the effects of the Nyquist filter.
  • the visual signal will have both in-phase and quadrature components because the television receiver's Nyquist filter has made the visual spectrum unsymmetrical about its carrier.
  • An ideal synchronous detector that is phase locked to the visual carrier in the television receiver will only respond to the in-phase components of the visual signal and will thus ignore the quadrature data signal as well as the quadrature components of the visual signal.
  • the data receiver of the invention of U.S. Patent application 09/062,225 filed April 17, 1998 should ideally perform in a complementary manner. That is, the data receiver will not have a Nyquist filter. Thus the visual spectrum in the double sideband region will remain symmetrical about its carrier and thus will not have any quadrature components. The data spectrum, however, will have both in-phase and quadrature components due to the action of the Compensator Subsystem in the data transmitter.
  • An ideal synchronous detector that is phase locked to ninety degrees relative to the visual carrier would respond only to quadrature data components and ignore the visual signal in the double sideband region which is completely in-phase with the visual carrier as well as the in-phase components of the data signal.
  • the visual signal has quadrature components outside the double sideband region. Consequently, the data signal demodulated by the data detector will be contaminated with the quadrature components of the visual signal.
  • the quadrature components must be strongly attenuated by a filter to prevent their detection by the data synchronous detector. Ideally, this filtering would eliminate the visual quadrature components. But practical filters will not accomplish this.
  • Such detection of the quadrature components of the visual signal by the data detector would result in "eye closure" of the data signal. As is appreciated by those of ordinary skill in the data detection arts, this will reduce the overall system performance margin and lead to data detection difficulties.
  • the present invention alleviates these problems by synthesizing a double side band signal in such a manner that both the visual signal and the d ⁇ La signal are each symmetrical with respect to their carriers in both amplitude and phase. This allows synchronous detection techniques to be applied to cleanly separate the data signal from the visual signal.
  • the double sideband synthesis is accomplished by reversing the received spectrum and adding it back to itself. Reversing the received spectrum interchanges the upper sideband and the lower sideband in a manner that causes the lowest frequency of the lower sideband in the received spectrum to become the highest frequency of the upper sideband in the reversed spectrum. Likewise, the highest frequency of the upper sideband in the received spectrum will become the lowest frequency of the lower sideband in the reversed spectrum.
  • One method of implementing spectrum reversal is presented in the preferred embodiment. Other methods of spectrum reversal can be implanted by those of ordinary skill in these arts.
  • While the main focus of this invention is the improved recovery of data signals, the invention has benefits for television receiver design even for signals that do not include data. Also, this invention will improve the reception of data signals that are phase modulated onto the visual carrier.
  • FIGURE la is a graph of a television signal spectrum normalized to 0.0 Hz.
  • FIGURE lb is a graph of the output of a flat response television receiver.
  • FIGURE lc is a graph of an idealized and typical TV receiver response curve.
  • Figure 2a is a graph of Data & Video Modulator Signals.
  • Figure 2b is a graph of Data Receiver Signals.
  • Figure 2c is a graph of television Receiver Signals.
  • Figure 2d is a block diagram of Data Detection receiver using a Filter. The receiver is subject to "Rude Video" interference.
  • Figure 3a is a graph of the Data Spectrum subjected to Data RF Tx Filter.
  • Figure 3b is a graph of the Data Spectrum subjected to TV Receiver's Nyquist Filter. This yields a Q component only (note: in the receiver the spectrum is reversed at IF).
  • Figure 4a is a block diagram of the Data Demodulator and an optional NTSC demodulator using the present Synthesis of a Double Sideband Signal invention.
  • Figure 4b is a graph of the NTSC spectrum and the Data Spectrum after the tuner and before Nyquist filter in a conventional television receiver.
  • the spectra is shown “as is " in the receiver; i.e. opposite to at RF).
  • the two spectra are shown separately for purposes of illustration. In the actual system, the two spectra are combined in a manner that does not allow them to be viewed separately.
  • Figure 4c is a graph of the TV Receiver's Nyquist filter characteristic.
  • Figure 4d is a graph of the NTSC spectrum and the Data Spectrum after the tuner, Nyquist filter, and Precision Phase-Correct Delay.
  • Figure 4e is a graph of the NTSC spectrum and the Data Spectrum at output of Spectrum Reverser.
  • Figure 4f is a graph of the NTSC spectrum and the Data Spectrum at output of Summer.
  • Figure 5 a block diagram of the Spectrum Reverser.
  • Figure 5b is a graph of the IF spectrum as seen at IF frequencies (the reverse is seen at RF frequencies.)
  • Figure 6a is a block diagram of a Precision Phase-Correct Delay Example using exactly the same filters as used in the Spectrum Reverser.
  • Figure 6b is a graph of the IF spectrum as seen at IF frequencies (the reverse is seen at RF frequencies.)
  • Figure 5e is a block diagram of the Spectrum Reverser Without filters.
  • Figure 5f is a graph of the IF spectrum as seen at IF frequencies (the reverse is seen at RF frequencies.)
  • Figure 6e Precision Phase-Correct Delay Without Filters.
  • Figure 6f is a graph of the IF spectrum as seen at IF frequencies (the reverse is seen at RF frequencies.)
  • Figure 6i is a graph of the spectrum resulting from adding the spectra of Figs 5h and 6h.
  • Figure 6j is a graph of the spectrum of a cosine wave at the IF frequency.
  • Figure 6k is a graph of the spectrum of Figure 6i heterodyned by a cosine wave at the IF frequency and a low pass filter to receive the baseband signal.
  • Figure 7a is a graph of the television spectrum and the data spectrum at the output of a Nyquist Filter and a Precision Phase-Correct Delay as see at IF (at RF, spectrum is reversed).
  • Figure 7b is a graph of the television spectrum and the data spectrum at the output of the Spectrum Reverser.
  • Figure 7c is a graph of the sum of the television spectrum and the data spectrum found at the output of Precision Phase-Correct Delay and at the output of the Spectrum Reverser.
  • Figure 8a is a block diagram of the Data Demodulator and an optional NTSC demodulator using the present Double Sideband Synthesis invention but without a Nyquist Filter. This Figure is similar to Figure 4a.
  • Figure 8b is a graph of the television spectrum and the data spectrum at the output of a Precision Phase-Correct Delay as see at IF (at RF, spectrum is reversed) but without first passing through a Nyquist filter.
  • Figure 8c is a graph of the television spectrum and the data spectrum at the output of the Spectrum Reverser, but without first passing through a Nyquist filter.
  • Figure 8d is a graph of the sum of the television spectrum and the data spectrum found at the output of Precision Phase-Correct Delay and at the output of the Spectrum Reverser, but without first passing through a Nyquist filter.
  • Figure 9a is a graph of the Data Spectrum after the tuner and Precision Phase- Correct Delay but without first passing through a Nyquist filter. Note: This spectrum has both Q components and I because it is not symmetrical about the carrier. (Spectra shown "as is” in the receiver; i.e. opposite to at RF).
  • Figure 9b is a graph of the Data Spectrum at output of Spectrum Reverser, but without first passing through a Nyquist filter.
  • Figure 9c is a graph of the Data Spectrum at output of the Summer. This spectrum has only a Q component because it has been made symmetrical about the carrier frequency. The delayed and reversed data spectra prior to summation are shown to illustrate that they add to the summer output spectra.
  • Figure 10a is a block diagram of the Data Demodulator of the receiver using the present Double Sideband Synthesis invention. Note that this figure is identical to Figure 8a.
  • Figure 10b is a graph of the television spectrum and an expanded bandwidth data spectrum at the output of a Precision Phase-Correct Delay as see at IF (at RF, spectrum is reversed) but without first passing through a Nyquist filter.
  • Figure 10c is a graph of the television spectrum and the expanded bandwidth data spectrum at the output of the Spectrum Reverser, but without first passing through a Nyquist filter.
  • Figure l Od is a graph of the sum of the television spectrum and the expanded bandwidth data spectrum found at the output of Precision Phase-Correct Delay and at the output of the Spectrum Reverser, but without first passing through a Nyquist filter.
  • Figure 1 1 a is a graph of the amplitude transfer function of the Data RF Tx Filter found at the transmitter.
  • Figure 1 l b is a graph of the amplitude transfer function of the NTSC Vestigial Sideband Filter found at the transmitter.
  • Figure l ie is a graph of the amplitude transfer function of the Composite
  • Figure 1 1 d is a graph of an expanded bandwidth Data Spectrum extending to 1.25 MHz and the Composite Filter function of Figure l i e along with the resulting non symmetric Filtered Data Spectrum.
  • Figure 12a is a graph of the application of the present Double Sideband Synthesis invention to receive the expanded bandwidth Data Spectrum of Figure 1 Id.
  • the present invention yields perfect reconstruction of the data spectrum between + 0.75 MHz and modest distortion between + 0.75 MHz and +1.25 MHz. This modest distortion could be pre-distorted at the transmitter.
  • Figure 12b is a graph of the expanded data spectrum after the television receiver ' s Nyquist Filter.
  • the expanded bandwidth Data spectrum has been made symmetrical and has been reduced in bandwidth to the double sideband bandwidth of the NTSC signal.
  • the data spectrum is all in quadrature with the visual signal and a Synchronous detector in the television receiver will be blind to this data signal.
  • Figure la describes the standard analog NTSC television signal comprising a
  • Video Carrier 102 that is Amplitude Modulated (AM), a Sound Carrier 103 that is Frequency Modulated (FM) and a Color Carrier 105, which is modulated with an in- phase and a quadrature phase components.
  • the lower sideband of the television signal is unattenuated to a frequency 750 KHz below the visual carrier.
  • the signal is severely attenuated at frequencies more than 1.25 MHz below the visual carrier.
  • the signal is rolled off between 750 KHz and 1.25 MHz below the visual carrier in a manner that can be realized with practical filters.
  • This filtering results in a vestige of the lower sideband leading to the term "Vestigial Side Band" (VSB) modulation.
  • the VSB technique was motivated by the desire to provide increased picture resolution while remaining within the allocated 6 MHz frequency band for a television channel.
  • the details of the VSB implementation were governed by practical consideration for the economic realization of the required filters.
  • Figure lb illustrates the difficulty this VSB technique causes in the demodulation process. Because the region 104 of Figure l a between 750 KHz below the visual carrier and 750 KHz above the visual carrier is ordinary double sideband modulation and the region 106 in Figure la between 1.25 above the visual carrier and 4.0 MHz above the visual carrier is single sideband modulation, the demodulated signal will have twice the strength in the baseband frequencies which are less than 750 KHz compared to the signals in the baseband frequencies between 1.25 MHz and 4.0 MHz.
  • Figure lc illustrates the solution adopted in analog television receivers to deal with this problem.
  • the signal is weighted with a filter that is antisymmetric about the visual carrier and results in half strength passed through the filter at the visual carrier frequency 102. This normalizes the strength of the demodulated signal from zero baseband frequency to its highest frequency.
  • This type of filter, with an antisymmetric characteristic, is called a Nyquist filter.
  • Figure 2a describes the data and video modulator signals at the signal origination point.
  • Figure 2b Illustrates the processing of the signals in a data receiver of the invention of U.S. Patent application 09/062225 filed April 17, 1998 while Figure 2c shows the processing of these signals in an ordinary television receiver.
  • a conventional double sideband signal has only in-phase components. It has no quadrature components. Also, a signal which is not a conventional double sideband structure has both in-phase and quadrature components.
  • the visual signal of Figure la between 750 KHz below the visual carrier and 750 KHz above the visual carrier is a conventional double sideband signal and has only in-phase components.
  • the signal with frequencies higher than 750 KHz above the visual carrier is not a conventional double sideband signal and thus has both in-phase and quadrature components. This is illustrated in the center section of Figure 2a.
  • the left part of the top section of Figure 2a shows the VSB filter characteristic 202 that converts a conventional double sideband television signal into the VSB signal 204 of the center section of Figure 2a.
  • the top section of Figure 2c shows the television receiver ' s Nyquist filter characteristic 206 and the center section of Figure 2c shows the resulting television signal spectrum 208 after passing through the television receiver's Nyquist filter 206.
  • the left portion of the bottom section of Figure 2a shows a data signal 210 double sideband modulated onto a quadrature phase shifted visual carrier 212 and the quadrature phase shifted carrier then completely suppressed.
  • This data signal 210 has only quadrature components; no components of this data signal exist in-phase with the television signal's visual carrier 214. If this signal was to passthrough the Nyquist filter 208 of the center section of Figure 2c, its double sideband nature would be destroyed resulting in the creation of components in-phase with the visual carrier 214 of the television signal. These components would become visible as interference on the television screen. This is undesirable and is likely to be unacceptable unless the interference is reduced to such a low level as to become invisible to viewers under normal circumstances.
  • the data signal is predistorted in a manner such that passage through the receiver's Nyquist filter 206 will convert it back into a conventional double sideband signal 216 with only quadrature components. This signal will then be ignored by the television receiver's synchronous demodulator.
  • Data RF Tx Filter Data Radio Frequency Transmit Filter 218
  • This filter characteristic 218 has the shape of the Nyquist filter used in the receiver 206 but with the frequencies reversed. This causes the transmitted data signal spectrum 220 to appear as in the bottom section of Figure 2a. That spectrum has both in-phase and quadrature components.
  • the data demodulation process of the prior art is shown in Figure 2b.
  • Synchronous demodulation is employed to separate the quadrature data signal from the in-phase visual signal.
  • the visual signal is conventional double sideband only in the range between 750 KHz below the visual carrier and 750 KHz above the visual carrier. Outside that range, the signal will have components 222 in quadrature which will be demodulated by the data signal synchronous demodulator 224 and will interfere with the detection of data; i.e. the discrimination of the discrete levels of the data signal at baseband after demodulation.
  • the filtering process were to be made completely symmetrical by the use of a reversed VSB filter in the data receiver, the video would have only components that are in-phase with the visual carrier and would not be detected. Snce the VSB filter 202 is not precisely specified, there would be some difficulty in choosing a proper reversed VSB filter design.
  • the relative strength of the visual signal relative to the data signal makes the precision of this choice important.
  • the data signal 216 is of much lower strength than the visual signal 208, the degree of precision in matching the transmitting data filter 218 (which ideally is the reverse of the receiver ' s Nyquist filter 206) is much less critical. The situation further benefits from the fact that the data signal 210 is uncorreiated to the visual signal 204 and imperfections will appear as noise rather than as some annoying pattern or image.
  • Figure 3a shows a data double sideband spectrum 302 (of raised cosine shape, chosen for illustrative purposes only and not as a limitation).
  • a data transmit filter characteristic 304 shape is shown (as a linear filter, chosen for illustrative purposes only and not as a limitation).
  • the resulting transmitted data signal 310 has been made unsymmetrical and therefore will have both in-phase and quadrature components.
  • Figure 3b shows the transmitted data signal 308 passing through the television receiver's Nyquist filter 306 and being converted once again into a symmetrical spectrum shape 312 that will have only spectral components that are in quadrature to the visual signal.
  • a synchronous demodulator in the television receiver will not respond to the data signal. Since the data signal is of much lower strength, any inaccuracies in converting it into a symmetrical spectrum will result in relatively minor interference, not visible under ordinary viewing condition.
  • FIG 4 shows one embodiment of the principle of the current invention which is called "Data Separation from Video by Synthesis of a Double Sideband Signal” and synchronous demodulation.
  • the invention is installed in a device that is to optionally receive television signals.
  • the TV receiver has a Nyquist Filter 406 whose output is split into three paths.
  • One path 402 feeds a mechanism for recovering the carrier signal.
  • a phase locked loop 404 is shown here for illustrative purposes but not as a limitation.
  • the recovered carrier signal is used in a mixer 408 for synchronous demodulation of the visual signal. It is also phase shifted 410 for use in another mixer 412 for synchronous demodulation of the data signal.
  • the second path 414 feeds a Spectrum Reverser block 416.
  • FIG. 4b shows the received television 422 and data signals 424 prior to Nyquist filtering 426.
  • the television signal 422 is shown as an NTSC signal for illustrative purposes but not as a limitation.
  • the television signal 422 and the data signal 424 are shown as two separate signals for purposes of illustration.
  • Figure 4c shows the television receiver's Nyquist filter 426.
  • Figure 4d shows the television signal 428 and the data signal 430 after the television receiver's Nyquist filter 426.
  • the data signal 430 has become symmetrical in its frequency range and thus has only quadrature components.
  • the television signal 448 in this same frequency range has become unsymmetrical having both in-phase and quadrature components.
  • Figure 4e shows the output 432, 434 of the Spectrum Reverser 416 that flips the spectrum around the visual carrier frequency 436.
  • Figure 4f shows the sum of the precision delayed spectrum and the reversed spectrum 438, 440.
  • both the visual signal 438 and the data signal 440 have become double sideband.
  • the visual signal 438 has only in-phase components throughout and the data signal 440 has only quadrature components throughout.
  • the two signals can be completely separated with synchronous demodulation without the need for precision filters.
  • the data path no longer requires a well designed filter to remove quadrature components of the visual signal 438 from the data path.
  • Figure 5 describes an implementation of a Spectrum Reverser 502.
  • the Spectrum Reverser 502 is not the invention itself. It will be appreciated that other methods of implementation of spectrum reversal would also be effective in implementing this invention. Other methods of implementation will be understood by those of ordinary skill in these arts.
  • the first Local Oscillator 504 operates at a frequency N times the receiver ' s intermediate frequency, IF.
  • Mixer # 1 506 multiplies this oscillator's 504 cosine wave output with the combined video and data signal.
  • Figure 5b shows the combined video and data signal 508.
  • Figure 5c shows the result when N is set equal to 3. This choice is for illustrative purposes only and is not a limitation.
  • the mixer 506 behaves as a doubly balance mixer yielding and output which is comprised of the sum frequencies and the difference frequencies.
  • the input components are balanced out and do not appear at the output.
  • the local oscillator signal 510 is depicted in Figure 5c as a dashed vector to indicate its location at the input to the mixer 506, but to also indicate that it is not present in the output signal.
  • the sum frequencies 512 which form the upper sideband in Figure 5c are retained with the Band Pass Filter # 1 514 and the lower sideband 516 is rejected by that filta- 514. Note that the spectrum 512 is not reversed at this point, but merely shifted to another frequency.
  • Next Local Oscillator 518 # 2 operates at a frequency (N + 2) times the IF frequency.
  • Local Oscillator # 2 518 in this illustration operates at five times the IF frequency. This choice is for illustrative purposes only and is not a limitation.
  • Mixer # 2 520 multiples this oscillator's cosine wave output with the output of Band Pass Filter # 1 514.
  • Figure 5 d shows output of Mixer # 2 520.
  • the mixer behaves as a doubly balance mixer yielding and output which is comprised of the sum frequencies and the difference frequencies. The input components are balanced out and do not appear at the output.
  • the local oscillator 518 signal is depicted in Figure 5d as a dashed vector 520 to indicate its location at the input to the mixer, but to also indicate that it is not present in the output signal.
  • FIG. 6 describes an implementation of a Precision Phase-Correct Delay.
  • the Precision Phase-Correct Delay is not the invention itself. It will be appreciated that other methods of implementation of a Precision Phase-Correct Delay would also be effective in implementing this invention. Other methods of implementation will be understood by those of ordinary skill in these arts.
  • Comparison of Figure 5a, Figure 5c, and Figure 5d respectively with Figure 6a, Figure 6c, and Figure 6d illustrates that the respective filters 514 and 614, 524 and 624 are identical and the mixers 406 and 606, 420 and 620 are identical. Thus the propagation time through this circuit will be identical to that of the Spectrum Reverser 502 to the precision of the matching of the components.
  • Optional Phase Adjusters 630, 632 have been added to compensate for mismatches in the implementation.
  • the Local Oscillators 504, 518, 604 operate at different frequencies, but that does not impact the propagation delay through the system.
  • the explanation of Figure 5a through Figure 5d provides an understanding of the operation of Figure 6a through Figure 6d.
  • the information signals in the vestigial sideband portion of the received spectra are correlated with the information signals in the unattenuated other side.
  • the voltages add when spectrum reversal and addition is done.
  • the noise in these two sidebands is uncorrelated and the noise powers add. This results in an advantageous signal to noise ratio improvement.
  • the filters of Figure 5a and Figure 6a 514, 524, 614, 624 aid in understanding the operation.
  • the primary function of the filters is to prevent overloading the mixers 506, 606, 520, 620. Ideal mixers would not require these filters. That is, mixers with sufficient dynamic range would not require these filters.
  • the filters 514, 524, 614, 624 are a source of expense and complexity and delay that can be avoided with adequate mixer design.
  • Figure 5e shows the Spectrum Reverser 502 of Figure 5a, but without the band pass filters 514, 524.
  • Figure 6e shows the Precision Phase-Correct Delay 634 of Figure 6a, but without the band pass filters 614. 624.
  • Figure 6j is the output of the Phase Shifter 410 of Figure 4a that feeds the Mixer 412 connected to the Data Output.
  • Figure 6k shows the output of that Mixer 412. Note that it is the sum and difference of the IF frequency and the spectra of Figure 6i. If the result is then low pass filtered with a relatively simple low pass filter 652, only the data baseband spectra 654 remains.
  • the low pass filter 652 is simple and inexpensive because the closest interfering spectrum 656 is at twice the IF frequency, some ninety MHz away from the baseband data spectra which consists of less than a few MHz. In fact, normal parasitic reactances will attenuate the higher frequencies. If these higher frequency components were not completely removed, they would only have the effect of slightly closing the eye pattern when the data signal is detected and converted into a digital stream. Depending on the system signal to noise ratio, this may be quite acceptable.
  • Figure 7 shows the spectra in a larger, easy to see form.
  • Figure 7a is the output of the Nyquist Filter 406 and the Precision Phase Correct Delay 420.
  • Figure 7b shows the output of the Spectrum Reverser 416.
  • Figure 7c sums the spectra at the outputs of the Precision Phase-Correct Delay 420 and the Spectrum Reverser 416.
  • Both the television 738 and the data spectra 740 are double sideband after the Summer.
  • the television signal 738 has just in-phase components and the data signal 740 has only quadrature components. These components are easily separated with synchronous demodulation techniques.
  • Figure 8 shows a further simplification in cases where high quality video is not required or where just a data receiver is implemented. In that case, the Nyquist filter
  • the increased signal strength in the visual signal will be of advantage when the visual signal is used as a "training signal" for the data.
  • the adjustment will be aided by the increase signal strength in the range of frequencies occupied by the data.
  • Nyquist filter 406 saves expense and complexity. This is especially of importance in an integrated circuit implementation where filters are a challenge.
  • the absence of Nyquist filter 406 facilitates implementation with a just a few or even just one integrated circuit.
  • Figure 9a displays the transmitted data spectrum 924.
  • Figure 9b shows the reversed data spectrum 934.
  • Figure 9c shows the addition of the two spectra 924, 934 yielding a symmetrical spectrum 940 with only quadrature components.
  • the Nyquist Filter 406 attenuates data signal 424 frequencies that are closer to the band edge while simultaneously attenuating the NTSC signal 422. Since the NTSC signal 422 and the data signal 424 are combined, it is not possible to attenuate portions of the NTSC signal 422 and not attenuate portions of the data signal 424 where they both occupy the same frequencies. This attenuation of the d ⁇ ta. frequencies in the receiver reduces the signal to noise ratio and that lowers the margin of data recover performance.
  • the implementation of Figure 8 avoids this reduction in signal to noise ratio by utilizing the entire received data spectrum rather than rejecting a significant portion of it. In both implementations, the signals in the sidebands are correlated so their voltages add while the noise is uncorrelated and so the noise powers add. The result is an important improvement in signal to noise ratio.
  • Figure 10 demonstrates the results of extending the data spectrum beyond the normal double sideband region of the NTSC signal 1060 (which consists of those frequencies between 750 KHz above the visual carrier and 750 KHz below the visual carrier).
  • Figure 1 1 shows the spectra details.
  • Figure 1 1a is the Data RF Transmit filter 1 1 18. Note that it severely attenuates the data signal between 750 KHz above the visual carrier and 1.25 MHz above the visual carrier.
  • Figure l ib is the NTSC VSB filter 1 102 that strongly attenuates the data (and visual) signal between 750 KHz below the visual carrier and 1.25 MHz below the visual carrier. At frequencies more than 1.25 MHz below the visual carrier, the VSB filter 1 102 severely attenuates the data and visual signals.
  • Figure 1 lc is the composite filter function 1 166 of these two filters at the transmitter.
  • Figure l id repeats this composite filter function 1 166, shows a raised cosine data spectra 1 168 (for illustrative purposes only and not as a limitation) and the result 1170 of the composite filter function operating on the data spectra 1 168.
  • the transmitted data spectra 1 170 is highly unsymmetrical. It slumps 1 172 between 750 KHz below the visual carrier and 1.25 MHz below the visual carrier. It is severely attenuated between 750 KHz above the visual carrier and 1.25 MHz above the visual carrier. However, the information- carrying bandwidth of this signal extends to 1.25 mHz. This is 50% more bandwidth than the previous implementations discussed.
  • Figure 12a shows the received data spectra 1270, the same as in Figure l id. It also shows a spectrum-reversed data spectra 1272 and the sum 1274 of the received 1270 and reversed spectra 1272. For comparison, the original data spectra 1276 is also shown in Figure 12a. Within the NTSC double sideband region, + 750 kHz around the carrier, the shape of the sum 1274 of the received 1270 and the reversed spectra 1272 is identical to the original data spectrum 1276. It will be noted that there is some minor distortion in the reconstructed data spectra outside of this region.
  • the spectra slumps in the region between 1.25 MHz below the visual carrier and 750 KHz below the visual carrier and in the region between 1.25 MHz above the visual carrier and 750 KHz above the visual carrier. This distortion is likely not serious and will result in only a slight closing of the eye pattern of the data. It is noted that this could be compensated with a predistortion at the point of origination and this effect eliminated.
  • Figure 12b displays the data signal 1216 in existing television receivers.
  • the television receiver's Nyquist filter 1206 severely attenuates the data in the region between 750 KHz below the visual carrier and 1.25 MHz below the visual carrier. This causes the data spectra 1216 in the television receiver to become symmetrical and have only quadrature components. It also limits the data spectrum 1216 in the television receiver to the NTSC double sideband region. Synchronous demodulation will separate the desired visual components from the data components 1216. Since the visual signal is much stronger than the data signal 1216, small asymmetries in the data spectra will result in only a small impact on the video. Since the data is uncorrelated with the video the small in-phase contribution only adds a trivial amount of noise to the video. This will not be objectionable under nearly all practical circumstances.

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  • Engineering & Computer Science (AREA)
  • Signal Processing (AREA)
  • Multimedia (AREA)
  • Computer Networks & Wireless Communication (AREA)
  • Digital Transmission Methods That Use Modulated Carrier Waves (AREA)
  • Television Systems (AREA)

Abstract

L'invention porte sur l'amélioration des performances de récepteurs de données utilisés pour la démodulation de signaux de données, décrits dans la demande internationale WO 99/55087, recourant à une nouvelle méthode de traitement de spectres pour synthétiser la totalité d'un spectre en bande latérale double. Les nouvelles techniques de l'invention présentent plusieurs avantages importants dont, non exclusivement, l'augmentation du rapport signal/bruit, la relaxation des contraintes de conception des circuits, la création de circuits intégrés de signaux mixtes, et un débit de données plus élevé. L'invention permet également d'améliorer la démodulation des signaux de télévision.
PCT/US2000/033479 1999-12-22 2000-12-08 Amelioration d'un recepteur augmentant la capacite d'information de systemes existants de transmission de communications WO2001047260A1 (fr)

Priority Applications (5)

Application Number Priority Date Filing Date Title
JP2001547862A JP2003518839A (ja) 1999-12-22 2000-12-08 既存の通信伝送システムのための拡大された情報容量に関する受信器の改良
CA002395810A CA2395810A1 (fr) 1999-12-22 2000-12-08 Amelioration d'un recepteur augmentant la capacite d'information de systemes existants de transmission de communications
EP00984146A EP1249125A1 (fr) 1999-12-22 2000-12-08 Amelioration d'un recepteur augmentant la capacite d'information de systemes existants de transmission de communications
AU20818/01A AU2081801A (en) 1999-12-22 2000-12-08 Receiver improvement for expanded information capacity for existing communication transmissions systems
MXPA02006191A MXPA02006191A (es) 1999-12-22 2000-12-08 Receptor mejorado de la capacidad de informacion expandida para sistemas de transmision de comunicacion existentes.

Applications Claiming Priority (2)

Application Number Priority Date Filing Date Title
US17138499P 1999-12-22 1999-12-22
US60/171,384 1999-12-22

Publications (1)

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WO2001047260A1 true WO2001047260A1 (fr) 2001-06-28

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EP (1) EP1249125A1 (fr)
JP (1) JP2003518839A (fr)
AU (1) AU2081801A (fr)
CA (1) CA2395810A1 (fr)
MX (1) MXPA02006191A (fr)
WO (1) WO2001047260A1 (fr)

Cited By (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
EP1675388A1 (fr) * 2004-12-22 2006-06-28 Angel Iglesias S.A. Modulateur de télévision double bande latérale
WO2007040572A1 (fr) * 2005-09-28 2007-04-12 Thomson Licensing Transmission de l'information sur un canal auxiliaire
US7354657B2 (en) 2002-09-30 2008-04-08 The Curators Of University Of Missouri Integral channels in metal components and fabrication thereof
WO2013003215A1 (fr) * 2011-06-28 2013-01-03 Qualcomm Incorporated Procédé et systèmes de forçage à zéro optimal et égaliseurs de domaine de fréquences mmse pour signaux complexes et vsb

Citations (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
EP0253623A2 (fr) * 1986-07-14 1988-01-20 Matsushita Electric Industrial Co., Ltd. Appareil de traitement de signal multiplexé
US4958230A (en) * 1989-08-11 1990-09-18 General Electric Company Method of transmitting auxiliary information in a television signal
WO1999055087A1 (fr) * 1998-04-17 1999-10-28 Encamera Sciences Corporation Capacite d'informations etendue pour systemes de transmission de communications existants

Patent Citations (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
EP0253623A2 (fr) * 1986-07-14 1988-01-20 Matsushita Electric Industrial Co., Ltd. Appareil de traitement de signal multiplexé
US4958230A (en) * 1989-08-11 1990-09-18 General Electric Company Method of transmitting auxiliary information in a television signal
WO1999055087A1 (fr) * 1998-04-17 1999-10-28 Encamera Sciences Corporation Capacite d'informations etendue pour systemes de transmission de communications existants

Cited By (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US7354657B2 (en) 2002-09-30 2008-04-08 The Curators Of University Of Missouri Integral channels in metal components and fabrication thereof
EP1675388A1 (fr) * 2004-12-22 2006-06-28 Angel Iglesias S.A. Modulateur de télévision double bande latérale
WO2007040572A1 (fr) * 2005-09-28 2007-04-12 Thomson Licensing Transmission de l'information sur un canal auxiliaire
WO2013003215A1 (fr) * 2011-06-28 2013-01-03 Qualcomm Incorporated Procédé et systèmes de forçage à zéro optimal et égaliseurs de domaine de fréquences mmse pour signaux complexes et vsb
US8782112B2 (en) 2011-06-28 2014-07-15 Qualcomm Incorporated Methods and systems for optimal zero-forcing and MMSE frequency domain equalizers for complex and VSB signals

Also Published As

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EP1249125A1 (fr) 2002-10-16
CA2395810A1 (fr) 2001-06-28
MXPA02006191A (es) 2002-12-09
JP2003518839A (ja) 2003-06-10
AU2081801A (en) 2001-07-03

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