WO2001023906A1

WO2001023906A1 - Distance measuring method

Info

Publication number: WO2001023906A1
Application number: PCT/DE2000/003356
Authority: WO
Inventors: Patric Heide; Martin Vossiek
Original assignee: Siemens Aktiengesellschaft
Priority date: 1999-09-27
Filing date: 2000-09-27
Publication date: 2001-04-05
Also published as: DE19946161A1; EP1216424A1

Abstract

According to the inventive method, the distance from at least one test object (MO) is measured using a CW microwave sensor (MS). The invention is characterized in that at least one active reflector (AR) is fixed to the at least one test object (MO). Said active reflector is used to receive, modulate and emit a signal sent by the CW microwave sensor (MS, RS).

Description

description

Distance measurement procedure

The invention relates to a method for measuring a 7-distance and a 7-distance change of an object with the aid of microwaves, 7 uses thereof, and devices for reflecting microwave signals.

For example from J. Detlefsen: "Radartechnik", Springer

Verlag Berlin, 1998, it is known to use radar waves to measure a relative distance or a relative speed between a radar device and one or more test objects, in particular by measuring the pulse transit time and by means of FMC measurement.

Microwave-based, and especially radar-based, 7-range and speed measurement methods offer advantages over an alternative waveform such as ultrasound or laser due to a low radio attenuation and a high degree of independence from the spread of temperature, pressure, humidity, etc.

In the pulse transit time method, a short radar pulse is emitted in the direction of a measurement object and received again as a reflected pulse after a transit time τ. The transit time of the radar pulse is proportional to the / distance between the radar device and the measurement object.

In the FMCW (Frequency Modulated Contmuos Wave) method, a linear or step-wise frequency-modulated radar signal is emitted. With stepwise modulation, at least two different frequency values are approached for a distance measurement. Between the transmitted and received signal on the radar device there is a frequency or a phase shift corresponding to the transit time τ. The FMCW method is the most widespread in commercial radar sensors. Works a general C (Contιnous Wave) radar, so also z. B. a Doppler radar with a monofrequency radar signal with a frequency f _{HF /} so the phase difference φ between the transmitted signal and a received signal reflected by the measurement object

φ = 2 π • f _H τ, (1)

where τ represents the total transit time of the radar signal. In addition to constant offset influences, the transit time τ is directly linked to the distance d from the radar sensor to the reflector unit via the propagation speed c of the radar waves. The following applies:

τ = τ _φ +. (2) c

Due to the periodicity of the phase (and because of the generally unknown offset τ ₀ ffs), a phase measurement at only one radar frequency is only suitable for determining differential changes in distance.

An absolute position value can e.g. B. by continuously determining a change in distance following a calibration measurement relative to a calibration reference point.

In the case of a continuous measurement, care must be taken that the measurement is carried out so quickly that there is no such great distance change between two measurements that results in a phase change greater than 180 °. This condition corresponds to the well-known sampling theorem.

Continuous tracking of the phase is often referred to as Doppler measurement, where the time derivative of the phase corresponds to the Doppler frequency. The Doppler frequency is proportional to the relative speed between the radar sensor and the measurement object in the direction of the signal transmission. F _o r an absolute distance measurement must be at least two phase values which are determined at least two different radar frequencies evaluated. If Δf _HF _denotes the frequency difference between two radar frequencies f _HF ι and f _{HF2, the} following applies to the difference Δφ between the measured phase values φl and φ2:

Aφ = 2 π Af _HF τ.

If Δf is not chosen too large, Δφ is unique within a large distance range. To ensure an optimal measurement effect, however, Δf should not be chosen to be unnecessarily small for a distance measurement. A simultaneously high measuring effect, combined with a large ambiguity range, can be achieved by using still further measuring frequencies (and thus further phase measured values).

In the limit case of a continuous frequency modulation, the radar frequency is measured with the time change and the phase change. In the case of linear frequency modulation, the known methods for FMCW signal processing are used, as are disclosed, inter alia, in WO 99/10757. All radar methods that evaluate signal frequencies for distance measurement are based on the above-mentioned phase relationships, since a frequency is only a time derivative of the phase.

These and other explanations of systems theory are described in M. Vossiek, T. v. Kerssenbrock, P. Heath, "Si gnal Processing Methods for Millimetrewave FMCW radar with high Distance and Doppler resolution", 27 ^th European Microwa- ve Conference, Jerusalem, Israel, pp. 1127-1132, 1997.

Realization options for FMCW radar sensors with different topologies are, for example, m: B. Zimmermann et al: "24 GHz Microwave Close-Range Sensors For Industrial Measurement Applications ", Microwave Journal, May 1996; in: EP 0 647 857 AI, WO 99/10757; in: M. Vossiek et al. (See above); and in: M. Nalezmski, M: Vossiek, P. Heide: "Novel 24 GHz FMCW Front-End with 2.45 GHz Ξ AW Reference Path for High-Precision Distance Measurements", 1997 IEEE MTT-S Int. Microwave Symp., Denver, USA, pp. 185-188.

As a rule, a high level of sensitivity combined with a long range is desirable. The range depends, among other things, on the transmission power and the level of the antenna. The radio-technical approval regulations limit the permissible output per area. The reachable measuring range is therefore typically a few tens of meters in practice.

In particular when measuring large distances, the measurement accuracy may be restricted by objects in the store. If there are other objects besides the measurement object in the propagation path or in the environment of the measurement object, the wave propagation can be disturbed because interfering interactions of the desired reflections of the measurement object to be measured and the measurement environment result, for example by multipath wave propagation. This case will occur in particular if a radio antenna has an omnidirectional characteristic, that is to say if there are many objects in the detection range of the antenna at the same time. Such an interference can only be reduced to a limited extent by intelligent signal evaluation.

It is known for the calibration measurement m of the antenna measurement technology that metallic reference reflectors, e.g. B. triple reflectors are used. These passive reflectors have a high scattering cross section compared to their geometrical dimensions, ie a high reflectivity. It is an object of the present invention to provide a possibility for reducing interference influences in the distance measurement by means of microwaves, in particular radar waves.

It is a further object of the present invention to provide a possibility for distance measurement by means of microwaves, in particular radar waves, with a long range.

These tasks are solved by a method according to claim 1, an application of the method according to claim 17 and by a device according to claim 19. Advantageous refinements can be found in the subclaims.

For this purpose, a method for distance measurement is used, in which a distance to at least one object is measured by means of a CW microwave sensor and at least one active reflector is attached to the at least one object, which rather receives, modulates and signals emitted by the CW microwave sensor emits again after the modulation. Amplitude modulation, frequency modulation or any combination of both types of modulation is possible as modulation.

This method is suitable for all CW microwave sensors, in particular CW radar sensors such as the Doppler radar and the FMCW radar. For simplification, the following versions are described above all with the help of radar sensors. Of course, the use of other types of microwaves is not excluded.

Of course, not only a distance, but also a change in distance and / or a position and / or any time derivative of these variables can be determined. To measure the distance and the change in distance, it is sufficient to use only one radar sensor. One, generally my three-dimensional, but also two-dimensional, position can be z. B. by means of a single radar sensor and performing a Kalibπermessung or by means of two radar sensors with triangulation determination.

An active reflector is from the area of communication and flight location, for example as a secondary radar (secondary surveillance radar SSR) from Memke Gundelach, "Taschenbuch der Hochfrequenztechnik", chapters S8, 1.7 and 1.8, 5th edition, Berlin, Heidelberg, New York: Springer 1992, known. An active reflector unit for testing the measurement quality of radar systems with a synthetic aperture is known from US Pat. No. 5,512,899.

Typically, the radar signal emitted by the radar sensor is picked up by the active reflector by means of an antenna, passed to a modulator (eg via a directional element) and then sent back to the radar sensor via the antenna of the active reflector. An output of the modulator can, for. B. be connected to a third terminal of the directional element.

The modulator can be an amplifier by which the radar signal in the active reflector is amplified in proportion to the signal level of the incoming radar signal. Such a pure amplitude amplification has the effect that the radar pulse sent back from the active reflector to the CW radar sensor is more pronounced in comparison to the interference signals. This makes it possible to suppress interference signals and increase a measuring range similar to a passive reflector. In contrast to the passive reflector, the height of the reinforcement is largely independent of the surface of the reflector and is also largely freely selectable.

The modulation can also bring about a frequency modulation, for example a simple frequency shift, by means of which the radar signal returned from the active reflector to the radar device is frequency modulated. As a result, the radar signal output by the active reflector can advantageously be separated from the interference signals in the frequency range of the radar sensor, for example by bandpass filtering. The filtering or demodulation can, for. B. with an electronic circuit or algorithmic m a processor.

This method has the advantage that interference influences are reduced and a long range can be achieved. It is also flexible and comparatively inexpensive.

Each active reflector favorably transmits with a frequency modulation which is characteristic of it. In this way, signals from several active reflectors can be distinguished and separated. In addition to a distance or position determination, an object identification can thereby also be carried out.

So z. B. equipped several objects with at least one active reflector. The characteristic modulation is expediently the same for all active reflectors attached to a measurement object, but different within a group of measurement objects.

It is particularly advantageous, among other things, to increase the measuring range if the radar signal is frequency-modulated as well as amplitude-modulated in the active reflector. For this purpose, a frequency modulator can be followed by an amplifier in the active reflector. The amplifier can then be connected to the third connection of the straightening element.

In particular, if there is a transmission path between the CW radar and the active reflector, it is advantageous if a distance value and / or a change in distance between the measurement object and the CW radar is determined by separating the radar signal emitted by the CW radar and the Radar off received radar signal a phase difference or a temporal change in the phase difference is evaluated.

If there are two different transmission paths between the CW radar and the active reflector, at least two distance values can be determined and these distance values can be combined using geometric equations (e.g. triangulation), thus determining the spatial position of the object.

It is also advantageous if the radar signal is amplitude-modulated in the amplifier by switching the amplifier with a modulation signal with a clock frequency fM, that is to say switching it on and off, and additionally the radar signal received in the CW radar by means of a high-pass filter or a Bandpass filtering is filtered and then demodulated.

It is particularly advantageous if the clock frequency fM is between 100 kHz and 10 MHz, which is typically significantly higher than the measuring frequency.

This amplitude modulation has the advantage that the useful signal emitted by the active reflector appears in the spectrum of the sensor signal on the CW radar as a modulated signal at a modulation frequency. The interference signals, however, are unmodulated and occur in the baseband. A simple high-pass or band-pass filtering with subsequent demodulation effectively suppresses the interference signals. The demodulation is of course matched to the modulation in the active reflector.

In the case of amplitude modulation, it is not necessary to vary the amplification of the signal only in a dual manner (ie “on” or “off”). Gain changes in analog steps can also be used for modulation. In the case of amplitude modulation, it is advantageous not to carry out the change in gain with a constant clock ratio, but rather according to certain pseudo-random code sequences. The known code sequences such as Barker codes, shift register sequences (M sequences), Golay codes, Gold codes or Huffmann sequences can be used for this purpose, among others. If several active reflectors are present at the same time, preference is given to using sequences which are not correlated or only very slightly correlated, so that a correlation in the CW radar enables separation and unambiguous assignment of the reflected signals.

It is also advantageous if the active reflector changes the radar signal by means of frequency modulation, in that it contains a mixer module which frequency mixes the incoming radar signal with a reference signal of frequency f. The frequency-changed signal is sent back to the radar sensor and thus occurs in the spectrum of the CW radar at a changed frequency. Appropriate bandpass filtering with subsequent demodulation can be used to suppress the stocho echoes in a manner similar to that in the case of amplitude modulation. It is advantageous, in particular to increase the measuring range, if the frequency-modulated radar signal is emitted simultaneously from the active reflector.

It can also be advantageous if, instead of a mixer, a phase shifter is used which changes the phase of the signal arriving at the active reflector. The phase is changed, for example, as m R. Mausl, digital modulation method, telecommunications, Heidelberg: Huthig Buch Verlag GmbH, 1991, called.

It can furthermore be advantageous to generate the radar signals in the active reflector by a defined time offset. The time delay is expediently so great that interference signals, for example caused by reflections on in the measuring range, have largely subsided due to propagation vaporization in the free space. If the time delays are also selected differently for each reflector, the signal components can be clearly separated using known CW radar evaluation methods.

When dimensioning a delay element used to delay the transit time, care must be taken to ensure that there is no overlap of the transit times of different signal components of the reflector in the distance range given for the corresponding application.

It is particularly advantageous if the delay element is designed with one or more delay lines in the form of a surface wave component (SAW). Such OFWs are, for example, n C. Ruppel, L. Rhemdl, S. Berek, U.

Knauer, P. Heide, M. Vossiek, "Design Fabrication and Application of Precise Delay Lines at 2.45 GHz, IEEE, Ultrasonics Symposium, San Antonio, USA. 1996.

If the delay line is implemented with surface wave components, it is particularly advantageous to also apply a resonator structure directly to a substrate on which the delay line is built, from which the modulation frequency or the modulation signal can be derived. This optimally ensures that runtime changes due to temperature drift and aging of the substrate result in a proportional change in the modulation frequency. Because the transit time and the modulation frequency are generated on the same substrate with the same basic physical principle, they are automatically coupled in the desired manner. Dimensions of SAW delay elements and SAW resonators can be found in the relevant literature. The substrate preferably consists of lithium niobate or quartz.

It is particularly favorable to combine amplitude modulation with a time delay. Here, the sales Strengthening z. B. m regular intervals on and off. The modulation frequency is preferably chosen so that it is mvers coupled to the delay time of the delay element, z. B. a delay of 1 μs is selected at a modulation frequency of 1 MHz.

Such an arrangement can prevent the amplified and modulated signal from again entering the input circuit of the active reflector, from being amplified again and thus from overdriving or feedback coupling.

Caused swellings. By coupling the delay to the modulation frequency, it can be ensured that the input circuit is always switched off when the amplified and delayed signal is sent back to the radar sensor.

Of course, the modulation of the radar signal in the active reflector is not limited to the modulation methods described above, further explanations of modulation methods can be found, for example, in Mausl et al.

All of the coding methods mentioned can also be used in any combination. It is particularly advantageous for a highly accurate measurement of distance and / or position to connect the change in time of the phase at one radar frequency, that is to say the differential distance measurement, to the phase difference values in the case of a plurality of Raαar frequency values, that is to say the absolute distance measurement. The absolute measured values can e.g. B. can be corrected by the much more precise differential change values. It is important to ensure that the measuring process is sufficiently fast.

For an exact distance measurement it is also important that the radar frequency and in particular the frequency change can be set and maintained very precisely. In addition to analog control loops and calibration devices with reference delay elements, phase control ω co M to h → P ¹ cn o Cπ o cπ o CJi

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Identification identifier of the autonomous vehicle. K also can further data _ö of the autonomous vehicle, such as its destination or special transport orders are transmitted.

Another advantageous application of the method is to determine the position of a container in a high-bay warehouse. This use is similar to determining the position of autonomous vehicles, except that at least two interrogation devices are preferably arranged within a rack storage aisle, with which the height and position of the container is then determined as described.

For this purpose, the container, typically a transport box or a unit of the transport vehicle, which contains the transport box, is provided with an active reflector unit. This means that the transport box can be taken from a specific position on the high rack or placed in a specific position. Data about the box to be removed or data about the removed box (for example, content, size, etc.) can be transmitted from the interrogation device to the transport unit or vice versa using the communication arrangement described.

The method for determining the position can also be used very advantageously for determining the position of tools. Here, a tool is provided with an active reflector, preferably as close as possible to its mechanical point of application to the workpiece. Depending on the degree of freedom of the movement of the tool, the distance of the tool to the respective radar system is then determined using one or more methods or arrangements as described above. The exact position of the tool is then determined and / or regulated by combining the distance values.

Tools include, for example, tools for material processing, such as turning, milling, drilling, punching and / or cutting tools, gripping tools or also medical in- instruments into consideration. The task of the measuring system is to determine the position and / or the position of the tool in space as precisely as possible. The advantage over a mechanical measuring system is that the position of the tool can be measured directly on the tool without contact and an error due to mechanical deformation (torsion, bending, etc.) is avoided. The principle of the active reflector is particularly suitable because a passive reflector arrangement generally does not provide sufficiently precise measurement results due to a large number of strong reflections on the metal surfaces customary in machine tools. The active reflector prevents interference from disturbing reflections, so that even highly reflective environments can be measured safely and precisely.

In all of the arrangements mentioned, both the location of the reflector and radar device and the number of active reflectors and radar sensors can be interchanged. The information that can be obtained is only determined by the number and geometric position of the measuring paths.

The method is equally suitable for measuring a change in distance, a distance, a (two- or multi-dimensional) position or position and / or a speed.

In the following examples, the method for distance measurement is schematically explained in more detail.

FIG. 1 shows a device for distance measurement, FIG. 2 shows an FMCW radar sensor,

FIGS. 3a to 3d show embodiments of an active reflector,

FIG. 4 shows a sketch of a spectrum of a signal received at the radar sensor.

FIG. 2 is a schematic circuit diagram of a typical microwave measuring arrangement in the form of a CW microwave sensor MS m form of an FMCW radar sensor RS shown with oscillator (VCO), signal generator / linearizer (SGLIN), radar receiver (EMIX) and antenna (ANT). There is also an evaluation unit (FFT) for fast Fourier Transformation (FFT) and a reference unit (REF).

Via a control signal, e.g. For example, a voltage applied from the outside, the frequency of the oscillator VCO can be set. The signal emitted by the oscillator VCO is emitted as a radar signal via the antenna ANT and the resulting reflections are received again (indicated by the arrows). Between the oscillator VCO and the antenna ANT is the radar receiver EMIX, which transmits a first part of the signal emitted by the oscillator VCO to the antenna ANT and carries out a frequency mixing with the received signal with a second part of the signal. The radar or Microwave receiver EMIX is designed as a detector mixer.

This radar sensor RS supplies a sensor signal SES, the phase or frequency of which is proportional to a distance dr of the measurement object MO. By evaluating phase values and / or phase difference values and / or frequency values, the distance dr from measurement objects and / or the distance change and / or the speed can be determined according to known methods.

In this exemplary embodiment, a linear frequency modulation is used to determine the object distance dr, which is evaluated by means of a fast Fourier transformation m of the evaluation unit FFT. The FFT evaluation unit receives the sensor signal from the EMIX radar receiver. The result of the fast Fourier transformation is an echo profile that represents the radar reflections as a function of the distance. A signal derived from the oscillator VCO is fed into the signal generator / linearizer SGLIN, which controls the frequency of the oscillator VCO in such a way that linear frequency modulation results. A comparison with a stable reference REF usually serves as the basis for controlling the modulation voltage.

In this figure there is a measurement object MO and a plurality of storage objects SO m at approximately the same distance dr, e.g. B. dr = 100 m - 200 m, to the antenna ANT of the radar sensor RS.

In the spectrum of the sensor signal, the individual echoes overlap to form an overall echo that can no longer be discriminated. In the worst case, the reflection of the storage objects SO is greater than the reflection of the measurement object MO. With moving Whether _j ects MO, SO, it may also come through interference effects to a strong amplitude fluctuation of echo. Overall, this has the consequence that the measurement accuracy is reduced or the measurement is prevented.

In FIG. 1, the measurement object MO is now equipped with an active reflector AR. The active reflector AR has, as subcomponents, an antenna A2, a directional element RE and a modulator MOD. The modulator MOD contains an amplifier AMP and a frequency modulator FMOD connected downstream of this.

The radar sensor RS can, but need not, correspond to the radar sensor RS in FIG. 2. For clarity of the measurement situation, the antenna ANT of the radar sensor RS is also shown.

The radar signal emitted by the radar sensor RS is picked up by the active reflector AR by means of the antenna A2, passed via the directional element RE to the amplifier AMP and from there the frequency modulator FMOD. The frequency-modulated and amplified signal is then m a third connection of the Directional element RE fed and then sent back to the radar sensor RS via the antenna A2 of the active reflector AR.

The received signal is filtered or demodulated again in a filter unit FIL / DEM connected downstream of the radar sensor RS and then entered as a sensor signal SES in an evaluation unit FFT. Of course, filter unit FIL / DEM and evaluation unit FFT can also be integrated in the radar sensor RS.

FIG. 3a shows an active reflector AR which generates an amplitude modulation. In contrast to the active reflector AR in FIG. 1, the frequency modulator MOD is missing here.

The amplitude modulation is generated by switching the amplifier AMP with a modulation signal MSIG with a clock frequency fM, that is to say here: on and off. The clock frequency fM is chosen to be significantly higher than the measuring frequency, here: 100 kHz <fM ≤ 10 MHz. As a result of this amplitude modulation carried out in the active reflector AR, the useful signal advantageously occurs in the spectrum of the sensor signal SES as a modulated signal at a modulation frequency. The interference signals, however, are unmodulated and occur in the baseband.

The interference signals are thus effectively suppressed by simple high-pass or band-pass filtering in the filter and / or demodulation unit FIL / DEM with subsequent demodulation. The demodulation is of course matched to the modulation in the active reflector AR.

With amplitude modulation, it is generally not necessary to merely vary the amplification of the signal. Amplification changes in analog steps can also be used for modulation. In the case of amplitude modulation, it can also be advantageous not to carry out the amplification change with a constant clock ratio, but with certain pseudo-random modulation sequences. The known code sequences, such as Barker codes, shift register sequences (M sequences), Golay codes, Gold codes or Huffmann sequences, can be used for this purpose, among other things.

If a plurality of active reflectors AR are present at the same time, preference is given to using sequences which are not correlated or only very slightly correlated, so that a correlation in the radar sensor RS enables the reflected signals to be separated and clearly assigned.

Also by coding the pulse sequences, for. B. the above-mentioned code sequences, data are transmitted. The data transmission can also take place in such a way that the amplifier clock is switched over in its frequency (high / low) or in its clock ratio.

A further active reflector AR is shown as a sketch in FIG. 3b. This now contains a mixer module MIX, which mixes the signal output by the directional unit RE with a reference signal of the frequency fR, which can also be referred to as the modulation frequency, and forwards it to the amplifier AMP.

Depending on the design of the mixer module MIX, it is a single-side band mixer or a double-side band mixer. In the case of a single-sideband mixer, the incoming signal is sent back to the radar sensor RS with a frequency offset and occurs with an offset frequency fR in the spectrum of the sensor signal SES. Suitable bandpass filtering with subsequent demodulation can suppress interference signals. In this exemplary embodiment, data can be transmitted, for example, by changing the modulation frequency fR, which corresponds to a so-called FSK (frequency key shiftmg) modulation.

FIG. 3c shows a further active reflector AR. This now has a controllable phase shifter PHS, which changes the incoming signal with respect to the phase via a phase signal PSIG, which can also be viewed as a modulation frequency. The phase is based on known schemes (see, for example, Mausl. Et al.).

In this exemplary embodiment, the phase modulation (phase key shift mg) known from communication technology is favorable for data transmission.

FIG. 3d shows a further active reflector AR, in which the signals in the active reflector AR are delayed by a defined time offset in the delay element DEL, which determines the time delay, and which is connected between the directional element RE and the amplifier AMP. It is advantageous if the delay element DEL is designed with one or more delay lines in the form of a surface wave component (SAW).

FIG. 4 shows a frequency spectrum of a signal received at the radar sensor as a plot of a signal level in arbitrary units against a frequency f m of any units.

The signal emitted by the radar sensor MS, RS is given by

Stx (t) = cos ((ω ₀ + 0.5μt) t) (1),

where ω ₀ and f _{0 represent} the modulation start frequency ("sweep start frequency") and μ the tuning rate ("sweep rate") provides. The signal received by the radar sensor MS, RS is a sum of several time-delayed images of the transmitted signal; however, only the signal component emitted by the active reflector AR is _modulated with a frequency ω _m0 d or f _mod . As a result, the signal received by the radar sensor MS, RS can be represented as

s _rx = s _tx (t-τ _r ) - cos ((ω _mod -t + φ _m0 ) + ∑ ^* s _tx (t-τ _n ) (2)

with τ _r the transit time from the radar sensor MS, RS to the active reflector RS and back and τ _n the transit time of the other objects. For simplicity, changes in the amplitude level are not taken into account. Also for simplification, the rectangular amplitude modulation is approximated by cosine modulation (frequency modulation), so higher terms are neglected. In practice, higher terms can be suppressed using a simple bandpass filter.

The radar receiver EMIX multiplies that from the radar device

MS, RS broadcast and received signals, resulting in a superimposed signal

S _m (t) = cos (ω _mθ d * t + μ-τ _r -t + ω ₀ -τ _r ) + cos (ω _mod -t + μ-τ _r -t + ω ₀ -τ _r ) + ( 3)

∑π cos (ω ₀ -τ _n + μ-τ _n -t)

leads. Higher frequency components and phase terms, which are not important for distance measurement, are neglected here.

The two spectral components mcl, mc2 of the active reflector AR are centered around the modulation frequency f _mod , while the other signals are located in the baseband ("baseband"). So if the modulation frequency f _mo and the

Sweep rate μ are chosen appropriately, the desired, from active reflector AR, the signal emitted can be clearly separated from the interference signals.

For this purpose, a typical spectrum of received signals is plotted in this figure, in which the frequency components according to Eq. (3) are plotted.

_From a consideration of the two spectral lines ml, mc2 caused by the active reflector AR, there is a distance dr between the radar receiver MS, RS and the active reflector AR

Δf = μ-τ _r / π → dr = c-τ _r = π-c-Δf / (2-μ) (4)

Δφ ^* = 4 π f ₀ τ _r → dr = c-Δφ / (8-π-f ₀ ) (5),

with Δf the frequency difference and Δφ the phase difference. Due to the periodicity of the phase, the phase difference Δφ from Eq. (5) cannot be used to measure an absolute distance dr, but it is very useful for precisely determining a change in distance. It makes sense to determine the absolute distance dr from equation (4) and to refine the measurement using Eq. (5).

In this method, instead of an absolute measurement of frequency or phase, a difference measurement .DELTA.f or .DELTA..phi. Is carried out, which is obtained from the two adjacent modulation components mcl, mc2. This means that from Eq. (4) and (5) values do not depend on the modulation frequency f _mθd . A very precise measurement is therefore possible even when the modulation frequency fmod drifts.

A measurement inaccuracy can be caused by a changing time delay of the delay element DEL, z. B. is caused by a temperature change of an SAW.

In a SAW, the delay time is proportional to the modulation frequency f _mθ so that a changing time delay can be recognized and compensated for by monitoring the modulation frequency f _mod . The modulation frequency f _mod , which corresponds to the average of the two frequency components reflected by the active reflector AR, can be measured very precisely over many measurement cycles.

Claims

1. A method for measuring a distance, in which a distance to at least one measurement object (MO) is measured by means of a CW microwave sensor (MS), characterized in that at least one active reflector (AR) is connected to the at least one measurement object (MO) ) is attached, which receives a signal emitted by the CW microwave sensor (MS, RS), modulates it and then emits it.

2. The method according to claim 1, wherein

the signal emitted by the CW microwave sensor (MS, RS) is received by the active reflector (AR) by means of a reflector antenna (A2),

is directed via a directional element (RE) to a modulator (MOD) and is modulated by the latter,

- Is given to a third connection of the directional element (RE), - is emitted via the reflector antenna (A2).

3. The method according to claim 1 or 2, in which the signal received by the CW microwave sensor (MS, RS) is demodulated in accordance with the modulation of the active reflector (AR).

4. The method according to claim 3, wherein

- Several measurement objects (MO) are each equipped with at least one active reflector (AR), the at least one active reflector (AR) modulating a signal m that it receives in a characteristic manner, and

- The CW microwave sensor (MS, RS) demodulates the signal it receives in accordance with the modulation of the active reflectors (AR) and evaluates it separately.

5. The method as claimed in claim 1, in which a value for a distance (dr) and / or a change in the distance (dr) is determined for a transmission path between the CW microwave sensor (MS, RS) and the active reflector (AR), by evaluating a phase difference and / or a temporal change in the phase difference between the signal emitted by the CW microwave sensor (MS, RS) and the signal received by the CW microwave sensor (MS, RS).

6. The method of claim 5, wherein a distance dr according to an equation dr = π-c-Δf / (2-μ) and a change in the distance (dr) based on an equation dr = c-Δφ / (8-π -f ₀ ) is determined.

7. The method according to any one of claims 1 to 6, in which between CW microwave sensor (MS, RS) and active reflector (AR) at least two distance values are determined via different transmission paths and these distance values are combined using geometric equations, and so the spatial position of the measurement object (MO) is determined.

8. The method according to any one of claims 2 to 7, in which

- The signal received by the active reflector is amplitude modulated by switching the amplifier (AMP) with a modulation signal with a clock frequency fM, and - the signal received in the CW microwave sensor (MS, RS) is filtered by means of high-pass filtering or bandpass filtering and is then demodulated.

9. The method according to claim 8, in which a digital clock signal with a modulation signal

Clock frequency fM between 100 kHz and 10 MHz is used.

10. The method according to any one of claims 8 or 9, wherein the clock frequency is determined by Barker codes, shift register sequences, Golay codes, Gold codes or Huffmann sequences.

11. The method according to any one of claims 2 to 10, in which

the signal received by the active reflector (AR) is introduced into a mixer module (MIX), and m is frequency-offset with a reference signal of frequency fR,

- The signal received by the CW microwave sensor (MS, RS) is filtered by means of bandpass filtering and then demodulated.

12. The method according to any one of claims 2 to 11, in which

the signal in the active reflector (AR) is introduced into a phase shifter (PHS) and is frequency-shifted in this,

- The signal received in the CW microwave sensor (MS) is filtered by means of a bandpass filter and then demodulated.

13. The method according to any one of the preceding claims, wherein the signal in the active reflector (AR) is introduced into a delay element (DEL), the signal received in the CW microwave sensor (MS, RS) is filtered by means of a bandpass filtering and then demodulated.

14. The method according to claim 13, in which a surface wave component is used as the delay element (DEL).

15. The method according to any one of the preceding claims, in which an FMCW radar (RS) is used as the CW microwave sensor (MS).

16. The method according to any one of the preceding claims, in which the signals transmitted by the active reflector (AR) are modulated such that data are transmitted by a subsequent demodulation.

17. Application of the method according to one of claims 1 to

at least two CW microwave sensors (MS) are present at different positions, by means of which the distance to the active reflector (AR) connected to a test object (MO), in particular an autonomous vehicle or a transport condition, is measured, and

- The position of the autonomous vehicle is determined by a combination of the measured distance values.

18. Application according to claim 17, in which a plurality of active reflectors (AR) are present on the measurement object (MO), so that the position of the measurement object (MO) in space is determined by evaluating the position of the plurality of active reflectors (AR).

19. Active reflector, having

an antenna (A2) which is connected to a directional element (RE),

- A modulator (MOD), which is attached between an output and a third connection of the directional element (RE)

so that a signal received by the reflector antenna (A2) can be modulated by means of the modulator (MOD) and can then be emitted via the reflector antenna (A2).

20. Active reflector according to claim 19, wherein the modulator (MOD) contains an amplifier (AMP).

21. Active reflector according to one of claims 19 or 20, wherein the modulator (MOD) contains a frequency modulator (FMOD).

22. Active reflector according to claim 21, in which the frequency modulator (FMOD) is a mixer module (MIX) by means of which the signal fed to it can be frequency-shifted with a reference signal of the frequency fR.

23. Active reflector according to one of claims 20 to 22, in which a phase shifter (PHS) is present between the directional element (RE) and the amplifier (AMP).

24. Active reflector according to one of claims 19 to 23, in which between an output and a third terminal of the

Straightening element (RE) a delay element (DEL) is present.

25. Active reflector according to claim 24, in which the delay element (DEL) is realized by means of one or more surface components which are applied to a substrate, and on the same substrate a resonator structure for generating a modulation signal (MSIG, fR, PSIG) is applied.