WO2000045459A1

WO2000045459A1 - High frequency filter circuit

Info

Publication number: WO2000045459A1
Application number: PCT/JP2000/000466
Authority: WO
Inventors: Kunio Tochi; Kiyoshi Mizushima
Original assignee: Nikko Company
Priority date: 1999-01-28
Filing date: 2000-01-28
Publication date: 2000-08-03
Also published as: AU2321200A

Abstract

A filter circuit provided on a transmission line thereof with at least one bypass including a dielectric resonator (DRO), characterized in that a capacitor is connected in parallel with the DRO on at least one of the bypasses; whereby a higher-performance (improved out-of-band attenuation and steepness, and suppression of harmonics) and smaller-size DRO type filter circuit can be implemented; and an intermediate frequency band high-performance filter circuit capable of accommodating an increase in information transmission conventionally hardly accommodatable by an SAW filter can be provided.

Description

Specification

High frequency filter circuit Technical field

The present invention relates to a filter circuit, particularly to a high-frequency filter circuit using a dielectric resonator (DRO) as a circuit element, a circuit element including the filter circuit, and a method of manufacturing the filter circuit element. Background art

In recent years, mobile communication systems typified by mobile phones and PHS have rapidly spread, and the demand for small and compact filters having favorable frequency characteristics in the microwave region has been increasing.

Filter circuits that are relatively small and used in this band include laminated LC filters, surface acoustic wave (SAW) filters, and dielectric filters.

The multilayer LC filter is made by printing wiring, inductors (L), and capacitors (C) according to the circuit configuration on a green sheet made of a dielectric material, firing this after lamination. Since the low-pass filter (LPF) of the stacked type has low power consumption, it is indispensable, for example, as a transmission filter inserted between the antenna and the transmitter to remove unnecessary waves (spurious). Also, a high-pass filter

Various filters such as (HPF), band-pass filter (BPF), and band-stop filter (BEP) can be easily configured. However, multilayer LC filters generally have difficulties in reducing manufacturing costs due to complex structures and expensive multilayer materials. In addition, since the dimensions and shape of the internal electrodes before and after firing are unavoidably changed due to shrinkage of firing of the laminated material during manufacturing, there is a problem that there is a limit in improving the design accuracy. On the other hand, S AW fill evening are quartz, lithium niobate (L i N B_〇 _3), Tan lithium Tal acid (L i T A_〇 ₃₎ multiple pairs of interpolation to each other on the piezoelectric substrate or piezoelectric thin film such as An interdigital electrode (interdigital electrode: IDT) is formed by providing a powerful transducer. When an electromagnetic wave signal is input to the SAW filter, it is converted into an acoustic signal by the input-side transducer, propagates on the crystal surface, is converted back into an electromagnetic wave signal by the output-side transducer, and is output. Currently, the SAW filter is generally used in the intermediate frequency band of mobile terminals and base stations for mobile communication because it is easy to reduce the size and weight.

However, in the SAW filter, the propagation speed of the signal wave is basically an elastic wave (at most several thousand m / s depending on the crystal material and propagation mode used), so the SAW filter has various problems in principle. Contains.

First, there is a problem of flatness of the group delay time in the pass band. If there is a difference in the group delay time (signal propagation speed) within the pass bandwidth, all the frequency components of the signal wave cannot be transmitted at the same time. In the S AW filter, the difference in the group delay time between each frequency component is large because elastic waves whose signal propagation speed is fundamentally fundamental are fundamental. However, in the case of the laminated LC type and the dielectric filter, the signal transmission speed is light speed (3 × 10 8 ms) because the signal transmission is an electromagnetic wave, which is orders of magnitude faster than the elastic wave. Therefore, the difference in the group delay time between the frequency components in the signal wave is significantly smaller than that in the SAW filter. The CDMA method is the dominant next-generation mobile communication method. In the CDMA method, signal waves are encoded and spread over the entire bandwidth. Therefore, the flatness of the group delay time becomes more important as the bandwidth is widened. However, it is difficult in principle to achieve the required flatness of the group delay time with the SAW filter.

Second, there is the problem of manufacturing costs. The size of the resonator depends on the resonance wavelength at the operating frequency. Since (wavelength) = (signal wave propagation velocity) Z (frequency), the lower the propagation velocity, the shorter the wavelength and the smaller the size of the resonator. S AW The electrode pattern width at the filter becomes a fine thin film electrode of several meters or less, and the finer the pattern forming technology becomes, the higher the applicable frequency band becomes. The formation of electrode patterns relies on expensive semiconductor manufacturing equipment and is a major factor that hinders cost reduction.

Third, there is the problem of temperature characteristics. In general, a piezoelectric material having a small electromechanical coupling coefficient has excellent temperature characteristics, but a piezoelectric material having a large electromechanical coupling coefficient tends to have poor temperature characteristics. For example, filters for intermediate frequencies require high temperature stability, and currently, quartz substrates with excellent temperature stability are used. However, since the electromechanical coupling coefficient of quartz is relatively small, the size of the element increases when trying to increase the bandwidth. On the other hand, as a material coupling coefficient can realize the required broadband and miniaturization in together largely known L i Nb_〇 _3, there is no application in the size fry real temperature coefficient.

Fourth, there is a problem that it is difficult to improve the out-of-band attenuation in a high frequency band. For example, in a reception filter in a terminal such as a wideband CDMA (using the 2 GHz band), 380M from the center frequency is used to suppress the image signal (image signal).

An out-of-band attenuation of about 70 dB is required at frequencies separated by Hz, but no SAW filter substrate material that satisfies this condition is currently known.

DRO filters are widely used as dielectric filters. As shown in Fig. 2, one or more dielectric resonators (DRO) 2 are placed on a substrate 1 and terminals 3 pulled out from the inner conductor (the inner surface of the resonator hole) of the DRO are connected to the substrate. Connected to the upper electrode 5. The outer conductor (outer surface) of the DRO is coupled to GND.

In the bandpass filter (BPF) shown in Fig. 2, the electrodes 5 are connected to each other on the substrate surface by a coupling capacitor 4a, and are also connected to the electrode 6 at the end of the substrate via a coupling capacitor 4b. . Electrode 6 wraps around the back surface (see Fig. 2 (b)) and forms mounting terminal electrodes surrounded by insulating layer 7 as necessary. I do.

Fig. 3 shows a circuit diagram of a conventional DRO-type BPF. The one-stage filter shown in Fig. 3 (a) has the basic structure, but a multi-stage BPF shown in Fig. 3 (b) is generally used. The DRO-type BPF in Fig. 2 corresponds to the case where n = 2 in Fig. 3 (b). Here, Cc i etc. is the coupling capacitor 4 (4 a,

4b) is the coupling capacity corresponding to

DRO-type BPFs have the advantages of low insertion loss, having an allowable power of several watts or more, and being able to handle high-frequency bands of 1 GHz or more. It is used.

In the conventional DRO-type BPF, the center frequency is almost equal to the resonance frequency of the DRO. For this reason, the center frequency of the BPF can be set relatively freely by adjusting the length of the DR〇.On the other hand, to reduce the size of the BPF at the same center frequency, a dielectric material with a high dielectric constant is applied to the DRO. We have to rely on techniques that improve the electromagnetic wave shortening rate (1Er). However, there are limits to improving the permittivity. For example, when a dielectric material having the highest dielectric constant (approximately 110 in dielectric constant) is currently used in practical use, an intermediate frequency band (IF band) of 500 MHz or less is used. In this case, the length of 0 shaku becomes 14 mm or more, and it cannot be designed with a dimension that is practically effective as a BPF for mobile terminals. For this reason, the applicable band is limited to the high-frequency band where the length of one wavelength is short, in spite of its excellent features.

DRO also requires high Q value and temperature stability. 01 High ぃ (3 values reduce the insertion loss of the BPF, and high temperature stability suppresses the temperature change of the center frequency. Therefore, in order to reduce the size and performance of the DR〇 type BPF, Materials that achieve high dielectric constant, Q value and temperature stability are required, but it is theoretically experienced that it is extremely difficult or impossible to satisfy these three conditions at a high level. It has also been suggested. Furthermore, generation of harmonics is one of the problems of the 0 shaku type 8 PF. That is, in mobile communication equipment, etc., it is a general specification that harmonics are suppressed at least up to three times the center frequency of the BPF. Is low. As a result, for example, in the case of the above-mentioned PHS, etc., an LPF for removing harmonics is required after the BPF, which has reduced the advantage of the DRO-type BPF that the insertion loss is small.

The coaxial DRO shown in Fig. 2 also has manufacturing problems. In other words, DR〇 shown in Fig. 2 is obtained by piercing a resonator hole along the axis in a prismatic dielectric core and firing it, and then sintering the inner surface of the resonator hole and the outer surface of the dielectric core. A conductor layer is provided on the end face (excluding the end face), and one end of a bent conductor bar or plate is inserted into the resonator hole and connected to the conductor layer on the inner surface to form a terminal. However, changes in properties due to shrinkage and deformation during firing are inevitable.

For O, it was necessary to adjust the characteristics by polishing the end face, so mass production was poor, and it was difficult to reduce the price.

Furthermore, the characteristic impedance Z of the DRO. Increasing the diameter is effective in improving the steepness of the BPF characteristic. However, in a coaxial DRO, it is necessary to increase the outer diameter or decrease the inner diameter in order to increase Zo. However, a resonator with a large outer diameter is not practical, and a resonator with a small inner diameter (less than 0.5 mm) is difficult to manufacture. Therefore, the characteristic impedance Z of a practical coaxial DRO. Is about 10Ω, and there is a limit in improving the characteristics of the filter circuit using this.

A structure in which circuit elements are connected to the DRO in series or in parallel has also been proposed. For example, a circuit configuration in which a capacitance is connected in series with a DRO and an attenuation pole is provided on the low frequency side is known (for example, Japanese Patent Application Laid-Open No. 5-102707). Also, in Japanese Patent Application Laid-Open No. 7-147503, a capacity or inductor and a switch are provided in parallel with A BPF that can be used as a duplexer that can switch the center frequency more is described. However, in Japanese Patent Application Laid-Open No. 7-147503, since a switch is provided on the GND side in the capacity, the effect of an increase in GND impedance cannot be avoided, and BPF characteristics cannot be obtained.

As described above, there are problems with all types of filters used in mobile communications, etc., but in the past, 0-short-length type 8 was used for processing 112-band signals, In the frequency band, the problem has been avoided by a combination such as using a SAW filter. However, next-generation communications such as broadband CDMA, MMAC (multi-media mobile access), and large-capacity optical communications are characterized by high-speed information transmission, and it is essential to use higher frequencies and broaden the frequency band. Becomes Therefore, the use of SAW filters is a bottleneck in improving information transmission capacity. Therefore, there is a need for a filter circuit that can be used even in the intermediate frequency band, and has a small size and excellent characteristics.

In addition, with the use of higher frequencies in filters and higher sophistication of required characteristics, conventional inspections have been conducted on all types of filter circuits, including multilayer LC filters, SAW filters, and dielectric filters. Methods are creating the problem that it is difficult to guarantee the required properties.

Specifically, conventionally, a high-frequency filter element has been inspected using a dedicated inspection jig according to the outer shape of the element and the arrangement of the electrode terminals. Therefore, the measurement data includes the jig characteristics (insertion loss, reflection loss, crosstalk loss) and the time variation of measuring instrument accuracy. Until now, these factors have been minimized by improving the characteristics of jigs, and the characteristics of the products themselves have been evaluated by comparison with data based on standard samples. Such measurements are required for the measurement of advanced characteristics required for next-generation communication terminals, such as measurement of large attenuation, measurement of minute reflected power and insertion loss, and measurement in the quasi-millimeter wave band. It is beyond the limits of the method. For this reason, a new inspection method that can sufficiently respond to these advanced characteristics Is required. Disclosure of the invention

The present inventors have made it possible to apply a dielectric filter (DRO type filter) in a wide frequency range including an intermediate frequency range, where it has been considered that the SAW filter is overwhelmingly advantageous. We studied the structure of a DRO-type filter that achieves the demand for lower frequency and smaller size while maintaining good characteristics. As a result, (1) by connecting a capacitor in parallel with DR フィル, it is possible to shift the center frequency of the DR〇 filter to a lower frequency side, which also has the effect of improving the characteristics. (2) In particular, in the case of a DRO-type filter, the terminal inductance of the DRO has a large effect on the high-frequency characteristics, and by resolving this, a great effect can be obtained. (3) When filters are multi-staged, various effects such as suppression of harmonics can be obtained by adjusting the admittance of the DR〇 and the parallel capacitance for each stage. (4) In particular, the above (2) In the circuit configuration where the terminal inductance is shifted to the transmission line based on the knowledge of (3), by adopting the design method of (3), the design method by simulation becomes possible, and various useful characteristics are realized. What you can do, 5) Furthermore, by using a dielectric resonator having a structure in which a dielectric is sandwiched between parallel conductive plates (parallel plate type dielectric resonator), a high characteristic impedance can be easily realized, and the filter is steep. (6) Furthermore, by forming a large number of filter circuits satisfying these characteristics on a substrate at the same time and dividing them into individual filter circuit chips, (7) In the manufacturing method of the above (6), when a large number of filter circuits are formed on the substrate, the inspection is refined by using an air prober for product inspection. · We found that efficiency could be easily realized, and completed the present invention.

That is, the present invention provides the following filter circuit, filter circuit element, and filter Provided is a method for manufacturing a circuit element.

1. A filter circuit comprising at least one bypass including a dielectric resonator (DRO) on a transmission line, wherein at least one of the bypasses has a capacity connected in parallel with the DRO. Filter circuit.

2. The admittance of the DRO included in the bypass connecting the capacitor in parallel with the dielectric resonator (DRO) is set to a value different from the admittance of the DRO included in any of the other bypasses, and The admittance of the capacity connected in parallel with the DRO is set to a value that eliminates the difference between the DR0 admittance and the admittance of each side road is set to the same value as described in 1 above. Phil evening circuit.

3. In a filter circuit having n (n is an integer of 2 or more) bypass circuits including a dielectric resonator (DRO) on the transmission line, k of them (k is an integer of 1 or more and less than n) The admittance of the DRO included in the bypass to which the capacitor is not connected is set to a value different from the admittance of the DRO included in the bypass, and The admittance of the capacity connected in parallel is set to a value that eliminates the difference between the DRO admittances, whereby the admittance of the n side roads is set to the same value. circuit.

4. The terminal of the dielectric resonator (DRO) has a two-branch structure from the end of the DRO, and the two ends generated by the branch are inserted into the transmission line to directly connect the DRO to the transmission line. The filter circuit according to any one of 1 to 3.

5. The capacitor as described in any one of 1 to 4 above, wherein the capacitor connected in parallel with the dielectric resonator (DRO) has a capacity to shift the center frequency of the filter to a low frequency side of 50 MHz or more. Phil evening circuit.

6. The filter circuit according to any one of 1 to 5, wherein the dielectric resonator (DRO) has a structure in which a dielectric is sandwiched between parallel conductor plates.

7. A filter circuit element including the filter circuit according to any one of 1 to 6 above. Child.

8. (1) After dividing lines that define a plurality of filter circuit formation regions are provided vertically and horizontally on the inorganic material substrate, (a) a conductor line connecting the electrodes and the electrodes is formed in each region. (B) mounting a dielectric resonator (DRO) and a capacitor, and (c) connecting a DRO and a capacitor to the electrode to form a bypass including a dielectric resonator (DRO) on a transmission line. Forming, in each of the filter circuit forming regions, a filter circuit having a structure in which at least one bypass path is connected to a capacitor in parallel with DR on at least one side path;

(2) sealing the filter circuit;

(3) A method of manufacturing a filter circuit element, comprising a step of dividing the substrate into individual filter circuit elements along the division line.

9. The method for manufacturing a filter circuit element according to the item 8, including a step of inspecting a plurality of filter circuits formed on the substrate with an aprober after the sealing step and before dividing the substrate.

10. The method for manufacturing a filter circuit element according to the above item 8 or 9, wherein the dielectric resonator (DRO) has a structure in which a dielectric is sandwiched between parallel conductor plates.

BRIEF DESCRIPTION OF THE FIGURES

FIG. 1 is a circuit diagram of a DRO-type BPF according to the present invention.

FIG. 2 is a circuit diagram schematically showing the structure of a conventional DRO-type BPF ((a) is a perspective view, and (b) is a back view).

FIG. 3 is a circuit diagram of a conventional DRO-type BPF.

Fig. 4 (a) is a circuit diagram that does not consider the terminal inductance of the DRO-type BPF, and Fig. 4 (b) is a circuit diagram that considers the terminal inductance. FIG. 5 is a circuit diagram of a DRO-type BPF schematically showing movement of a terminal inductance of DR〇 to a transmission line according to the present invention. FIG. 6 is an end view schematically showing the structure of the split terminal type DR〇. FIG. 7 is a perspective view schematically showing a structure of a split terminal type DR〇.

FIG. 8 (a) is a perspective view of a parallel plate type dielectric resonator that can be used in the present invention, and FIG. 8 (b) is a cross-sectional view of an open-ended parallel plate type dielectric resonator. FIG. 8 (c) is a cross-sectional view of a parallel-plate type dielectric resonator with a short-circuited tip.

FIG. 9 (a) is a schematic diagram illustrating a balanced input / output configuration of a parallel plate type dielectric resonator that can be used in the present invention, and FIG. 9 (b) is a schematic diagram illustrating an unbalanced input / output configuration. .

FIG. 10 is a circuit diagram of a BPF using a parallel plate type dielectric resonator. FIG. 11 is a perspective view of a parallel-plate dielectric resonator of a continuously changing characteristic impedance type.

FIG. 12 (a) shows a conventional circuit configuration of the band rejection filter (BEP), and FIG. 12 (b) shows a corresponding circuit configuration example of the band rejection filter (BEP) of the present invention.

FIG. 13 (a) shows a balanced polar filter circuit having a pole on the low frequency side according to the present invention, and FIG. 13 (b) shows a corresponding unbalanced polar filter circuit. .

FIG. 14 (a) shows a balanced polar filter circuit having a pole on the high frequency side according to the present invention, and FIG. 14 (b) shows a corresponding unbalanced polar filter circuit.

FIG. 15 is a schematic diagram showing an example of a duplexer configured by a combination of the circuit of FIG. 13 and the circuit of FIG.

FIG. 16 is a graph showing the transmission characteristics of a DRO-type BPF according to the present invention.

Fig. 17 is a graph showing the characteristics of DRO-type BPF.

Figure 18 is a graph showing the characteristics of a DRO-type BPF.

FIG. 19 is a graph showing characteristics of a DRO-type BPF according to the present invention. FIG. 20 is a graph showing characteristics of a DR〇 BPF according to the present invention.

FIG. 21 is a graph showing the characteristics of a DR〇 type BPF according to the present invention.

Fig. 22 is a graph showing the characteristics of DRO-type BPF.

Fig. 23 is a graph showing the harmonic characteristics of the DRO-type BPF.

FIG. 24 is a graph showing the harmonic characteristics of the DR〇 BPF according to the present invention.

FIG. 25 is a graph showing the high-frequency characteristics of a multi-stage BPF in which the DR〇 specifications of each stage are changed according to the present invention.

FIG. 26 is a plan view schematically showing a configuration of a microwave band small filter according to the present invention.

FIG. 27 is a plan view schematically showing the configuration of the intermediate frequency band small filter according to the present invention.

FIG. 28 is a graph showing transmission characteristics of a 2.5 GHz optical communication clock extraction filter according to the present invention.

FIG. 29 is a schematic view illustrating a method for manufacturing a filter circuit according to the present invention. Detailed description of the invention

1. Basic structure

The present invention relates to a filter circuit comprising at least one bypass including a dielectric resonator (DRO) on a transmission line, wherein at least one of the bypasses includes a DRO and a DRO as shown in FIG. 1 (a). This is a filter circuit in which a capacity is provided in parallel. That is, in a filter circuit including a DRO in a bypass, a circuit portion including the DRO [when there are two or more such bypasses, the circuit portion in at least one bypass (see the next section). ] Is replaced with the circuit element of Fig. 1 (a). DRO can be either short-circuited or open.

Even if a capacity is provided in parallel with the DRO, the characteristics of the DRO itself do not change, but due to the presence of the parallel capacity, the center frequency F of the fill is set. Is shifted to the low frequency side. To remove. Therefore, according to the present invention, it becomes possible to shorten the resonator length in the target frequency band.

For example, the circuit diagram shown in FIG. 1 (b) is obtained by applying the present invention to the conventional DRO type BPF shown in FIG. 3 (a). Assume that a DRO with a resonance frequency of (= F) and a length of λ is required when constructing a frequency BPF. On the other hand, according to the present invention, when the parallel capacity is provided, the center frequency of the filter shifts to the lower frequency side. Therefore, in order to construct a filter having the center frequency of the pass band in FIG. 1 (b), a DRO having a resonance frequency of f ₂ (> F! = Fi), and thus a length of λ _{2 and} λ 丄 is sufficient. Note that the coupling capacitors C c and C c ₂ are not necessarily required.

When the above configuration is adopted according to the present invention, the effect of lowering the center frequency and the effect of improving the out-of-band attenuation can be obtained. That is, there is an effect of the same property as the increase in the characteristic impedance of the dielectric resonator. Therefore, the characteristics can be improved by applying the present invention to the conventionally used DR〇. In addition, DR〇 composed of a dielectric material that has not been practically used due to poor dielectric constant, Q value or temperature stability can also be used. For example, there are many dielectric materials that have not been put to practical use because of their high dielectric constant and Q value but poor temperature stability.However, such a dielectric material should be applied to DR〇 and temperature compensated by Cs. It is also possible to improve the performance of the DRO BPF. Further, as will be described in an embodiment to be described later, harmonics relatively increase in frequency as the Cs capacitance increases, and the LPF following the BPF required in the conventional DRO-BPF becomes unnecessary.

2. Multi-stage filter circuit

The present invention can be applied to a multistage filter circuit.

That is, in the present invention, in a filter circuit provided with two or more bypasses including a dielectric resonator (DRO) on a transmission line, at least one of the bypasses is provided. And a filter circuit characterized by connecting a capacitor in parallel with the DRO.

For example, FIG. 1 (c) is an application of the present invention to the multi-stage BPF of FIG. 3 (b). As shown, DR〇 ₁ ~ DRO _n in parallel with capacity C s ~. The DRO-type BPF has a multi-stage configuration that is connected via the inter-stage coupling capacity C C1 to _C c _{n + 1} . Fig. 1 (c) shows an example in which capacity is provided on all the sideways, but only k (l≤k≤n-1) of these sideways are parallel to the DRO. You may connect the capacity.

In the multistage filter circuit according to the present invention configured as described above, it is possible to obtain excellent characteristics that cannot be realized by the conventional multistage filter circuit. The following describes the multi-stage BPF shown in Fig. 3 (b) as an example.

In the multistage BPF shown in Fig. 3 (b), the input admittance of each stage Υ] [j is 1 to n. The admittance is the center frequency F. Is the same as the admittance YDROj of the DR〇 of each stage, but if the specification of DR〇 of any stage is different from the others, different values will be mixed in Υ ”and the insertion loss will be Increase. Therefore, it is required that the DRQ of each stage has the same specifications, and that Υ ”, that is, YDRC ^, be all equal.

On the other hand, in the present invention, the input admittance Υ ”of each stage is equal to the sum of the admittance YDROj of each stage DR〇 and the admittance Yes j of the parallel capacitor. Therefore, even if the DRO specification of one of the side roads is different from the others, the conditional expression: Yi-Ys Yg — ^ Yn can be satisfied by changing the specification of the capacity. In other words, it is possible to use different specifications of DRO and parallel capacity at each stage.

Specifically, ① Parallel capacity is provided on all side roads including DRO, and the admittance of DRO (or two or more) of one of the side roads is the admittance of DRO of other side roads. Different values, and the difference in the admittance (2) If some sideways are included (for example, if there are n sideways, one or more and less than n sideways), a capacitor is provided in parallel with the DRO, and the admittance of these DROs Is different from the admittance of the DRO on the side road where no parallel capacity is provided. DRO admittance is also different between the established side roads, and this includes the case where the difference in admittance is canceled by the admittance of the parallel capacity.

As described above, in the present invention, since DROs and parallel capacitors having different specifications can be used in each stage, characteristics that cannot be realized by a conventional multi-stage filter using DROs having the same specifications can be obtained. For example, by using a DRO and a parallel capacitor so that harmonics cancel each other at each filter stage, a BPF circuit in which harmonics are significantly suppressed or eliminated can be obtained. Also, in the conventional DRO-type BPF of Fig. 3, the DRO of each stage has the same specifications, so the pole frequency of the BPF may be changed to a constant value with respect to the passband characteristics. Although it is not possible, if the DRO type BPF circuit according to the present invention uses DR〇 with different specifications for each stage, the pole frequency can be arbitrarily designed. Therefore, adjustment of the out-of-band attenuation and improvement of the steepness of the filter can be realized.

The conditions for this can be determined by simulations such as circuit analysis after setting parameters such as the applied frequency and the number of filter stages.

= Y ₂ = Y ₃ = ~ = Y _n , while the DRO and parallel capacity of different specifications are used in each stage in the filter of the present invention, the steepness adjustment and the It is possible to improve the out-of-band attenuation, etc. In addition, the filter characteristics can be realized as required.

Here, the BPF is described as an example, but the same applies to other types of filter circuits, for example, LPF, HPF, and BEP. By using a DRO and a parallel capacity with the specified specifications, it is possible to adjust steepness and improve the out-of-band attenuation, etc., and to realize fill characteristics as required.

3. DRO (dielectric resonator)

In the present invention, the DRO includes a distributed constant line through which a TEM wave propagates. The line ends may be short-circuited or open, and there is no limit on the number of resonant lines per DRO. The structure of the DRO is not particularly limited except for a terminal structure and a parallel plate type dielectric resonator described later.

The tip of the DRO may be short-circuited or open. However, when the center frequency is shifted to the lower frequency side by about several hundred MHz or more by using the large capacity C s, that is, when the difference between the resonance frequency of the DRO and the center frequency of the BPF increases, as a result, the BPF characteristic becomes It was found that the BPF characteristics deteriorated due to the influence of the inductance of the DRO pin. That is, when the present invention is applied to a filter circuit using a DRO having a conventional terminal structure (for example, FIG. 2), for example, the circuit in FIG. 4 (a) [the circuit in FIG. 1 (b) It is necessary to consider the effect of the terminal inductance L as shown in Fig. 4 (b). (More precisely, the terminal inductance of the capacity is also taken into account.) There is a need) . The influence of the terminal inductance located on the bypass is significant, and even the minute inductance (less than 0.5 nH) given by the 1-mm long wire-pound gold wire has a large effect on the filter characteristics. In particular, the increase in insertion loss is large, and it becomes difficult to apply the design method. As shown in Fig. 4 (b), when the terminal inductance is located on the side road, the design value of the filter and the measured value greatly differ, but the reason has not been analyzed in detail at present. .

In the present invention, by moving the DR ^ terminal position from the bypass to the transmission line, the effect of the terminal inductance L of 01 ^ 0 on the zero-scale 0 characteristic is virtually eliminated, and C s is reduced without deteriorating the characteristic. Of BPF We succeeded in shifting the center frequency to the intermediate frequency range below 500 MHz. In addition, by incorporating the effect of terminal inductance into the inductance on the transmission line, only the DRO and the capacitance need to be considered as bypass parameters, and the design values and the measured values match well. I do.

To move the position of the DRO terminal from the bypass to the transmission line, as shown in Fig. 5, the DRO terminal has a two-branch (split) structure from the end of the DRO, and the branch end is inserted into the transmission line. In this way, DR〇 is connected directly to the transmission line.

For example, as shown in FIG. 6 (a), a DR09 having a branch terminal 8 (8a and 8b) formed in a substantially V shape is used, and an electrode forming a part of a transmission line on a substrate 11 is used. Connect the terminals 8a and 8b to 10a and 10b, respectively. Fig. 6 is an end view of a DRO with a circular cross-section where the inner conductor is installed so as to be substantially parallel to the substrate (support members etc. are omitted.). Other structures are possible as long as the signal is moved from the bypass to the transmission line.

For example, FIG. 7 shows a configuration example using a microstrip line line. In this example, electrode pads 14a and 14b are provided on both sides of the end of the inner conductor 12 and the input side and the output side of the transmission line are connected via wire pounds 18a and 18b, respectively. , I ZO.) The internal conductor 12 on the dielectric layer 16a may be covered with the dielectric layer 16b (the dielectric layer 16b is shown separated from the internal conductor 12 in the figure). In this case, the dielectric layer 16b is shorter by a length (AL) corresponding to the electrode pads 14a and 14b, exposing the electrode pads. The conductor layers 20a and 20b provided on the surfaces of the dielectric layers 16a and 16b, respectively, are connected to ground (GND). Microstrip line is short Entanglement · It may be open.

Further, in the present invention, a parallel plate type DRO can be used, which will be described later.

4. Parallel capacity evening

The center frequency depends on the DRO, coupling capacitance, number of filter stages, etc., but basically shifts to lower frequencies as Cs increases. Capacity of the sub connexion, C s (or 〇 3 _1-5 _11), depending on the fill evening applications, generally 0. 5 pF or more, preferably 1 pF or more, more preferably from about 3 pF or more . Below the lower limit, the shift amount of the center frequency of the BPF is small (at most about ten and several MHz in the GHz band), and the meaning of lowering the frequency is poor. In addition, the effect of improving out-of-band attenuation and the like cannot be sufficiently obtained. The upper limit is not particularly limited.

C s (or 3 ₁ to 3 ₁₁ ) may have any configuration, but is provided as a separate element outside the DRO in order to more effectively obtain the effects of the present invention. For example, the capacity is mounted on the DRO-mounted board as a capacity board, or a multilayer capacity board is applied to the DRO-mounted board to make the capacity Cs an inner layer.

Cs can also have the same branch terminal structure as the DRO. It has been difficult to accurately simulate the influence of the Cs terminal inductance on the bypass, but by moving the terminal inductance to the transmission line, a circuit analysis method using ordinary transmission equations can be used. The filter characteristics according to the design values can be realized.

5. Parallel-plate dielectric resonator

Further, the present inventors have found that a structure 36 in which a dielectric 34 is sandwiched between parallel conductor plates 30 and 32 as shown in FIG. 8 is useful as a dielectric resonator. It should be noted that either the open-ended type in which the bipolar plates are insulated (directly) [Fig. 8 (b)] or the short-circuited type in which the ends are connected [Fig. 8 (c)] may be used. Such a structure is conventionally used as a dielectric resonator. Not.

A parallel plate dielectric resonator (referred to as a “parallel plate dielectric resonator” in this specification) has the following features.

First, a parallel plate type dielectric resonator can easily obtain a large characteristic impedance and can improve the sharpness of the filter. That is, Z. Although the steepness of the filter circuit can be improved by increasing, the characteristic impedance Z that is practical for a coaxial DRO. Was less than 10 Ω, and there was a limit in improving the characteristics. In the case of coaxial type DRO, it is possible to increase Z by increasing the outer diameter or decreasing the inner diameter. It is not impossible to make the inner diameter 10 Ω or more, but if the outer diameter is made larger, it is necessary to reduce the size, but reducing the inner diameter has problems in manufacturing technology and cost. Therefore, it is actually Z of DR〇. There is a limit to the improvement of the steepness of the Phil Evening due to. On the other hand, a parallel plate dielectric resonator can easily obtain a Zo of 10 Ω or more. If a dielectric with a dielectric constant of about 40 is used, 40 Ω or more is possible.

Second, parallel plate dielectric resonators can be manufactured simply by cutting a dielectric substrate of an appropriate length with electrodes on both sides, so mass production is easy. In addition, no characteristic adjustment is required. Furthermore, terminals can be eliminated because they can be mounted directly on the substrate.

Third, the filter circuit can be made balanced by using a parallel plate type dielectric resonator. That is, in general, when two high-frequency lines or devices are connected, not only the characteristic impedance at the connection point but also the distribution of the electromagnetic field must be the same. Currently, balanced power amplifiers, low-noise amplifiers (LNA = Low Noise Amps), and mixers that are driven at low voltage to achieve a wide dynamic range and high gain are being put into practical use. By balancing high-frequency circuits (such as mobile phone transmitters and receivers), a circuit configuration that eliminates the need for MM ICs such as a power switch and a DC ZDC converter for negative voltage generation is possible. Realize compact size and low price. Balanced high-frequency circuits are promising as the next-generation high-frequency circuit technology, and the BPF applied here also needs to be balanced. However, balanced SAW filters are difficult to design and expensive. It is possible to convert the electromagnetic field mode of an unbalanced S AW filter to a balanced type by applying a balanced-unbalanced conversion circuit and apply it to a balanced high-frequency circuit. 5 to 7 are used. Therefore, this method hinders the miniaturization and cost reduction of the balanced high-frequency circuit, and is less practical.

However, as shown schematically in Fig. 9 (a), when a parallel plate type dielectric resonator is used, a balanced input / output configuration can be obtained which is insulated from GND at high frequencies. For this reason, as shown schematically in Fig. 10 (a), it is possible to construct a balanced BPF circuit using a parallel plate type dielectric resonator. In addition, in FIG. 10, the DRO terminal has the above-mentioned divided terminal structure in each stage, as shown by a broken line portion in the first stage.

A balanced circuit can be configured by modifying an unbalanced circuit to have a plane of symmetry. Fig. 10 (b) shows the unbalanced circuit diagram corresponding to Fig. 10 (a). The D RO in the unbalanced BPF may be a coaxial D RO or a parallel plate type dielectric resonator used in an unbalanced input / output configuration [Fig. 9 (b)]. The dimensions of the parallel plate type dielectric resonator are not particularly limited. However, in order to reduce the size of the entire filter, the thickness (distance between the electrode plates) is 1 mm or less, the width is 1 to 10 mm, and the length is 1 to 1 Omm. Preferably, the thickness is about 0.5 mm or less, the width is about 1 to 5 mm, and the length is about 1 to 5 mm. As shown in FIG. 11, the width of the parallel plate type dielectric resonator may be continuously changed in the length direction. Thus, the characteristic impedance Z. A unique DRO with a continuous change in the width direction is obtained. By using a parallel-plate dielectric resonator of continuous impedance type, it is possible to suppress harmonics.

The dielectric material and the electrode material are used in the conventional coaxial DR 同軸, respectively. Material is available. The electrode can be applied to the dielectric substrate by an existing method for applying a conductor layer, such as printing a thick film or forming a thin film.

6. Application to various filter circuits

The present invention can be applied to a filter circuit having a center frequency or a cutoff frequency in the range of several tens of MHz to 100 GHz.

In the above description, BPF is mainly used as an example, but the present invention is applicable to various filter circuits. For example, FIG. 12 (a) shows a conventional circuit configuration of a band rejection filter (BEP), and FIG. 12 (b) shows a corresponding circuit configuration example of a band rejection filter (BEP) of the present invention.

Further, the present invention is not limited to the modification of the conventional filter circuit. Several examples of unique and useful circuit configurations of the present invention are shown in FIGS.

FIG. 13 and FIG. 14 are examples of a polar BPF. The polar filter increases the amount of attenuation in a specific frequency region by providing a resonance point (pole) in the out-of-band region of the filter. Fig. 13 is a circuit with increased low frequency attenuation, and Fig. 14 is a circuit with increased high frequency attenuation. FIGS. 13 (a) and 14 (a) show balanced circuit configurations. In these figures, BL-DRO means a balanced dielectric resonator. Can be used. Figures 13 (b) and 14 (b) show the unbalanced circuit configuration.

Further, the circuits of the present invention can be used in combination. For example, FIG. 15 shows an example of a duplexer composed of a combination of the circuit of FIG. 13 and the circuit of FIG.

In general, a mobile phone operates a reception frequency band and a transmission frequency band close to each other. For example, in the CDMA system (USA), the transmission frequency band is assigned to 824 to 849 MHz, and the reception frequency band is assigned to 869 to 894 MHz. Despite the fact that they are only 20 MHz apart, The signal and the transmitted signal must be separated so as not to cause interference. For this purpose, a duplexer is provided to separate (demultiplex) the received wave from the transmitted wave.

The circuit in FIG. 13 can be used as a receiving filter and the circuit in FIG. 14 can be used as a transmitting filter. Therefore, a duplexer can be configured by combining these circuits. Since the antenna is an unbalanced element, a balanced-unbalanced converter is required in the configuration of Fig. 15 (a) using a balanced BPF.

In the present invention, it is also possible to obtain completely new characteristics by combining the above-described elements. For example, a synchronization signal extraction filter for 2.5 Gbps optical communication, which was conventionally configured by inserting a 50 Ω coaxial DRO on a transmission line, is exemplified. This will be described in detail in an embodiment described later.

7. Manufacturing method of filter element

The element including the filter circuit according to the present invention can be manufactured by a known method, but is preferably manufactured by the method shown in FIG.

In short, simply, a large number of filter circuit elements or circuit elements including filters are formed on one substrate, and individual circuit element pieces are cut out from the aggregate substrate.

In the embodiments shown in these drawings, first, perforations 44 are formed along a boundary 42 that defines each region on the inorganic insulator substrate 40 [FIG. 29 (a)]. Examples of substrate materials include alumina, particularly 96 alumina containing 96% alumina by weight. Inorganic dielectric materials other than alumina can also be used. When a large capacity capacitor is to be formed inside the substrate, a laminated substrate using a material having a high dielectric constant is used. The thickness of the substrate is selected according to the dimensions of the substrate and the required strength. The dimensions and shape of the board are determined by the size of the elements to be mounted.

The perforations can be formed in any manner. For example, drilling with a laser beam Holes, drilling with a drill, and the like. It may be fired after punching the ceramic green sheet.

Next, electrodes and conductor lines 46 are formed on the front surface of the substrate, and electrode terminals are formed on the back surface (not shown) of the substrate (FIG. 29 (b)). If necessary, a conductor may be printed, filled and / or plated in the perforations and used as through holes. A perforation may be provided in the previous step in addition to the perforation for the boundary, and this may be used as a through hole. These electrodes and conductor lines can be formed by, for example, thick film printing. If there is no firing step after electrode formation, highly accurate electrode formation by thin film formation and other metallization methods can be used. The conductor material is a conventional one.For example, in thick film printing, conductive base for thick film such as Ag, Ag-Pt, Ag-Pd, Au, Cu, Ni, etc., thin film formation Examples include, but are not limited to, Au, Cu, Al, Ni, and the like. These techniques may be combined, and a method in which an electroless plating (for example, electroless gold plating) is applied to a thick-film printed electrode is preferably used. Applying electroless gold plating has the effects of (1) improving wire bonding, (2) improving solderability, and (3) improving conductor resistance.

After forming the electrodes, etc., mount DR〇48, coupling capacity 50, and parallel capacity 52 on the board, and if the element contains other components, mount them as well, and connect them by wire bonding, etc. I do. It will be understood from FIG. 29 (c) that the parallel plate type dielectric resonator is particularly advantageous since it can be connected to the substrate without using terminals. As shown in FIG. 29 (c), the parallel plate type dielectric resonator may have a parallel plate mounted perpendicular to the substrate surface, or may be mounted parallel to the substrate.

Next, each substrate region is sealed [FIG. 29 (d)]. The sealing may be performed by resin sealing or a method of bonding a cap member.

The cap is used to house the element when it is brought into close contact with the surface of the substrate. What is necessary is just to have the shape and height which form the space of sufficient dimension. Caps made of inorganic material, organic material, and metal can be used for sealing. Examples of inorganic materials include alumina and quartz glass, and examples of organic materials include plastic and epoxy, and examples of metal materials include iodine, phosphor bronze, and copper.

The cap member only needs to have sufficient strength as a protective member, but if a metal cap is applied, the high-frequency element will be surrounded by a good conductive material, and the cap can be grounded to GND. Almost perfect shielding effect can be obtained. Also, there is a high shielding effect without grounding.

If a metal cap is used, it must be insulated from the conductor tracks on the substrate surface. For insulation between the metal cap and the conductor line, an insulating layer may be provided between the cap and the conductor line by thick film printing or the like. For example, an organic polymer resin layer may be provided on a portion of the substrate to be bonded to the cap to improve the adhesiveness of the cap and to simultaneously insulate the cap.

Thereafter, the module is separated into individual modules packaged on the substrate surface. Separation into individual chips can be performed by a conventional method such as sewing. The method of dividing along the above-mentioned perforations is preferable, and the divided perforations can be used as a conductive portion to the back surface or a side electrode. According to this method, the aggregate substrate is divided while the edge of each substrate region is structurally reinforced by the bonding of the cap member, so that the stress at the time of division tends to concentrate on the division line, and the division is performed for each filter element region. Can be realized more reliably.

In the above description, it is assumed that the collective substrate is not provided with through-holes. Various modifications and changes such as changing the order of the steps, and replacing the operation in each step with a conventional equivalent operation are also included in the scope of the present invention. 8. Filing circuit inspection method

In the above-described manufacturing method, the inspection of the manufactured filter circuit can be made more efficient.

This is done by using a wafer prober as an inspection means for a pre-packaged or packaged filter circuit assembly.

Wafer probers are widely used for precision measurement, intermediate inspection, and shipping inspection of semiconductor products, and include a test head, a probe card, and a stage that can be moved in the XY and vertical directions. The probe force is a replaceable part attached to the test head, and includes a plurality of wires and a plurality of probe pins electrically connected to the wires. In the inspection of a semiconductor wafer by a wafer prober, the semiconductor wafer before being diced is placed on a stage, and the probe pins are brought into contact with the electrodes of the semiconductor products collectively formed on the wafer, so that a current flowing between the electrodes is obtained. Is measured, and defective products are identified by inspection of electrical characteristics.

The inspection method of the present invention is characterized by using a wafer prober, which is known as a semiconductor wafer inspection apparatus, but has not been used for inspection of a high-frequency filter. Specifically, the above-mentioned collective substrate is placed on the stage of a wafer prober with the device mounting area side down, and probe pins are brought into contact with the electrodes on the back surface of the substrate to test individual high-frequency devices. . Before mounting the device, various inspections can be performed on the front side of the board.

As a specific inspection method, a semiconductor wafer inspection method using a wafer prober can be used almost as it is. For example, even when the input / output electrode positions of the filter element or the high-frequency circuit element including the filter are changed in accordance with the desired electrical characteristics, it may be performed by exchanging the probe card or the probe pin or changing the control program. Therefore, it is possible to inspect various products at low cost. Wafer prober inspection processing capacity is 1 per product area even in precision measurement No more than a second. A multi-prober may be applied to simultaneously detect a plurality of high-frequency circuit elements. In this case, the inspection time per product area is reduced to 1/2 to 1/3 seconds.

As described above, in the conventional laminated high-frequency element (or package), inspection required the production of a dedicated jig corresponding to an individual module due to its structure. However, in the method of the present invention, a wafer probe was used. In this case, since the collective substrate is used instead of the wafer, a dedicated jig for the individual module is not required. In addition, since the wafer probe has a movable stage, the large-size transfer equipment used in the conventional automatic inspection of high-frequency circuit elements (or packages) is unnecessary, and the probe can be used semipermanently. It is.

Furthermore, in the wafer probe, the calibration up to the probe tip can be performed by using the calibration substrate. As a result, the intrinsic characteristics of the product can be accurately measured. It is also easy to inspect temperature characteristics. Quality assurance of temperature characteristics has always been a major problem in ceramic electronic components.If inspections other than room temperature were to be performed using automatic inspection equipment that used conventional large-sized transport equipment, the entire inspection system would need to be in a constant temperature Major changes, such as keeping the data, were not practical. However, in general, a wafer probe is a small device that can be carried around and can be calibrated at the probe level such as S-LT (short-open-load-through) calibration. Measurement can be performed easily. BEST MODE FOR CARRYING OUT THE INVENTION

Hereinafter, the present invention will be described more specifically with reference to examples.

Example 1

In the following example, an example in which the characteristics of the present invention are improved by applying the present invention to a DRO having a length of 1 mm, which has been widely used in the past, will be described. DR〇 is outer diameter 4mm x inner diameter 2mm x length 9.3mm Z. (Characteristic impedance): 5.91 Ω Er: 90.0 Using a product with Q value: 169 (Ag 10%), the circuit configuration shown in Table 1 in Fig. 1 (b) (The meaning of symbols such as C _sl C _cl etc.) Obeys the notation in Fig. 1 (b).) The two-stage filter circuit BPF13 and the three-stage filter circuit BPF45 were manufactured. The terminal structure of DR〇 is the same as the conventional DRO-BPF circuit.

The BPF 13 has the same configuration except that the value of Cs is changed to 0 pF 3.0 pF 5.9 pF, and each has a bandwidth of 25 MHz. Similarly, BPFs 4 and 5 have the same configuration except that the value of Cs is changed to 0 pF 2.8 pF, and have a bandwidth of 22 MHz.

For each of these circuits, the center frequency, insertion loss, attenuation, and in-band voltage standing wave ratio (VSWR Peak-Peak value (hereafter, VSWRp-p)) were measured. Table 2 summarizes the results.

As shown in the above results, according to the present invention, the effect of improving the out-of-band attenuation while maintaining the bandwidth is obtained.

Example 2

The following example shows an example in which the present invention is applied to a relatively small (5 mm long) DRO to improve its characteristics.

DR〇 is 2mm outside diameter x 0.7mm inside diameter X length 5.8mm, Z. : 7.41 Ω, Er: 114.4 Using a product with Q value: 142 (Cu 20%), the circuit configuration shown in Table 3 (The meaning of symbols such as C _sl C _cl follows the notation in Fig. 1 (b).) As a result, a two-stage DRO-BPF 67 was manufactured. Table 3

Table 4 summarizes the measurement results of the center frequency, insertion loss, attenuation, and in-band voltage standing wave (VSWRp-p) for each of these circuits. Table 4

As can be seen from the above results, according to the present invention, the effect of shifting the center frequency by about 200 MHz while maintaining the bandwidth and improving the out-of-band attenuation is obtained. In particular, the out-of-band attenuation on the low frequency side has been significantly improved.

Example 3

In the following example, an example is shown in which the center frequency is reduced by using a large capacity Cs.

DRO uses the product of the same specification as in Example 2, Ri by to the circuit configuration shown in Table 5 (means of symbols, such as C _sl, etc., and C _cl follow the notation in FIG. 1 (b).) DRO-BPFs 8 to 12 in a two-stage configuration were manufactured. The BPF was designed so that Cs and Cc would have a bandwidth of several tens of MHz in order to compare the differences in characteristics under the same conditions. However, BPF 9 is an example of a wideband BPF using a conventional terminal. Table 5

Cc capacitance [pF] Cs capacitance [pF]

BPF

C C1 C2 G C3 C S1 C S2

BPF-8 1.7 0.8 1J 0.0 0.0 BPF-9 6.0 2.6 6.0 0.0 0.0 BPF-10 6.0 3.6 6.0 20.0 20.0 BPF-11 6.0 3.9 6.0 30.0 30.0 BPF-12 6.0 4,1 6.0 40.0 40.0

Table 6 summarizes the measurement results of the center frequency, insertion loss, attenuation, and in-band voltage standing wave ratio (VSWRp-p) for each of these circuits.

For the BPFs 10 to 12 (Example), the measurement results of harmonics (FQ ') are also shown. Also shows the pass band characteristic [S ₂₁ (logarithmic value)] a in FIG. 16. In the figure, the bold solid line represents the characteristics of BPF 10, the thin solid line represents the characteristics of BPF 11, and the dashed line represents the characteristics of BPF 12. According to the present invention, the center frequency of the BPF is reduced to 500 MHz or less by connecting Cs of 40 pF, and the DRO having practical outer dimensions (outer diameter 2 mm × inner diameter 4 mm × length 5.8 mm) is used. Use in intermediate frequency band The DR〇 type BPF that can be used has been realized.

Example 4

In the following example, an example is shown in which a large center capacitance C s is used and the center frequency is reduced without deteriorating characteristics by improving the terminal structure of DRO and C s.

DRO uses the product of the same specification as in Example 3, also, by respectively similar to DRO- BPF 8 12 As shown in C _sl C _cl like also Table 7, the DR_〇 as shown in FIG. 5 A DRO-BPF 1317 with the same specifications as in Example 3 was manufactured except that the terminal inductance was moved to the transmission line.

Gs volume: E [pF]

C S1 C S2

0.0 0.0

20.0 20.0

30.0 30.0

40.0 40.0

The center frequency, insertion loss, attenuation, in-band ripple, in-band voltage standing wave ratio (VSWRp-p), harmonics (F. '), F for each of these circuits. 'ZF. Table 8 summarizes the measurement results of the ratio and in-band group delay ripple deviation (sec). FIG. 1721 shows the characteristics of each of the BPFs 1317. In the 17 21 Figure, A S ₂₁ (logarithm: Log Mag S2 1) as passing characteristic of the BPF shows a, B represents the SWR as a standing wave ratio, C is showed DEL AYS ₂₁ as the group delay characteristics . Group delay is the transmission time required for a signal wave to pass through a measurement sample.

When the characteristics of each BPF of the fourth embodiment are compared with those of the third embodiment, the insertion loss of the fourth embodiment is improved, and the attenuation outside the band is remarkably improved. In the fourth embodiment, the pass band characteristics are also improved. Fig. 22 shows the passband characteristics of BPFs 8 and 9 with the same DRO terminal structure as before and without Cs connection (Fig. 22 dashed line: BPF 8, Fig. 22 solid line: BPF 9). Fig. 23 shows the characteristic curves of BPFs 13 and 14 similar to those except that the DRO terminal structure was divided, but when Cs was not connected, the characteristics in the pass band were the same as those of the conventional terminal structure and the divided type. No difference is seen between the terminal structure. In these examples, the resonance frequency of the DRO and the center frequency of the BPF are almost the same. This is because when the inductance of the DRO terminal is located on the side road, the terminal impedance can be ignored because the impedance of DR〇 becomes infinite near the resonance frequency of DR〇. On the other hand, in the frequency range away from the resonance frequency, the effect of the terminal impedance becomes stronger. As a result, the pass band characteristics deteriorate in BPF 10 to 12, but the DR In Example 4 in which the inductance of the O terminal was moved to the transmission line, it is considered that the influence of the terminal impedance was reduced or eliminated and the passband characteristics were improved.

Further, the BPF circuit of the fourth embodiment is also excellent in harmonic characteristics. FIG. 24 shows the harmonic characteristics of the BPFs 13 and 15 to 17 (broken line in FIG. 24: BPF 13, thick solid line in FIG. 24: BPF 15, thin solid line: BPF 16, thick broken line: BPF 17). The harmonic frequency F that appears in the figure. 'And F. The ratio increases with an increase in C s as shown in Table 8. That is, the higher harmonics are used by using the large capacity Cs. In a conventional DRO-type BPF, an LPF is generally used in order to remove a harmonic, but the LPF becomes unnecessary due to the large capacity of Cs in the present invention.

Table 9 shows the main characteristics of BPF 17 according to the present invention in comparison with SAW filters. The SAW filter that is being compared here is a BPF developed for the latest CDMA, with dimensions of 11.5mmX 5. OmmX thickness 1.9mm.

The BPF 17 of the present invention not only has no inferior items in electrical characteristics, but also exhibits excellent characteristics of two or more digits in bandwidth; and three or more digits in group delay characteristics. In addition, BPF 17 has a surface area ratio of about 12, and outperforms S AW Filler in external dimensions. The dielectric material composing the DRO used for the BPF 17 has a dielectric constant of about 110, but if the latest high-frequency dielectric material with a dielectric constant of 200 is used, a smaller BPF can be manufactured. It is possible. Example 5

The following example shows that in a multi-stage BPF in which a capacitor is provided in parallel with a DRO according to the present invention, unique characteristics can be obtained by using different specifications for each DR〇 and capacity.

2-stage fill evening stages in the circuit diagram of FIG. 1 (c), DR_rei_1, using a dielectric resonator having the following specifications as DR02, the coupling capacitor Cc i and Cc ₃ 6. 0 pF, Cc ₂ the 4. l pF, the parallel capacitor C s丄a 4. 0 p F, to produce a BPF 18 to 〇 3 ₂ capacity as 0 (capacity no evening). This is the same as the BPF 17 (Example 4), except that the specifications of the DRO for each stage were changed, and the admittance of each stage was made the same by adjusting the parallel capacity.

DRO 1: Outer diameter 2mmX Inner diameter 0.7mmX Length 5.8mm

Z. : 7.41 Ω, E r: 1 14.4, Q value: 142 DRO 2: outer diameter 2 mm x inner diameter 0.7 mm x length 9.5 mm

_{Z Q: 5. 80 Ω, E} r: 180, ζΗ straight: 120

A graph of the high frequency characteristics of this circuit is shown by a solid line in FIG. In the drawing, the high-frequency characteristics of the BP 17 are also indicated by broken lines.

As shown in the figure, it can be seen that the pole frequency of the BPF 18 has been reduced from 1.5 GHz to 1.1 GHz as compared to the BPF 17.

Mobile phone receivers generally use the superheterodyne system, which generates an image signal in principle. To prevent erroneous operation, the receiving filter must have a characteristic that suppresses the image signal by 70 dB. Due to the strict characteristics, applicable design techniques are limited and many cutting-edge technologies are being used. However, as shown in FIG. 25, if the present invention is applied to a BPF circuit, the pole frequency can be arbitrarily designed. At the pole, the input impedance of the dielectric resonator is almost zero, so that an attenuation close to infinity is obtained, and it is easy to suppress the image signal by 70 dB. Therefore, when the present invention is applied to a BPF circuit, If the DRO and the parallel capacity are selected so that the frequency of occurrence and the frequency of the pole coincide, the image signal can be suppressed despite a simple circuit configuration. In addition, if the parallel capacity is adjusted so that the input admittance of each stage becomes the same at the center frequency of the BPF, the above object can be achieved without deterioration of the passband characteristics such as increase in insertion loss. .

Note that this example shows an example of the characteristics obtained by changing the specifications of the DRO in each stage by the method of the present invention, and other characteristics can be realized as necessary.

Example 6

In the following example, an example of manufacturing a small microwave band filter using a parallel plate type dielectric resonator is shown.

(1) Manufacture of parallel plate type dielectric resonator

Conductive paste (Ag) is applied to both sides of a dielectric substrate (dielectric constant: 110) with a thickness of 5 mm, fired, and then sewn. Manufacture vessels.

(2) Manufacture of microphone fill-band small filter

A rectangular area is provided on a 96mm alumina substrate of 75mm x 75mm, assuming a filter element of 3.8mm in length and width. Through-holes are provided along the boundaries between the regions and at the positions required for the wiring on the front and back, and parallel-plate dielectric resonators, coupling capacitors, and parallel capacitors are mounted on the substrate surface (Fig. 26). Then, a two-stage BPF circuit shown in FIG. 1 (c) is formed.

After that, each area is sealed with resin, and the terminals of the prober are sequentially brought into contact with the electrode terminals on the rear surface, and the filter characteristics are inspected. After 100% inspection, the device is divided along the division line to obtain a microwave band small filter device.

Microwave band small filter with excellent out-of-band attenuation of 70 dB at center frequency 2.14 GHz, bandwidth 60 MHz, and 2.52 GHz by the above manufacturing method Properties can be manufactured. The insertion loss is less than I dB and the group delay time is less than Ins.

Example 7

In the following example, an example of manufacturing a small filter in the intermediate frequency band using a parallel plate type dielectric resonator is shown.

(1) Manufacture of parallel plate type dielectric resonator

Conductive paste (Ag) is applied to both sides of a dielectric substrate (dielectric constant: 110) with a thickness of 1.0 mm, fired, and then sewn to form a parallel plate type dielectric resonator with a width of 1. OmmX and a length of 6.5 mm. To manufacture.

(2) Manufacture of small-sized intermediate frequency band filters

In the circuit shown in Fig. 1 (c), the capacity of the parallel capacity is required to be about 800 pF in order to manufacture an intermediate frequency band filter with a center frequency of about 80 MHz. Therefore, a multilayer substrate with an internal parallel capacitor is manufactured by sintering a high dielectric constant ceramic material (dielectric constant 180) at low temperature, and the composite capacitor and the parallel plate type dielectric of (1) are formed on this substrate. A resonator is mounted (Fig. 27). The manufacturing method and the inspection method are the same as in the sixth embodiment.

By adopting the parallel capacity and applying the parallel plate type dielectric resonator in the present invention, an intermediate frequency band filter smaller than the SAW filter can be realized. Specifically, excellent characteristics with a center frequency of 210 MHz, a bandwidth of 1.3 MHz, and an attenuation of 35 dB can be realized. In addition, as a result of signal propagation replacing slow surface acoustic waves with light-speed electromagnetic waves, phase linearity and group delay characteristics are improved by two orders of magnitude compared to the current state-of-the-art S AW filter.

Example 8

The following example shows an example of manufacturing a 2.5 GHz optical communication clock extraction filter using a parallel-plate dielectric resonator with a large characteristic impedance. In addition, by using a parallel plate type dielectric resonator with a short-circuited tip in this frequency band, When the filter for extracting the hook is designed, the length of the resonator becomes short at 2.5 mm, making it difficult to handle. Therefore, an open-ended parallel plate resonator was used.

A 2mm wide x 5.5mm parallel plate dielectric resonator (characteristic impedance: 30Ω) is cut by applying electrodes to both sides of a lmm-thick dielectric substrate (dielectric constant: 40) and cutting it. Was manufactured.

A single-stage BPF (BPF 19) shown in Fig. 1 (b) was constructed by mounting the above-mentioned parallel plate resonator, coupling capacitance and parallel capacitor on the substrate. Here, the combined capacity Cc i and Cc ₂ were both 0.16 pF, and the parallel capacity 5 was 7.9. FIG. 28 shows the transmission characteristics.

The obtained filter has the required characteristics (center frequency: 2.5 GHz, insertion loss: 8 d) with a volume of 1/100 (thickness: 2 mm, width: 4 mm, length: 7 mm) of the conventional product. B or less, reflection loss: 13 dB or less, bandwidth (Q value): 600 or more, temperature characteristic: 2.5 ppm / t :).

Optical digital signals are greatly distorted by medium- and long-distance transmission of optical fibers. Therefore, in the receiving unit, after conversion from the optical digital signal to the electric digital signal, only the synchronization signal is extracted by BPF and the waveform is shaped. This filter requires narrow-band characteristics and high temperature stability. Conventionally, a coaxial resonator and a dedicated filter circuit have been used, but the external dimensions are large because the resonator is long. The synchronous signal extraction filter according to the present invention is a filter that solves such a problem. Industrial applicability

The present invention achieves high performance (for example, improvement of out-of-band attenuation and steepness, suppression of harmonics) and downsizing of a DRO filter circuit. Therefore, it is useful in various filter circuits and circuits and devices including the filter circuit in the GHz band. In addition, the present invention provides a DRO-type filter circuit that does not cause a product size problem even in an intermediate frequency band where SAW filters have been overwhelmingly superior. Enable application. Therefore, it is possible to provide a high-performance filter circuit that is suitable for use in terminals and base stations in mobile communications such as mobile phones, and that can respond to an increase in the amount of information transmission that is expected to be difficult with a SAW filter. Becomes

Claims

The scope of the claims

1. A filter circuit comprising at least one bypass including a dielectric resonator (DRO) on a transmission line, wherein at least one of the bypasses has a capacity connected in parallel with DR〇. Phil evening circuit.

2. The admittance of the DRO included in the bypass connected to the capacitor in parallel with the dielectric resonator (DRO) is set to a value different from the admittance of the DR〇 included in any of the other bypasses, and By setting the admittance of the capacity connected in parallel with the DRO to a value that eliminates the difference between the DR〇 admittance, the admittance of each bypass is set to the same value. The filter circuit as described.

3. In a filter circuit with n (n is an integer of 2 or more) bypasses including a dielectric resonator (DRO) on the transmission line, k filters (k is an integer of 1 or more and less than n) are used. Connect the capacity in parallel with the DRO on the side road, and set the admittance of the DRO included in the side road to which the capacity is connected to a value different from the admittance of the DRO included in the side road that does not connect the capacity to the DRO. The admittance of the capacity connected in parallel is set to a value that eliminates the difference between the DRO admittances, so that the admittances of the n bypasses are set to the same value. The filter circuit as described.

4. The terminal of the dielectric resonator (DR〇) has a two-branch structure from the end of the DRO, and the two ends generated by the branch are inserted into the transmission line to directly connect the DRO to the transmission line. The filter circuit according to any one of claims 1 to 3.

5. The capacitor according to any one of claims 1 to 4, wherein the capacitor connected in parallel with the dielectric resonator (DRO) has a capacity to shift the center frequency of the filter to a lower frequency side by 50 MHz or more. Filler circuit described in Crab.

6. A dielectric resonator (DRO) with a dielectric sandwiched between parallel conductor plates The filter circuit according to any one of claims 1 to 5, wherein the filter circuit has a structure.

7. A filter circuit element including the filter circuit according to any one of claims 1 to 6.

8. (1) After dividing lines that define a plurality of filter circuit formation regions are provided vertically and horizontally on the inorganic material substrate, (a) a conductor line connecting the electrodes and the electrodes is formed in each region. (B) mounting a dielectric resonator (DRO) and a capacitor; and (c) connecting a DRO and a capacitor to the electrode to form a bypass including the dielectric resonator (DRO) on the transmission line. Forming, in each of the filter circuit forming regions, a filter circuit including at least one and having a structure in which a capacitor is connected in parallel with a DRO on at least one side path;

(2) sealing the filter circuit;

9. The method for manufacturing a filter circuit element according to claim 8, further comprising a step of inspecting a plurality of filter circuits formed on the substrate with an aprober after the sealing step and before dividing the substrate.

10. The method according to claim 8, wherein the dielectric resonator (DRO) has a structure in which a dielectric is sandwiched between parallel conductive plates.