WO1999044281A1

WO1999044281A1 - Method of driving ultrasonic motor

Info

Publication number: WO1999044281A1
Application number: PCT/JP1999/000888
Authority: WO
Inventors: Yoshiyo Wada
Original assignee: Star Micronics Co., Ltd.
Priority date: 1998-02-27
Filing date: 1999-02-25
Publication date: 1999-09-02

Abstract

A single transformer (T) comprises a common core constituting a closed circuit, and primary and secondary windings (Cai, Cbi) and (Cao, Cbo) of the respective phases. Driving signals of the phases are inputted to the respective primary windings (Cai, Cbi). In the transformer (T), the signals interfere with each other, and the interference influences the transformer output voltage of the secondary windings (Cao, Cbo) supplied to an oscillator (5v). However, the influence is so small that the voltages of the respective phase are slightly different from each other, and the predetermined phase difference is maintained. Therefore, without causing any practical problem, the ultrasonic motor can be well driven, and the number of transformers, which are large and most expensive in the drive system of the ultrasonic motor, can be decreased, reducing the cost and size.

Description

Akitoda

Ultrasonic motor driving method

The present invention relates to a method for driving an ultrasonic motor. Background art

The conventional ultrasonic motor includes a stay (stator) composed of an elastic body and a piezoelectric element (vibrator) provided on the back surface of the elastic body, and a pressurizing arrangement opposed to the stay. A rotor (movable element) made of an elastic body, a drive circuit that generates and outputs two-phase drive signals having different phases (for example, a sinusoidal voltage signal and a cos-wave voltage signal having a 90 ° phase difference), By supplying the two-phase drive signal to each of the two regions where the piezoelectric element is divided, the piezoelectric element is vibrated to generate a traveling wave on the surface of the stay. The rotor is driven to rotate by the traveling wave.

In the above ultrasonic motor, the voltage of the two-phase drive signal supplied to the piezoelectric element is increased to 5 to 24 V DC by the power amplifier in the drive circuit, but the drive signal is 15 V Since a voltage of 0 V to 300 V is required, a step-up transformer is required after the power amplifier in the drive circuit.

In this case, a configuration may be considered in which the drive signal of each phase is input to a single transformer to boost the voltage.However, since the drive signals of each phase interfere with each other in the single transformer, they are not suitable for actual use. Conventionally, transformers having the same drive signal transmission characteristics are provided for each phase. Disclosure of the invention

Thus, the conventional ultrasonic motor requires two transformers. Since the transformer is the most expensive in the drive system of the ultrasonic motor, there was a problem that the cost was increased.

Also, since the transformer itself is large, there is also a problem that miniaturization cannot be achieved if two transformers are used.

The present invention has been made to solve such a problem, and an object of the present invention is to provide a driving method of an ultrasonic motor that can drive an ultrasonic motor satisfactorily while reducing cost and size. And

In the method of driving an ultrasonic motor according to the present invention, two-phase drive signals having different phases are supplied to a vibrator provided on the back surface of the elastic body to vibrate the vibrator, and the vibration is generated on the surface of the elastic body. An ultrasonic motor drive method in which a mover is driven by a traveling wave, in which two-phase drive signals having different phases are applied to a common core constituting a closed magnetic circuit, and a primary and a secondary winding of each phase are provided. Are input to the respective primary windings of each phase of the single transformer wound, and are boosted to be supplied to the vibrator from the secondary windings of the respective phases. According to such an ultrasonic motor driving method according to the present invention, a single transformer in which the primary and secondary windings of each phase are wound on a common core forming a closed magnetic circuit, Since the drive signal of each phase is input to the primary winding, the drive signal of each phase interferes with each other in the single transformer. The effect of this mutual interference is that even when the transformer output voltage supplied to the vibrator is at a slightly different level in each phase, a relationship having a predetermined phase difference is maintained, so that there is no problem in actual use. Without driving the ultrasonic motor. As a result, the number of the most expensive and large transformers in the drive system of the ultrasonic motor is reduced. Here, in accordance with the ratio of the voltage at which one of the two-phase drive signals having different phases in the single transformer interferes with the other drive signal, the step-up ratio by the winding in the single transformer is determined. It is preferable that the output voltages of the two-phase drive signals supplied to the vibrator be made substantially the same by making each phase different. In particular, the output voltages of the two-phase transformers supplied to the vibrator differ over a predetermined range, and If it is considered that this will adversely affect the operation, it is preferable to perform correction according to the ratio of the interference voltage in a single transformer as described above.

Also, in order to invert the driving direction of the ultrasonic motor (rotation direction when the mover is a rotating body), this input is used to invert the input phase of the drive signal to the primary winding of a single transformer. In accordance with the inversion of the phases, it is necessary to interchange the boosting ratios that are different for each of the above-described phases so that the output voltages of the two-phase drive signals supplied to the vibrator are substantially the same.

A drive signal for the primary winding of a single transformer is a pulse wave signal, and a sinusoidal drive signal is applied to the vibrator by the resonance action of the capacitance of the vibrator and the inductance of the single transformer. Preferably, it is supplied. Such a pulse wave signal supply circuit has a more practical configuration when configuring the circuit. BRIEF DESCRIPTION OF THE FIGURES

FIG. 1 is a configuration diagram showing an ultrasonic motor employing the ultrasonic motor driving method according to the present invention.

2A and 2B are configuration diagrams showing a single transformer, FIG. 2A is a configuration diagram showing a two-phase drive three-leg transformer, and FIG. 2B is a configuration diagram showing a two-phase drive two-leg transformer. FIG.

FIGS. 3A to 3D are explanatory diagrams showing connection polarities of two-phase windings in a single transformer, and FIGS. 3E to 3H correspond to FIGS. 3A to 3D, respectively. It is a transformer output voltage waveform diagram.

FIGS. 4A to 4D are explanatory diagrams of the internal operation of the transformer corresponding to FIGS. 3A to 3D, respectively, and FIGS. 4E to 4H are transformers respectively corresponding to FIGS. 4A to 4D. FIG. 4 is a vector explanatory diagram of an output voltage.

FIGS. 5A to 5C are diagrams for explaining the connection polarity of windings in a one-phase transformer. 5A is a configuration diagram showing a one-phase three-leg transformer, FIG. 5B is an explanatory diagram showing the addition polarity of the winding, and FIG. 5C is an explanatory diagram showing the subtraction polarity of the winding.

6A to 6E are configuration diagrams showing a single transformer prepared for the experiment. FIG. 7 is a table showing measured mutual inductances, coupling coefficients, and leakage ratios of the transformers in FIGS. 6A to 6E.

FIG. 8 is a diagram showing the relationship between the coupling coefficient and the leakage ratio obtained in FIG.

FIG. 9 is a chart showing measured values of the output voltages of the respective phases of the transformers shown in FIGS. 6A to 6E and calculating and calculating a ratio of a difference between the measured output voltages.

FIG. 10 is a diagram showing the relationship between the output voltage difference ratio and the leakage ratio obtained in FIG. FIG. 11 is a configuration diagram showing a main part of a drive circuit to which a circuit for correcting in a direction to reduce the output voltage difference is added.

FIG. 12 is a configuration diagram illustrating a main part of a drive circuit including a circuit that corrects the output voltage difference in a direction to reduce the difference.

FIGS. 13A to 13P are timing charts for explaining the circuit operation of FIG.

FIG. 14 is a cross-sectional view showing the ultrasonic motor main body. BEST MODE FOR CARRYING OUT THE INVENTION

Hereinafter, a preferred embodiment of a method for driving an ultrasonic motor according to the present invention will be described with reference to the accompanying drawings. In the description of the drawings, the same elements or elements having the same functions will be denoted by the same reference symbols, without redundant description.

FIG. 1 is a configuration diagram showing an ultrasonic motor employing the ultrasonic motor driving method according to the present invention. This ultrasonic motor includes an ultrasonic motor main body 1 composed of a mechanical drive mechanism and a drive circuit 2 for driving the ultrasonic motor main body 1.

First, the configuration of the ultrasonic motor main body 1 will be described with reference to FIG. The ultrasonic motor main body 1 penetrates the center of the fixing base 3 and is fixed to the fixing base 3 The rotating shaft 4 rotatably supported by the bearings 7 provided, the outer peripheral circular opening 6 press-fitted and fixed to the rotating shaft 4, and the outer rim fixed to the fixing base 3 facing the rotor 6 It has a circular stay 5 and. The rotor 6 is rotated by a traveling wave traveling on the surface of the stator 5 (the upper surface in the figure) and traveling in the circumferential direction (details will be described later).

The stay 5 includes an annular vibrator 5 V made of a ceramic piezoelectric element, and an annular elastic body 5 e made of metal with the vibrator 5 V adhered to the outer periphery of the back surface. . An inner peripheral portion 5i of the elastic body 5e is fixed to the fixing base 3 by, for example, screwing, and an annular intermediate portion 5m between the outer peripheral portion 5o and the inner peripheral portion 5i is thin. In addition, the intermediate portion 5 m facilitates vibration of the outer peripheral portion 5 o and suppresses vibration transmission between the inner peripheral portion 5 i and the outer peripheral portion 5 o.

The rotor 6 is formed of an annular elastic body made of metal. The inner peripheral portion 6i of the rotor 6 is press-fitted and fixed to the rotary shaft 4, and the rotor 6 has an annular thin intermediate portion 6m between the outer peripheral portion 6o and the inner peripheral portion 6i.

A flange portion 4f is provided on the upper portion of the rotary shaft 4, and a snap ring 9 such as a C-ring is mounted on a lower portion thereof. A snap ring 9 is provided between the snap ring 9 and the bearing 7. The compression spring 11 is inserted via the spacer 10. The rotating shaft 4 and the mouth 6 are constantly urged downward by the compression spring 11 and the flange portion 4f.

The mouth 6 has a narrow annular convex portion 6 p on a surface of the outer peripheral portion 6 o facing the outer peripheral portion 5 o of the stay 5. When the elastic bodies 5e and 6 are brought into full contact, the vibration on the excitation side (stay side) is transmitted to the other side (rotor side) and no traveling wave is formed, but the elastic body 6 has convex sections 6p. And a traveling wave can be formed in the elastic body 5e.

An annular cushioning friction member 12 made of, for example, resin is interposed between the rotor side convex portion 6p and the outer peripheral portion 5o on the stay side, and the cushioning friction member 12 is brought into pressure contact with them. Ah You. That is, the rotor-side convex portion 6p presses the outer peripheral portion 5o on the stay side via the buffer friction member 12 by the elastic force of the compression panel 11. The cushioning friction member 12 is fixed to one of the convex portion 6p and the outer peripheral portion 5o on the stay side with an adhesive or the like, for example. To generate a normal traveling wave in the stay side elastic body 5e, and to avoid direct contact between metals (convex part 6p and outer peripheral part 5o). To prevent the generation of abnormal noise, and further improve the durability of the press contact part. The cushioning friction member 12 may not be fixed to either the convex portion 6p or the outer peripheral portion 5o, but may be interposed between the convex portion 6p and the outer peripheral portion 5o.

The piezoelectric vibrator 5V has regions divided into a plurality in the circumferential direction, and has been subjected to polarization processing in advance so that the polarization directions in the thickness direction are opposite to each other in adjacent regions. As shown in Fig. 1, these areas are divided into two parts, the sin side electrode part SS and the cos side electrode part CC, and the feedback between the sin side electrode part SS and the cos side electrode part CC. Detection electrode section FB is provided. The sine side electrode portion S S and the cos side electrode portion C C of the piezoelectric vibrator 5 V are connected to the drive circuit 2 as shown in FIG. The drive circuit 2 applies a sinusoidal voltage signal (sinusoidal voltage signal; one drive signal) between the ground and the sinusoidal electrode SS of the piezoelectric vibrator 5 V, and connects the ground with the cos-side electrode CC. During this time, a sinusoidal voltage signal (cos-wave voltage signal; the other drive signal) having a phase difference of 90 ° in electrical angle is applied simultaneously. As a result, the piezoelectric vibrator 5 V vibrates, and a traveling wave traveling in the circumferential direction due to the synthetic wave is generated on the surface of the outer peripheral portion 5 o on the stay side, and the traveling wave is generated by the stator 5. The configuration is such that the rotor 6 pressed against the rotor is driven to rotate.

Next, the drive circuit 2 will be described in more detail. As shown in FIG. 1, the driving circuit 2 generates an sinusoidal voltage signal and outputs the sinusoidal voltage signal, and shifts the sinusoidal voltage signal output from the oscillator 20 by 90 ° in electrical angle. Then cos Characteristics that amplify each voltage signal (two-phase drive signal with a different 90 ° phase) output from the 90 ° phase shifter 21 and the oscillator 20 and the 90 ° phase shifter 21 1 output as a wave voltage signal The power amplifiers Amp 1 and Amp 2, which are equal to each other, and the respective voltage signals output from the power amplifiers Amp 1 and Amp 2 are input and boosted, and the boosted voltage signals are converted to the sine of the piezoelectric vibrator 5 V. And a single transformer T that supplies each to the side electrode section SS and the cos side electrode section CC. The drive circuit 2 also includes a switch SW1 connected between the outputs of the oscillator 20 and the 90 ° phase shifter 21 and the inputs of the amplifiers Amp1 and Amp2, and has a single connection with the output of the power amplifier Amp1. And a switch SW2 connected between the two terminals of the primary winding Cai corresponding to the sin-side electrode section SS of the transformer T.

The switch SW 1 converts the sinusoidal voltage signal from the oscillator 20 and the cosine wave voltage signal from the 90 ° phase shifter 21 to the power amplifiers Amp 1 and Amp 2 to supply the sinusoidal electrodes SS and cos The phase relationship between the voltage signals supplied to the side electrode sections CC is exchanged with each other, and the rotation direction of the rotor 6 is reversed.

The switch SW2 inverts the voltage signal output from the power amplifier Amp 1 by 180 ° without changing the phases of the sin wave voltage signal and the cos wave voltage signal input to the power amplifiers Amp 1 and Amp 2. The rotation direction of the rotor 6 is reversed by supplying the primary winding Cai corresponding to the sin-side electrode section SS of the single transformer T.

2A and 2B show specific examples of the single transformer T. FIG. 2A is a configuration diagram showing a two-phase drive three-leg transformer, and FIG. 2B is a two-phase drive two-leg transformer. FIG.

This single transformer T is composed of a common core FC in which a vertically divided light core is connected via an insulating spacer Z, and an insulating switch in one of the outer iron parts (left side in the drawing) of the core FC. It is wound below the switch Z and connected to the switch SW2. The next winding Cai, the secondary winding Cao wound above the insulating spacer Z on one of the outer iron parts and connected to the above-mentioned sin side electrode part SS, and the other (right side in the drawing) of the outer iron part The primary winding Cbi wound below the insulating spacer Z in the above and connected to the power amplifier Amp 2, and the c above wound above the insulating spacer Z in the other of the outer iron parts. And a secondary winding Cbo connected to the 0s side electrode section CC. The gap d1 formed by the insulating spacer Z is used to control the magnetic flux of the transformer T, the amount of magnetic flux leakage that interferes with the windings of other phases (to be described in detail later), and the parameters for determining the inductance. It is evening.

Next, the output voltage of each phase of the transformer T will be described. 3A to 3D are explanatory diagrams showing the connection polarities of the two-phase windings Cai, Cbi, Cao, and Cbo at the transformer T. FIGS. 4A to 4D are FIGS. 3A to 3D. 4E to 4H are vector explanatory diagrams of transformer output voltages respectively corresponding to FIGS. 4A to 4D. For convenience of explanation, the voltage signal input through the primary winding Cai wound on one of the outer iron parts is a in, the voltage signal output through the secondary winding Cao is a, The voltage signal input through the primary winding Cbi wound on the other side is b in, the voltage signal output through the secondary winding C bo is b, and the voltage conversion signal is generated according to the input voltage signal a in according to the winding conversion ratio. The conversion voltage signal to be generated is denoted by ao, and the conversion voltage signal originally generated by the winding conversion ratio according to the input voltage signal bin is denoted by bo.

In FIG. 3A, the switches SW1 and SW2 are switched to the positions indicated by the solid lines in FIG. At this time, since the converted voltage signals ao and b0 originally generated by the winding conversion ratio are output voltage signals having a 90 ° phase difference, a vector diagram as shown in FIG. 4A is obtained.

In the state shown in FIG. 3A, it is assumed that voltage signals a in and b in having phases different by 90 ° are input to the transformer T through the primary windings Cai and Cbi, respectively. Here, for example, using the transformer T shown in FIG. 2A, if the input voltage signal a in is input to the transformer T through the next winding C ai, the core generated by the input voltage signal a in The magnetic flux in the FC is wound by the windings Cai and Cao through the central leg and the leg on which the windings Cbi and Cbo are wound (the right leg in the figure; hereinafter called the b-side leg). A loop (closed magnetic path) leading to the leg (the left leg in the figure; hereafter referred to as the a-side leg) is formed, and the voltage signal a is output to the sin-side electrode SS through the secondary winding Cao. You.

Among the magnetic fluxes generated by the input voltage signal ain, the magnetic flux that enters the b-side leg is a magnetic flux that is originally not desired to enter (a magnetic flux that interferes with the windings Cbi and Cbo). The smaller the flux, the less magnetic flux enters the b-side leg. Also, the magnetic flux generated by the input voltage signal bin also enters the a-side leg. In the present embodiment, a voltage component (a voltage component in which one drive signal of the two-phase drive signals interferes with the other drive signal in the transformer T) entering the other leg is defined as a leakage amount in the present embodiment. .

Here, assuming that the phase of the input voltage signal ain at the b-side leg is 180 ° out of phase with the output voltage signal b, it interferes with the converted voltage signal bo as the leakage voltage signal a 'shown in Fig. 4A. I do. Similarly, assuming that the leakage of the input voltage signal bin at the a-side leg is in phase with the output voltage signal a, it interferes with the converted voltage signal ao as the leakage voltage signal b 'shown in FIG. 4A.

That is, based on the above assumption, as shown in Fig. 4A, the electromotive forces a ', b, generated by the magnetic flux interference have a phase relationship of 90 ° with respect to the converted voltage signals ao, bo.

Therefore, the output voltage signals a and b are a combination of the vectors shown in FIG. 4A, and the output voltage signal a is assumed to be output as ao + b ′ as shown in FIG. Is assumed to be output as bo—a '.

Next, as shown in FIG. 3B, the polarity of the primary winding Cai supplying the input voltage signal ain is inverted by 180 ° and connected to the connection polarity state 1 shown in FIG. 3A, and in this state, When the input voltage signals ain and bin are applied, as shown in FIG. 4B, the converted voltage signal ao and the leakage voltage signal a are inverted by 180 ° with respect to FIG. 4A. Therefore, the output voltage signals a and b are a combination of the vectors shown in FIG. 4B, and the output voltage signal a is assumed to be output as ao−b ′ as shown in FIG. 4F. b is assumed to be output as bo + a '.

Next, as shown in Fig. 3C, the polarity of the primary winding Cbi for supplying the input voltage signal bin is inverted by 180 ° and coupled to the connection polarity state 1 shown in Fig. 3A. When the input voltage signals ain and bin are applied, as shown in FIG. 4C, the converted voltage signal bo and the leakage voltage signal b ′ are 180 ° out of phase with respect to FIG. 4A.

Therefore, the output voltage signals a and b are a combination of the vectors shown in FIG. 4C, and the output voltage signal a is assumed to be output as ao−b ′ as shown in FIG. 4G. Is assumed to be output as bo + a '.

Next, as shown in FIG. 3D, for the connection polarity state 1 shown in FIG. 3A, the primary winding C ai supplying the input voltage signal a in and the primary winding C bi supplying the input voltage signal b in When both polarities are inverted by 180 ° and coupled, and the input voltage signals ain and bin are applied in this state, as shown in Fig. 4D, the converted voltage signal ao and the leakage voltage signal a ', the converted voltage signal bo and Both leakage voltage signals b and 180 are 180 ° out of phase with respect to Fig. 4A.

Therefore, the output voltage signals a and b are a combination of the vectors shown in FIG. 4D, and the output voltage signal a is assumed to be output as ao + b ′ as shown in FIG. 4H, and the output voltage signal b is , Bo—a 'are assumed to be output.

The inventor has confirmed this by an experiment. FIGS. 3E to 3H are transformer output voltage waveform diagrams measured under the connection polarities shown in FIGS. 3A to 3D, respectively. A voltage signal (drive signal) of 32 kHz was used.

With the connection polarity shown in Fig. 3A, as shown in Fig. 3E, the voltage levels of the output voltage signals a and b are 4.8 V (p-p) and 3.6 V (p-p), respectively. Was.

In the connection polarity shown in Fig. 3B, as shown in Fig. 3F, the voltage levels of the output voltage signals a and b are 3.6 V (p-p) and 4.8 V (p-p), respectively. Met. With the connection polarity shown in Fig. 3C, the voltage levels of the output voltage signals a and b are 3.6 V (pp) and 4.8 V (pp), respectively, as shown in Fig. 3G. Was.

In the connection polarity shown in Fig. 3D, as shown in Fig. 3H, the voltage levels of the output voltage signals a and b were 4.8 V (p-p) and 3.6 V (p-p), respectively.

As described above, although the voltage levels of the output voltage signals a and b are slightly different, it has been confirmed that the phase relationship between the output voltage signals a and b does not differ from 90 ° as shown in FIGS. 3E to 3H. Was.

Incidentally, in the vector calculation formulas described in FIGS. 4E to 4H, since there is a relationship of ao = bo and a ′ = b ′, for example, the voltage levels a and b in FIG. Is substituted into the corresponding vector equation a = ao + b ', b = bo-a' explained in Fig. 4E, ao = bo = 4.2 V, a, = b, = 0.6 V is obtained.

In Fig. 4F, a = ao-b ', b = bo + a', so substituting the values obtained above gives a = 3.6V, b = 4.8V, which corresponds to Fig. 4F. It was confirmed that it matched the level of 3F.

In Fig. 4G, a = ao-b 'and b = bo + a'. Substituting the values obtained above gives a = 3.6V and b = 4.8V, which corresponds to Fig. 4G. It was confirmed that it matched the level in Fig. 3G.

In Fig. 4H, a = ao + b 'and b = bo-a'. Substituting the values obtained above gives a = 4.8V and b = 3.6V, which corresponds to Fig. 4H. It was confirmed that the level matched the level in Fig. 3H. Thus, since the results of the vector calculations based on FIGS. 4E to 4H agree with the experimental results shown in FIGS. 3E to 3H, the assumptions in FIGS. The assumption that the electromotive forces a 'and b' generated by the magnetic flux interference have a phase relationship of 90 ° with respect to the converted voltage signals ao and bo seems to be correct.

By the way, as described above, the amount of leakage is a voltage component that enters the other leg, In the present embodiment, the conversion voltage signal ao and the leakage voltage signal a, a ′ / ao, are used as the leakage ratio (one drive signal of the two-phase drive signal is replaced by the other drive signal). (Interference voltage ratio).

Next, the relationship between the leakage ratio and the transformer output voltage signals a and b will be described. Since the amount of leakage mentioned above is the degree of coupling between the windings that affects the other leg, it can be known from the inductances of the windings of the a leg and b leg and the mutual inductance of both. it can.

For example, in a one-phase drive three-leg transformer as shown in Fig. 5A, the winding inductance at the a-side leg is L1, and the winding inductance at the b-side leg is L2, as shown in Fig. 5B. Assuming that La is the winding inductance of the addition polarity connected to R and L o is the winding inductance of the subtraction polarity connected as shown in Fig. 5C, the mutual inductance M and the coupling coefficient K are expressed by 1), can be theoretically obtained by equation (2).

,, La— Lo ...

M =…)

K ^ ^ £ = ... (2) The inventor conducted experiments using various single transformers T shown in FIGS. 6A to 6E, and conducted mutual inductance 1, coupling coefficient K, and leakage ratio of various transformers T. a '/ ao was measured. Here, 50 kHz was used as the voltage signal (drive signal).

Transformer T shown in Fig. 6A is composed of three core FCs of the two-phase drive three-leg transformer shown in Fig. 2A and connected in the left-right direction as shown in Fig. 2A. A primary winding Cai that supplies the signal a in and a secondary winding Cao that outputs the output voltage signal a are wound, and the primary leg that supplies the input voltage signal b in to the other leg connecting the cores FC. A secondary winding Cbo for outputting the line Cbi and the output voltage signal b is wound.

The transformer shown in Figure 6B is the two-phase drive three-leg transformer shown in Figure 2A. The transformer T shown in FIGS. 6C and 6D is a two-phase drive three-leg transformer shown in FIG.

The transformer T shown in FIG. 6E uses the two-phase drive bipod transformer _c shown in FIG. 2B. _C The transformer T shown in FIGS. 6A to 6E is used. , d2, and the winding inductance L, the mutual inductance M, the coupling coefficient K, and the leakage ratio a '/ ao of the various transformers T were measured.The results shown from the middle to the bottom of Fig. 7 were obtained. .

That is, in the transformer T shown in FIG. 6A, the coupling coefficient = 0.163 and the leakage ratio = 0.012, and in the transformer T shown in FIG. 6B, the coupling coefficient = 0.310 and the leakage ratio = 0.035. In the transformer T shown in Fig. 6C, the coupling coefficient = 0.500, the leakage ratio = 0.063, and in the transformer T shown in Fig. 6D, the coupling coefficient = 0.666, the leakage ratio = 0.109. In the transformer T shown in 6E, a coupling coefficient = 0.737 and a leakage ratio = 0.143 were obtained, respectively.

FIG. 8 shows the relationship between the obtained coupling coefficient and the leakage ratio as a diagram. As is evident from Fig. 8, there is a strong correlation between the coupling coefficient and the leakage ratio. Next, the inventor measured each phase output voltage a, b of each transformer T shown in FIGS. 6A to 6E. The calculated values of the measured output voltages a and b of each phase and the ratio of the difference between the measured output voltages a and b are shown for each transformer T in FIGS. 6A to 6E. Fig. 10 shows the relationship between the output voltage difference ratio and the leakage ratio obtained in Fig. 9 as a diagram.

As is clear from FIGS. 9 and 10, a correlation was obtained between the leakage ratio and the output voltage difference ratio, as the leakage ratio increased, the output voltage difference ratio also increased. That is, as shown in FIG. 10, when the leakage ratio is 0.1, the output voltage difference ratio is 18%. C. Of course, there is no change in the 90 ° phase difference between the output voltage signals a and b.

The present inventor confirmed that the value of the output voltage difference ratio = 18% was within a range that would be acceptable for driving the ultrasonic motor by performing a rotation experiment on the ultrasonic motor. Was.

When the transformer T shown in FIGS. 6A to 6C is used, as shown in FIG. 9, the output voltage difference ratios are 2.3%, 6.7%, and 11.9%, respectively. However, since the value is lower than 18% based on the above-described ultrasonic motor rotation test, there is no problem in driving the ultrasonic motor.

The most important factor in driving the ultrasonic motor is that the phase difference between the transformer output voltage signals supplied to the 3: 1 11-side electrode unit 3 3 and the cos-side electrode unit CC should not differ from 90 °. (Of course, the frequency is the same), and the ultrasonic motor can be driven even if the transformer output voltage is at a slightly different level in each phase. In other words, the above-mentioned simple level is applied to the ultrasonic motor having such a required level (the phase difference of each phase is always 90 ° and the transformer output voltage may be slightly different in each phase). This means that a single transformer configuration can be used effectively. For example, when the frequency, output level, and phase of both signals are constantly changing, such as a stereo audio signal, the above single transformer configuration can be used. Can not be adopted.

As described above, in the present embodiment, since the drive system of the ultrasonic motor is the most expensive and the number of large transformers can be reduced from two to one without being hindered by the driving of the ultrasonic motor, Cost and size can be reduced. Accordingly, the degree of freedom in design can be improved, and the probability of failure can be reduced by reducing the number of parts.

On the other hand, when the transformer T shown in FIGS. 6D and 6E is used, as shown in FIG. 9, the output voltage difference ratios are 19.7% and 25%, respectively. The value is higher than the value of 18% based on the rotation experiment. In particular, when the transformer T shown in Fig. 6E is used, there is a certain difference from 18%, and it is considered that when driving the ultrasonic motor, adverse effects such as a decrease in rotational torque may occur. It is preferable to make a correction that makes the two-phase output voltage signals a and b substantially the same as described below.

Figure 11 shows a drive circuit with a circuit to correct the output voltage difference FIG. 2 is a configuration diagram showing a main part of FIG. The drive circuit 25 is connected between type inputs TP 1 and TP 2 for changing the winding ratio of each phase of the transformer T, and between the primary winding Cai and the power amplifier Amp 1 described above. And a switch SW3 for reversing the rotation direction of the rotor 6 that supplies the voltage signal output from the rotor 6 to the primary winding Cai by inverting the phase by 180 °.

The drive circuit 25 also connects the input tap end of the primary winding Cai to one of the terminals of the primary winding Cai (upper terminal in the drawing) or one of the tap inputs TP1 in conjunction with the switch SW3. The input tap end of the primary winding Cbi is connected to one of the terminals (the upper terminal in the figure) of the tap input TP2 or the secondary winding Cbi in conjunction with the switch SW4 and the switch SW3 for switching to And a switch SW5 for switching to one side.

The tap end positions due to the tap inputs TP1 and TP2 differ depending on the leakage ratio of the transformer T. Based on the relationship between the leakage ratio and the output voltage difference ratio shown in Fig. 10, the sin side electrode section SS and the cos side The drive signals a and b supplied to the electrode section CC are set so that the voltage levels are almost the same.

In FIG. 11, the oscillator 20, the 90 ° phase shifter 21 and the power amplifiers Amp 1 and Amp 2 described in FIG. 1 are omitted.

Next, the operation of the driving circuit 25 configured as described above will be described. When the rotor 6 rotates forward, the switches SW3, SW4, and SW5 are switched to the positions indicated by the solid lines in FIG. 11, and the input tap end of the primary winding Cai is switched to one terminal of the primary winding Cai. The input end of the primary winding Cbi is switched to the input TP2. That is, the winding ratio of the windings Cai and Cao corresponding to the sin side electrode portion S S is reduced, and the winding ratio of the windings Cbi and Cbo corresponding to the cos side electrode portion C C is increased. As a result, the voltage levels of the drive signals a and b become almost the same.

When the rotor 6 rotates in the reverse direction, the switches SW3, SW4, and SW5 are switched to the positions indicated by the dotted lines in FIG. 11, and the input tap end of the primary winding Cai is switched to the evening tap input TP1. PT / JP99 / 00888

In the primary winding Cbi, the input tap end is switched to one terminal of the primary winding Cbi. That is, the winding ratio of the windings Cai and Cao corresponding to the sin-side electrode unit S S is increased, and the winding ratio of the windings Cbi and Cbo corresponding to the cos-side electrode unit C C is reduced. As a result, the voltage levels of the drive signals a and b are almost the same even during the reverse rotation.

As described above, in the drive circuit 25 to which the correction circuit of FIG. 11 is added, the voltage levels of the drive signals a and b are made substantially the same according to the leakage ratio (output voltage difference ratio). However, the transformer T shown in FIGS. 6D and 6E, which can cause adverse effects such as a decrease in rotational torque when driving the ultrasonic motor, can be used with confidence.

FIG. 12 is a configuration diagram showing a main part of a drive circuit provided with a circuit for correcting the output voltage difference in a direction to reduce the difference, and FIGS. 13A to 13P are timing diagrams for explaining the circuit operation of FIG. It is a chart.

The drive circuit 30 has FETs 40 to 43, 50 to 50 instead of a configuration in which the mechanical switch SW3 switches the rotation direction of the rotatable 6 described in FIG. A similar operation is obtained by switching using 53.

That is, inputs al to a4 are pulse wave signal inputs on the primary winding Cai side, inputs b1 to b4 are pulse wave signal inputs on the primary winding Cbi side, and inputs al to a4 and bl to b4 are shown in the figure. Is generated by a digital circuit omitting the symbol and is output at a predetermined timing. Inputs a 1 and a 4 are push-pull inputs to FETs 40 and 43 that have a phase difference of 180 ° from each other, and correspond to the drive when the winding ratio of windings Cai and Cao is reduced. Reference numeral 3 denotes a push-pull input having a phase difference of 180 ° with respect to FETs 41 and 42, which corresponds to a driving operation in which the winding ratio of the windings Cai and Cao is increased.

The input system on the b side has a 90 ° phase difference with respect to the input system on the a side, and the input bl: b 4 is a push-pull input to FETs 50 and 53 having a phase difference of 180 ° from each other, and corresponds to a drive in which the winding ratio of windings Cbi and Cbo is reduced, while inputs b 2 and b 3 are mutually connected. Push-pull input to FETs 51 and 52 with a phase difference of 180 °, which is compatible with driving in which the winding ratio of windings Cbi and Cbo is increased.

When the mouth 6 is rotating forward, the input al and & 4 are activated, as shown in Fig. 13A and Fig. 13D. Therefore, the circuit that reduces the turns ratio operates, while the inputs a2 and a3 are As shown in 3B and Figure 13C, the FETs 41 and 42 are turned off at input 0, so the circuit that increases the turns ratio does not operate. Since the inputs al and a4 have a phase difference of 180 ° as shown in Figs. 13A and 13D, the push-pull addition is performed in the transformer T and the drive signal ( Output voltage signal) a is obtained.

Here, the inputs al and a 4 are 25% pulse width inputs, but due to the resonance action of the capacitance of the piezoelectric vibrator 5 v and the inductance L of the transformer T, a sinusoidal wave is applied to the vibrator 5 V. A practical drive signal a (similar to a sine wave) is supplied.

In addition, when the mouth 6 rotates in the normal direction, the inputs b 2 and b 3 operate as shown in FIGS. 13F and 13G, so that the circuit for increasing the winding ratio operates. In b4, as shown in Fig. 13E and Fig. 13H, the FETs 50 and 53 are turned off at input 0, so the circuit for reducing the winding ratio does not operate. As shown in Figures 13F and 13G, the inputs b2 and b3 have a 180 ° phase difference, so they are push-pulled in the transformer T and supplied to the cos-side electrode CC side. b is obtained. This drive signal b is a practical drive signal like a sine wave like the drive signal a.

Also, this drive signal b has a phase difference of 90 ° with respect to the drive signal a because the inputs b 2 and 3 have a phase difference advanced by 90 ° with respect to the inputs & 1 and a4. Have. In addition, as described above, the winding ratio of the windings Cai and Cao corresponding to the SS side of the sin side electrode unit is reduced, and the winding ratio of Cbi and Cbo corresponding to the CC side of the cos side electrode unit CC is increased. Therefore, the voltage levels of the drive signals a and b are almost the same.

That is, the drive signals a and b have a phase difference of 90 ° and the voltage levels are almost the same.

When the one-way mouth 6 reverses, the inputs a 2 and a 3 are input as shown in Figs. 13J and 13K in order to reverse the rotor 6 as shown in Figs. 13J and 13K. A, see Fig. 13D). On the other hand, for the inputs a 1 and a 4, as shown in FIGS. 13I and 13L, the FETs 40 and 43 are turned off at the input 0, so that the circuit for reducing the winding ratio does not operate. Inputs a 2 and a 3 have a 180 ° phase difference as shown in Figure 13J and Figure 13K. The signal a is obtained.

When the rotor 6 rotates in the reverse direction, the inputs b 1 and b 4 operate as shown in FIGS. 13M and 13P, so that the circuit for reducing the winding ratio operates. On the other hand, the inputs b 2 and b 3 As shown in Fig. 13N and Fig. 130, the FETs 51 and 52 are turned off at input 0, and the circuit for increasing the winding ratio does not operate. Since the inputs bl and b4 have a phase difference of 180 ° as shown in Figs. Can be The drive signal b has a phase difference of 90 ° with respect to the drive signal a because the inputs b 1 and b 4 have a phase difference of 90 ° with respect to the inputs a 2 and a 3. In addition, the winding ratio of windings Cai and Cao corresponding to the sin side electrode section SS side is increased, and the winding ratio of Cbi and Cbo corresponding to the cos side electrode section CC side is reduced. The voltage levels of the drive signals a and b are almost the same.

Thus, even in the drive circuit 30 in FIG. 12, the drive signals a and b are supplied without changing the 90 ° phase difference between the drive signals a and b according to the leakage ratio (output voltage difference ratio). Since the pressure levels are set to be almost the same, adverse effects such as a decrease in rotational torque may occur when driving the ultrasonic motor, and the transformer T shown in Figs. You can use it with your heart.

In addition, as compared to the drive circuit 25 shown in FIG. 11, a mechanical switch is not used, so that it is more practical in configuring a circuit.

As described above, the invention made by the present inventor has been specifically described based on the embodiment. However, the present invention is not limited to the above embodiment, and it can be variously modified without departing from the gist thereof. Needless to say, for example, in the above-described embodiment, the phase difference between the two-phase drive signals is 90 °, but is not limited to 90 °.

Further, according to the present invention, a piezoelectric vibrator is also fixed to the back surface of the mouth, as described in, for example, Japanese Patent Application Laid-Open No. By applying a drive signal, a traveling wave traveling in the circumferential direction is generated on the rotor surface, and the rotor is rotationally driven by the traveling wave generated on the rotor surface and the traveling wave generated on the stay surface described above. The present invention is also applicable to a type of ultrasonic camera.

Furthermore, it goes without saying that the present invention is similarly applicable to a linear type ultrasonic motor. Industrial applicability

The driving method of the ultrasonic motor according to the present invention comprises: a single transformer in which primary and secondary windings of each phase are wound on a common core forming a closed magnetic circuit; Thus, drive signals for each phase are input. Therefore, in the single transformer, the drive signals of each phase interfere with each other, and the effect of the mutual interference is that the transformer output voltage of the secondary winding supplied to the oscillator is slightly different in each phase. The relationship having a predetermined phase difference is maintained only by the level, so that the ultrasonic motor can be driven without any trouble in actual use. As a result, ultrasound In the motor drive system, the most expensive and large number of transformers can be reduced, and it is possible to drive the ultrasonic motor satisfactorily while reducing the cost and size.

Claims

Scope of word blue

1. Two-phase drive signals having different phases are supplied to the vibrator provided on the back surface of the elastic body to vibrate the vibrator, and the mover is driven by a traveling wave generated on the surface of the elastic body. A method of driving an ultrasonic motor,

The two-phase drive signals having different phases are input to the primary winding of each phase of a single transformer in which the primary and secondary windings of each phase are wound on a common core forming a closed magnetic circuit. A method for driving an ultrasonic motor, comprising: boosting the pressure and supplying the boosted voltage from the secondary windings of the respective phases to the vibrator.

2. The step-up by the winding in the single transformer according to the ratio of the voltage at which one of the two-phase drive signals of the single transformer interferes with the other drive signal. 2. The ultrasonic motor driving method according to claim 1, wherein the ratio is made different for each phase, and output voltages of two-phase drive signals supplied to the vibrator are made substantially the same.

3. In accordance with the inversion of the input phase of the drive signal to the primary winding of the single transformer, the boost ratios that are different for each of the phases are exchanged with each other, and the two-phase drive signal supplied to the vibrator is changed. 3. The method for driving an ultrasonic motor according to claim 2, wherein the output voltages are substantially the same.

4. The drive signal for the primary winding of the single transformer is a pulse wave signal, and a sinusoidal wave is applied to the vibrator by the resonance action of the capacitance of the vibrator and the inductance of the single transformer. The method for driving an ultrasonic motor according to claim 1, wherein a driving signal having a wave shape is supplied.