WO1997018624A1 - Fm modulator - Google Patents
Fm modulator Download PDFInfo
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- WO1997018624A1 WO1997018624A1 PCT/JP1996/001705 JP9601705W WO9718624A1 WO 1997018624 A1 WO1997018624 A1 WO 1997018624A1 JP 9601705 W JP9601705 W JP 9601705W WO 9718624 A1 WO9718624 A1 WO 9718624A1
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- circuit
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- resistor
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Classifications
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03C—MODULATION
- H03C3/00—Angle modulation
- H03C3/02—Details
- H03C3/08—Modifications of modulator to linearise modulation, e.g. by feedback, and clearly applicable to more than one type of modulator
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03C—MODULATION
- H03C3/00—Angle modulation
- H03C3/10—Angle modulation by means of variable impedance
- H03C3/12—Angle modulation by means of variable impedance by means of a variable reactive element
- H03C3/20—Angle modulation by means of variable impedance by means of a variable reactive element the element being a voltage-dependent capacitor
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03C—MODULATION
- H03C3/00—Angle modulation
- H03C3/10—Angle modulation by means of variable impedance
- H03C3/28—Angle modulation by means of variable impedance using variable impedance driven mechanically or acoustically
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03B—GENERATION OF OSCILLATIONS, DIRECTLY OR BY FREQUENCY-CHANGING, BY CIRCUITS EMPLOYING ACTIVE ELEMENTS WHICH OPERATE IN A NON-SWITCHING MANNER; GENERATION OF NOISE BY SUCH CIRCUITS
- H03B5/00—Generation of oscillations using amplifier with regenerative feedback from output to input
- H03B5/20—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising resistance and either capacitance or inductance, e.g. phase-shift oscillator
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03B—GENERATION OF OSCILLATIONS, DIRECTLY OR BY FREQUENCY-CHANGING, BY CIRCUITS EMPLOYING ACTIVE ELEMENTS WHICH OPERATE IN A NON-SWITCHING MANNER; GENERATION OF NOISE BY SUCH CIRCUITS
- H03B5/00—Generation of oscillations using amplifier with regenerative feedback from output to input
- H03B5/20—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising resistance and either capacitance or inductance, e.g. phase-shift oscillator
- H03B5/24—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising resistance and either capacitance or inductance, e.g. phase-shift oscillator active element in amplifier being semiconductor device
Definitions
- the present invention relates to an FM modulator for FM-modulating a sound and sending it out.
- the FM wireless microphone amplifies the sound collected by the microphone, performs FM modulation, amplifies the FM-modulated signal, and transmits it from the antenna.
- the microphone, low-frequency bandwidth circuit, and FM modulation It includes a circuit, a high-frequency amplifier circuit, and an antenna.
- the FM modulation circuit is generally configured using an LC oscillator that generates a sine wave with little distortion.
- LC oscillators such as Colpitts type LC oscillators.
- the capacity of the LC resonance circuit included in the LC oscillator is composed of a varicap (variable capacitance diode), and the capacitance is the voltage of the FM modulation signal.
- FM modulation is performed by changing the level according to the fluctuation of the level.
- this type of LC oscillator has the problem that if the oscillation frequency is greatly changed, the voltage level of the oscillation output also changes, and it is not practical as it is. Therefore, when using this type of LC generator, a circuit for keeping the amplitude of the FM carrier constant is required, and the circuit configuration becomes complicated. Disclosure of the invention
- the present invention has been conceived to solve such a problem, and an object thereof is to provide an FM modulator having a simple circuit configuration.
- the FM modulator of the present invention includes two all-pass type phase shift circuits including a differential amplifier and a CR circuit, and cascade-connects these two phase shift circuits to output the output of the phase shift circuit at a subsequent stage.
- a condenser microphone as a capacitor in the CR circuit included in one of the two phase shift circuits.
- FIG. 1 is a circuit diagram showing a configuration of an FM modulator of a first embodiment
- FIG. 2 is a diagram showing another configuration of the FM modulator
- FIG. 3 is a circuit diagram showing a configuration of an oscillator in which a capacitor microphone or the like included in the FM modulator shown in FIG. 1 is replaced with a capacitor having a fixed capacitance;
- Fig. 4 is a vector diagram related to the input / output voltage etc. of the preceding phase shift circuit shown in Fig. 3
- Fig. 5 is a vector related to the input / output voltage etc. of the subsequent phase shifting circuit shown in Fig. 3.
- Fig. 6 is a circuit diagram showing the configuration of the phase shift circuit including the condenser microphone.
- FIG. 7 is a circuit diagram showing a configuration of a phase shift circuit including an LR circuit
- FIG. 8 is a vector diagram relating to the input / output voltage of the phase shift circuit shown in FIG. 7, and FIG. 9 is a circuit diagram showing another configuration of the phase shift circuit including the LR circuit.
- FIG. 10 is a vector diagram relating to the input / output voltage of the phase shift circuit shown in FIG. 9, and FIG. 11 is a circuit diagram showing a configuration of an oscillator having a voltage divider circuit in the phase shift circuit.
- FIG. 12 is a circuit diagram showing another configuration of the oscillator,
- FIG. 13 is a circuit diagram showing a configuration of a phase shift circuit which can be replaced with the preceding phase shift circuit shown in FIG. 12,
- FIG. 14 is a circuit diagram showing the configuration of a phase shift circuit that can be replaced with the subsequent phase shift circuit shown in FIG.
- FIG. 15 is a circuit diagram showing a detailed configuration of an FM modulator according to a fourth embodiment.
- FIG. 16 is a circuit diagram showing a configuration of an oscillator including a phase inversion circuit.
- FIG. 17 is a circuit diagram showing another configuration of an oscillator including a phase inversion circuit
- FIG. 18 is a circuit diagram for reducing the detailed configuration of the FM modulator according to the seventh embodiment
- FIG. 19 is a diagram illustrating a capacitor microphone and the like included in the FM modulator shown in FIG. Is a circuit diagram showing a configuration of an oscillator replaced with a fixed capacity
- FIG. 20 is a vector diagram relating to input / output voltages and the like of the preceding phase shift circuit shown in FIG.
- Figure 21 is a vector diagram related to input and output pressure of the subsequent phase shift circuit shown in Fig. 19,
- FIG. 22 is a circuit diagram showing the configuration of a phase shift circuit including a condenser microphone
- FIG. 23 is a diagram showing the configuration of a phase shift circuit that can replace the preceding phase shift circuit shown in FIG. ⁇ Road map
- Fig. 24 is a vector diagram of the input / output voltage etc. of the phase shift circuit shown in Fig. 23, and Fig. 25 is a phase shifter that can be replaced with the subsequent phase shift circuit shown in Fig. 19 A circuit diagram showing the structure of the circuit,
- FIG. 26 is a vector diagram of the phase shift shown in FIG. 25 [port: input / output turret pressure of the road, etc.]
- FIG. 27 is a circuit diagram showing a configuration of an oscillator including a phase inversion circuit
- FIG. 28 is a circuit diagram showing another configuration of the oscillator including the phase inversion circuit
- FIG. 29 is a circuit diagram showing a configuration of the FM modulator according to the tenth embodiment.
- FIG. 30 is a diagram showing the FET and its peripheral circuits included in the FM modulator shown in FIG. Is a circuit diagram showing a configuration of an oscillator replaced with a fixed resistor,
- FIG. 31 is a vector diagram relating to the input / output IS pressure and the like of the preceding phase shift circuit shown in FIG. 30;
- FIG. 32 is a vector diagram relating to the human output voltage and the like of the subsequent phase shift circuit shown in FIG. 30,
- FIG. 33 is a circuit diagram showing a configuration of a phase shift circuit including a condenser microphone.
- FIG. 34 is a circuit diagram showing a configuration of a phase shift circuit including an LR circuit.
- FIG. 35 is a circuit diagram showing another configuration of the phase shift circuit including the LR circuit
- FIG. 36 is a circuit diagram showing a configuration of an oscillator including a phase inversion circuit
- FIG. 37 is a circuit diagram showing another configuration of the oscillator including the phase inversion circuit
- FIG. 38 is a circuit diagram in which a portion necessary for the operation of the phase shift circuit in the configuration of the operational amplifier is extracted.
- FIG. 1 is a circuit diagram showing a configuration of an FM modulator according to a first embodiment to which the present invention is applied.
- the FM modulator 1 shown in the figure includes two phase shift circuits 110 C and 30 C that perform a total of 360 ° phase shift at a predetermined frequency, and a phase shift circuit 30 C at the subsequent stage. And a feedback resistor 70 that feeds back the output of the first stage to the input side of the phase shift circuit 110 C of the preceding stage.
- the feedback resistor 70 has a finite resistance value from 0 ⁇ .
- the FM modulator 1 shown in FIG. 1 is provided with a condenser microphone as described later, and FM-modulates the voice collected by the condenser microphone and outputs it.
- an amplifier 2 and an antenna 3 are connected after the FM modulator 1, and the output of the FM modulator 1 is amplified by the amplifier 2 and transmitted from the antenna 3 to the air. It would be an FM wireless microphone.
- the signal may be transmitted to the transmission line 400 via the transmission driver 4 as shown in FIG.
- FIG. 3 shows a simplified investigation by replacing the condenser microphone 141-1 and capacitor 14-2 included in the FM modulator 1 shown in 3 ⁇ 41 ⁇ with a fixed capacitance 14 capacitor.
- 6 is a circuit diagram showing a configuration of a device 5.
- FIG. The oscillator 5 shown in the figure includes two phase shift circuits 10 C and 30 C that perform a total of 360 ° phase shift at a predetermined frequency, and outputs the output of the subsequent phase shift circuit 30 C in the preceding stage. And a feedback resistor 70 for feeding back to the input side of the phase shift circuit 10C.
- the phase shift circuit 10 C at the preceding stage constituting the oscillator 5 shown in FIG. 3 is composed of an operational amplifier (operational amplifier) 12, which is a type of differential amplifier, and the phase of a signal input to the phase shift circuit 10 C. Is shifted by a predetermined amount, and the resistor 16 and the capacitor 14 input to the non-inverting input terminal of the operational amplifier 12 and the input terminal of the phase shift circuit 10 C and the inverting input terminal of the operational amplifier 12 It is configured to include a resistor 18 inserted and a resistor 20 inserted between the output terminal of the operational amplifier 12 and the inverting input terminal.
- the voltage VC1 appearing at both ends of the capacity 14 is applied to the non-inverting human terminal of the step 12. Also, since there is no potential difference between the two input terminals of the operational amplifier 12, the potential of the inverting input terminal of the operational amplifier 12 is equal to the potential of the connection point between the resistor 16 and the capacitor 14. Therefore, the same voltage VR1 as the ⁇ 3 ⁇ 4 terminal voltage VR1 of the resistor 16 appears at both ends of the resistor 18.
- the relationship between the magnitude and the phase of the input / output voltage can be expressed by an isosceles triangle with the input voltage E i and the output voltage E o as the hypotenuse and the base as twice the voltage VR1. It can be seen that the amplitude is the same as the amplitude of the input signal regardless of the frequency, and the phase shift amount is represented by 01 shown in FIG.
- phase shift [nj path 30 C] of the subsequent stage constituting the generator 5 shown in FIG. 3 is input to an operational amplifier 32 which is a type of differential amplifier and to this phase shift I] path 30 C.
- a resistor 36 to shift the phase of the input signal by a predetermined amount to the non-inverting input terminal of the operational amplifier 32, and the input terminal of the phase shift circuit 30C and the inverting power of the operational amplifier 32.
- a resistor 38 is inserted between the output terminal of the operational amplifier 32 and the inverted terminal, and a resistor 40 is inserted between the output terminal of the operational amplifier 32 and the inverted human terminal.
- phase shift circuit 30 C The basic configuration of the phase shift circuit 30 C is the same as that of the preceding phase shift circuit 10 C, and the capacity 3 4 and the resistor 3 6 which constitute the CR circuit in the phase shift circuit 30 C are connected.
- the order of connection is opposite to the order of connection between the capacity 14 and the resistor 16 constituting the CR circuit in the phase shift circuit 10C.
- the input voltage E i And the output voltage E o is defined as the hypotenuse, and the base of the power K VC2 can be represented by an isosceles triangle. It can be seen that the phase shift amount is represented by 02 shown in FIG.
- phase of each of the two phase shift circuits 10 C and 30 C is shifted by a predetermined amount, and the amount of phase shift is determined by the entire two phase shift circuits 10 C and 30 C at a predetermined frequency.
- a signal totaling 360 ° is output.
- the output of the subsequent phase shift circuit 30 C is fed back to the input side of the phase shift circuit 10 C via the feedback resistor 70, and the loop has been completed by setting the loop gain of the feedback loop to 1 or more.
- a sine wave oscillation occurs at a frequency such that the sum of the phase shift views is 360 °.
- the gains of the phase shift circuits 10 C and 30 C become 1, and furthermore, in the feedback loop, The loop gain will be less than 1 because some peaks will occur. For this reason, in order to make the loop gain 1 or more, it is necessary to make the resistance value of the resistance 20 larger than the resistance 18 or make the resistance value of the resistance 40 larger than the resistance 38.
- FIG. 6 is a circuit diagram showing the configuration of a phase shift circuit including a condenser microphone
- FIG. 6 The configuration of the preceding phase shift circuit 110C included in the device 1 is shown.
- This phase-shift circuit 110 C is the same as the phase-shift circuit 10 C in the preceding stage included in the generator 5 shown in FIG. 3 except that a CR circuit comprising a resistor 16 and a capacitor 14 is connected to a resistor 16 and a capacitor microphone
- the right side shows the configuration in which a CR circuit consisting of 1 and 14 is replaced.
- This change in capacitance reflects the change in sound pressure picked up by the condenser microphones 14-11, and the time constant of the CR circuit included in one phase shift circuit 110C changes according to the sound pressure.
- the frequency of the oscillation output also changes.
- FM-modulated signals can be easily obtained by using the condenser microphone 14-1 in the-part of the CR circuit. Therefore, the circuit configuration itself of the FM modulator 1 can be simplified.
- each of the two phase shift circuits 110 C and 30 C constituting the FM modulator 1 is an all-pass circuit, and even if the frequency of the FM-modulated carrier is changed, the amplitude is changed. Is almost constant, and another configuration for preventing amplitude fluctuation is not required.
- a general condenser microphone is a microphone that uses the change in capacitance of a parallel plate capacitor, applies a predetermined DC voltage between parallel electrodes through a resistance of several tens of ⁇ , and applies electrostatic capacitance due to sound pressure.
- the change in voltage according to the change in capacitance is extracted as an electrical signal. Therefore, it is necessary to apply a predetermined DC voltage for normal use, but in the present embodiment, the condenser microphone 1411 is used as a capacitor whose capacitance changes according to the sound pressure. Therefore, it is not necessary to apply a predetermined DC voltage.
- a condenser microphone is included in the preceding phase shift circuit, but a condenser microphone may be included in the subsequent phase shift circuit. That is, the phase shift circuit 30 C of the subsequent stage constituting the oscillator 5 shown in FIG. 3 is replaced with the phase shift circuit 130 C shown in FIG. 6 (B) (capacity signal in the phase shift circuit 30 C). (The one using condenser microphone 34-1 and capacity 34-2 instead of 4) may be used.
- the two phase shift circuits are both configured to include the CR circuit, but one of the phase shift circuits not including the condenser microphone is replaced with a phase shift circuit including the LR tol path. You can also.
- FIG. 7 is a circuit diagram showing a configuration of a phase shift circuit 10L which can be replaced with the preceding phase shift circuit 10C included in the oscillator 5 shown in FIG.
- the phase shift circuit 10 L shown in FIG. 7 is different from the phase shift circuit 10 C shown in FIG. 3 in that a CR circuit consisting of the resistor 16 and the capacitor 14 is connected to the inductor 17 and the resistor 16. LR circuit. Considering the relationship between the magnitude and phase of the input / output voltage assuming that the voltage across the inductor 17 is VL1 and the voltage across the resistor 16 is VR1, as shown in Fig.
- the input voltage E i and the output voltage E o Can be represented by an isosceles triangle whose base is twice the voltage VL1.
- the amplitude of the output signal is the same as the amplitude of the human-powered signal regardless of the frequency, and the amount of phase shift is shown in Fig. 8. It can be seen that it is represented by 03 as shown. If the time constant of the CR circuit in the phase shift circuit 10C shown in FIG. 3 and the time constant of the LR circuit in the phase shift circuit 10L shown in FIG.
- the transfer functions of the phase circuits 10C and 10L are both (1-Ts) / (1 + Ts).
- s j ⁇ .
- phase shift circuit 10L is equivalent to the phase shift circuit 10C, and the phase shift circuit 10C can be replaced with the phase shift circuit 10L. Therefore, in the oscillator 5 shown in FIG. 3, the phase shift circuit 10 C in the preceding stage is replaced with the phase shift circuit 10 L shown in FIG. 7, and the phase shift circuit 30 C in the latter stage is replaced by a condenser microphone.
- phase shift circuit 10 C in the preceding stage is replaced with the phase shift circuit 10 L shown in FIG. 7, and the phase shift circuit 30 C in the latter stage is replaced by a condenser microphone.
- each of the two phase shift circuits constitutes an FM modulator including an LR circuit or a CR circuit. Can be.
- FIG. 9 is a circuit diagram showing a configuration of a phase shift circuit 30L that can be replaced with a subsequent phase shift circuit 30C included in the oscillator 5 shown in FIG.
- the phase shift circuit 30L shown in Fig. 9 is different from the phase shift circuit 30C shown in Fig. 3 in that the CRIHJ path consisting of the capacitor 34 and the resistor 36 is replaced by an LR circuit consisting of the resistor 36 and the inductor 37. It has a different configuration.
- the CRIHJ path consisting of the capacitor 34 and the resistor 36 is replaced by an LR circuit consisting of the resistor 36 and the inductor 37. It has a different configuration.
- the input voltage Ei and the output voltage Eo are The amplitude of the output signal is the same as the amplitude of the input ⁇ signal regardless of the frequency, and the position ffl shift is shown in Fig. 10. You can see that it is represented by 04. Assuming that the time constant of the CR circuit in the phase shift circuit 30C shown in FIG. 3 and the time constant of the LR circuit in the phase shift circuit 30L shown in FIG. The transfer functions of the circuits 30 C and 30 L are both 1 (1-T s) / (1 + T s).
- phase shift circuit 30L is equivalent to the phase shift circuit 30C, and the phase shift circuit 30C can be replaced with the phase shift circuit 30L. Therefore, in the generator 5 shown in FIG. 3, the subsequent phase shift circuit 30C is replaced with the phase shift circuit 30L shown in FIG. 9, and the preceding phase shift circuit 10C includes a condenser microphone.
- phase shift circuit 110C shown in FIG. 6 (A) each of the two phase shift circuits can constitute an FM modulator including an LR line or a CR line.
- each phase shift circuit shown in Fig. 3 the output of the operational amplifier is directly fed back to the human side of the operational amplifier via a resistor.However, a voltage divider is connected to the output terminal of each operational amplifier to divide the voltage. The output may be fed back to the input side of the operational amplifier.
- FIG. 11 is a circuit diagram showing a detailed configuration of an oscillator provided with a voltage dividing circuit in a phase shift circuit.
- the voltage dividing circuit composed of the resistors 21 and 23 is connected to the output terminal of the operational amplifier 12 in the phase shift M path 210C shown in FIG. It is connected to the anti-fc human input terminal of the operational amplifier 12 via the resistor 20.
- the output terminal of the operational amplifier 32 in the phase shift circuit 230 C is connected to a voltage dividing circuit composed of the resistors 41 and 43, and the voltage dividing output terminal is connected to the operational amplifier via the resistor 40. 32 Connected to 2 inverting input terminal.
- phase shift circuit 230 C Even if the frequency changes, it is possible to shift only the phase by a predetermined fidelity while keeping the width of the output pressure E o constant.
- the resistance values of the resistors 18 and 20 are the same and the resistance values of the resistors 38 and 40 are the same. Even if there is, the loop gain of the feedback loop formed by cascading the two phase shift circuits can be reliably set to 1 or more, and the oscillation operation can be stabilized.
- FIG. The FM modulator can be configured in the same manner as.
- FIG. 11 shows an example in which a voltage dividing circuit is connected to the output terminals of the operational amplifiers 12 and 32 in the phase shift circuits 10 C and 30 C shown in FIG.
- a voltage divider is connected to the output terminals of the operational amplifiers in the phase shift circuits 10 L and 30 L shown in the figures and Fig.
- FIG. 12 is a circuit diagram showing another configuration of the oscillator.
- the oscillator 5B shown in the figure is configured to include two phase shift circuits 410C and 430C that perform a total of 360 ° phase shift at a predetermined frequency.
- the oscillator 5A shown in Fig. 11 changes the frequency of the input AC signal by setting the resistances ⁇ of the resistors 18 and 20 in the preceding phase shift circuit 10C to the same value. The amplitude change at the time of is suppressed.
- the phase shift circuit 4100 C in the preceding stage included in the oscillator 5 B shown in FIG. 12 does not have a shunt in the phase shift circuit and has a higher resistance than the resistance of the resistor 18 ′.
- the use of the phase shift circuit 410C is set to a value more unique than 1.
- the gain of the phase shift circuit 430C is set to 1 It is set to a larger value.
- gain fluctuation may occur depending on the frequency of the input signal.
- the phase shift circuit 410C when the frequency of the input signal is low, the phase shift circuit 410C becomes a voltage hollow path, and the gain at this time becomes 1 times.
- the phase shift circuit 4 10 when the frequency is high, the phase shift circuit 4 10 (is an anti-amplifier, so the gain at this time is -m times (m is the resistance ratio between the resistance 20 'and the resistance 18').
- the gain of the phase shift circuit 410C also changes, and the amplitude of the output signal fluctuates.
- Such amplitude fluctuation can be suppressed by connecting the resistor 19 to the inverting input terminal of the operational amplifier 12 and matching the gain when the frequency of the input signal is low with that when the frequency of the input signal is high.
- the input signal is set by setting the resistance of the resistor 19 to mr / (m-1).
- the phase shift circuit 4300C is also connected to the inverting input terminal of the operational amplifier 32.
- the capacity in the CR iP] circuit which is provided by either of the two phase shift circuits 41 C or 43 C shown in FIG.
- An FM modulator can be configured in the same manner as in the figure.
- the oscillator 5B shown in FIG. 12 has a cascade connection of phase shift circuits 4110C and 4330C including a CR circuit
- the CR circuit can be replaced with an LR circuit.
- the phase shift circuit 4 10 L shown in FIG. 13 is the same as the previous phase shift circuit 4 10 C shown in FIG. 12, and the phase shift circuit 4 10 C is a phase shift circuit. It can be replaced with 410 L.
- the phase shift circuit 430 L shown in FIG. 14 is equivalent to the phase shift circuit 430 C of the subsequent stage shown in FIG. It can be replaced by 30 L.
- the capacity 34 in 0 C may be configured using a condenser microphone.
- the combined phase shift of the two phase shift circuits is set to 360 ° at a predetermined frequency, but the two phase shift circuits are cascaded.
- the FM modulator may be configured by connecting a non-inverting circuit that does not change the phase to a part of the feedback loop formed as described above.
- FIG. 15 is a circuit diagram showing a detailed configuration of a fourth embodiment of the FM modulator.
- the FM modulator 1A shown in the figure is the same as the FM modulator 1 shown in FIG. 1 in that the phase shift circuit 110C and the phase shift circuit 30C are connected in cascade, and the subsequent phase shifter is used.
- FIG. 1 shows that the non-inverting circuit 50 is connected to the output side of the circuit 30 C.
- the non-inverting circuit 50 includes an operational amplifier 52 and resistors 54 and 56, and has a predetermined gain according to a resistance ratio of the two resistors 54 and 56. Therefore, the loss at the time of forming the closed loop can be compensated by this gain, and the loop gain of the feedback loop can be easily set to 1 or more.
- the non-inverting path 5 It is also possible for 0 to have a function as a power amplification stage.
- FIG. 15 as an example, the configuration in which the non-inverting circuit 50 is connected to the FM modulator 1 shown in FIG. 1 has been described, but the above-described various phase shift circuits are cascaded in an arbitrary order.
- the non-inverting circuit 50 shown in FIG. 15 may be connected to various FM modulators configured as described above.
- the phase shift amount is controlled by two phase shift circuits, and the operation of adjusting the frequency at which the juice becomes 360 ° is performed.
- a phase inversion circuit is provided in a closed loop. By closing the aperture, the total amount of phase shift by the two phase shift circuits is 180.
- the vibration operation may be performed at a frequency such that:
- FIG. 16 shows an In] path of an oscillator configured by cascading two phase shift circuits and a phase inversion Lnj path.
- the oscillator 5C shown in the figure is composed of a cascade connection of two stages of the preceding phase shift circuit 10C in the generator 5 shown in FIG. 3, and an operational amplifier 82 and resistors 84, 8 A phase inversion circuit 80 composed of 6 is connected, and the output of the phase inversion circuit 80 is fed back to the human-powered side of the preceding phase shift circuit 10 C via a resistance 70.
- the oscillator 5C shown in FIG. 16 shows an example in which the phase shift circuit 10C is cascaded
- the oscillator 5C shown in FIG. 3 may be cascaded with the subsequent phase shift circuit 30C.
- FIG. 17 is a circuit diagram showing another configuration of the oscillator including the phase inversion IHJ path therein.
- a phase shift circuit 30C at the subsequent stage in the oscillator 5 shown in FIG. 3 is vertically connected by two stages, and a phase inverting circuit 80 is connected to the subsequent stage.
- the output of the inversion circuit 80 is fed back to the human side of the preceding phase shift circuit 30 C via the feedback resistor 70.
- the phase inverting circuit 80 Since the signal is inverted by the phase inverting circuit 80, when the total phase shift amount of the two phase shift circuits 30 C is 180 °, the phase shift amount when making a round of the closed loop is 3 60 °, and a predetermined oscillation operation is performed by setting the loop gain of the feedback loop at this time to 1 or more.
- the condenser microphone By replacing one of the two phase-shift circuits 30 C included in the oscillator 5 D with the phase-shift circuit 130 C shown in FIG. 6B, sound is collected by the condenser microphone. It is possible to configure an FM modulator using the converted voice for the FM modulation signal. Alternatively, one of the two phase shift circuits 30 C included in the oscillator 5 B is replaced with a phase shift circuit 130 C shown in FIG. 6B and the other is shown in FIG. The FM modulator may be configured by replacing the phase shift circuit with 30 L.
- FIG. 18 is a circuit diagram showing a detailed configuration of the seventh embodiment of the FM modulator.
- the FM modulator 1B shown in the figure includes two phase shift circuits 710C and 630C that perform a total of 360 ° phase shift at a predetermined frequency, and a phase shift circuit 6 in the subsequent stage.
- a non-inverting circuit 650 that amplifies and outputs the output of the 3 0 C at a predetermined amplification level without changing the phase, and the output of the non-inverting circuit 65 0 is the input side of the previous phase shift circuit 7 10 C
- a resistor 670 that feeds back the current.
- This resistor 670 has a finite resistance value from 0 ⁇ .
- the capacitor 672 connected in series with the resistor 670 is for blocking DC current, and its impedance is extremely small at the operating frequency, that is, it has a large capacitance. I have.
- the FM modulator 1B is configured to include a condenser microphone (details will be described later), and uses the sound obtained by the condenser microphone as an FM modulation signal, and outputs an FM-modulated signal as an oscillation output. ing.
- an amplifier 2 and an antenna 3 are connected after the FM modulator 1B, and the output of the FM modulator 6 is amplified by the amplifier 2 and the antenna If you send it out of the air from 3, it becomes an FM wireless microphone.
- the signal may be transmitted to the transmission line 400 via the transmitting driver 4 as shown in FIG.
- FIG. 19 shows the case where the condenser microphone 614-1 and the capacitor 614-2 included in the FM modulator 1B shown in Fig. 18 are replaced with a fixed capacitance 614.
- FIG. 3 is a circuit diagram showing a configuration of the vibrator.
- the oscillator 5E shown in FIG. 3 has two phase shift circuits 6 10C and 630C that perform a phase shift of 360 ° by a total at a predetermined frequency, and the phase of the output signal of the subsequent phase shift circuit 630C. And a resistor 670 that feeds back the output of the non-inverting circuit 650 to the input side of the previous phase shift circuit 6100C. ing.
- phase shift circuit 610C is composed of a FET 612 connected to the input terminal of the phase shift circuit 610C, A capacitor 6 14 and a resistor 6 16 connected in series between the source 2 and the drain; a resistor 6 18 connected between the drain of the FET 6 12 and the positive power supply; and a FET 6 1 and a resistor 620 connected between the source and ground.
- the resistor 626 in the phase shift circuit 610 C is for applying an appropriate bias voltage to the FET 612.
- at least the FET 612 and the FET 632 described later may be replaced with a bipolar transistor.
- the resistance values of the two resistors 620 and 618 connected to the source and the drain of the FET 612 described above are set to be substantially equal, and the resistance of the input voltage applied to the gate is reduced to the AC component.
- a signal with the same phase is output from the source of the FET 612, and a signal whose phase is inverted and whose amplitude is equal to the signal output from the source is output from the drain of the FET 612.
- the amplitude of the AC voltage applied to the source and drain is Ei.
- a series circuit composed of a capacitor 614 and a resistor 616 is connected between the source and drain of the FET 612, and the voltage appearing at the source and drain of the FET 612 It's resistance 61 6 or capacity evening 6 1
- the signal synthesized via 4 is output from the phase shift circuit 6100C.
- the voltage VC1 appearing at both ends of the capacitor 614 and the voltage VR1 appearing at both ends of the resistor 616 are 90 ° out of phase with each other. Since it is equal to the voltage 2 Ei between the source and the drain of 2, as shown in Fig. 20, twice the voltage Ei is the hypotenuse, and the voltage VC1 across the capacitor 614 and the voltage VR1 across the resistor 616 are : A right triangle that forms the two intersecting sides will be formed. Assuming that the potential difference between the connection point of the capacitor 6 14 and the resistor 6 16 and the ground level is taken out as the output voltage Eo, this output voltage Eo starts from the center point of the semicircle shown in FIG.
- the phase shift circuit 630C of the subsequent stage constituting the oscillator 5E shown in FIG. 19 includes a FET 632 whose gate is connected to the input terminal of the phase shift circuit 630C, and a source ⁇ ⁇ ⁇ ⁇ drain of the FET 632.
- the resistor 646 in the phase shift circuit 630 C is for applying an appropriate bias voltage to the FET 632, and the capacitor 648 inserted between the phase shift circuits 630 C and 6100 C is.
- the current is ffl.
- This phase shift circuit 630C has the same basic configuration as that of the previous phase shift circuit 610C, and connects the CR circuit composed of the resistor 636 and the capacitor 634 in the preceding phase shift circuit 610C. The difference is that it is opposite to the connection of the CR circuit consisting of the capacity 6 14 and the fan 6 6.
- phase shift circuits 61 0 C and 63 0 C are shifted at a predetermined frequency.
- a signal in which the sum of the phase shift amounts to 360 ° is output by the whole.
- the non-inverting circuit 650 shown in Fig. 19 has a resistor 654 connected between the drain and the power supply, and a FET 65 connected between the source and ground. 2, transistor 658 with base connected to the drain of FET 652 and collector connected to the source via resistor 660, and appropriate bias voltage applied to FET 652 And a resistor 6 62.
- the capacitor 664 provided in the preceding stage of the non-inverting circuit 650 shown in FIG. 19 is used to remove the DC component from the output of the subsequent phase shifting circuit 630 C. And only the AC component is manually input to the non-inverting circuit 650.
- the F ET 652 When the AC ⁇ 2 is input to the gate, the F ET 652 outputs an opposite-phase ⁇ sign from the drain. Also, when the opposite phase ⁇ is input to the base of the transistor 658, the phase of the signal further inverted, that is, the phase of the signal input to the gate of the FE ⁇ 652 becomes in-phase. Is output from the collector, and this in-phase signal is output from the non-inverting circuit 650. The output of the non-inverting circuit 650 is taken out from the output terminal 92 as the output of the oscillator 510, and is fed back to the input side of the preceding phase shift circuit 610C via the resistor 670. I have.
- the amplification of the above-described non-inverting circuit 650 is determined by the resistance values of the above-described resistors 654, 656, and 660. By adjusting the resistance values of these resistors, the amplification degree of FIG.
- the loop gain of the feedback loop formed by including the two phase shift circuits 61 0 C and 63 0 C and the resistor 67 0 shown in Fig. 10 can be set to 1 or more h.
- FIG. 22 is a circuit diagram showing a configuration of a phase shift circuit including a condenser microphone.
- FIG. 22 (A) shows a configuration of a preceding phase shift circuit 710C included in the FM modulator 1B. I have.
- This phase shift circuit 710C is a capacitor microphone which is composed of a capacitor circuit 614 and a resistor 616 in the former stage phase shift circuit 610C included in the oscillator 5E shown in FIG. It has a configuration in which it is replaced with a CR circuit consisting of 614-1 and capacity 614-2 and resistor 616.
- This change in the electrostatic landscape reflects the change in the rising sound pressure of the condenser microphone 6114, and the time constant of the CR circuit included in one of the phase shift circuits 7 10 C according to the sound pressure changes. Because of the change, the frequency of the oscillation output also changes. In other words, by using the condenser microphone 614-1 as a part of the CR circuit, FM-modulated signals can be easily obtained.
- a condenser microphone is not provided in the preceding phase shift circuit, but a condenser microphone may be included in the subsequent phase shift circuit. That is, the phase shift circuit 630 C of the subsequent stage constituting the oscillator 5 E shown in FIG. 19 is replaced with the phase shift circuit 730 C shown in FIG. 22 ( ⁇ ) instead of the capacitor 634 in the phase shift circuit 630 C.
- the condenser microphone 634-1 and the capacity 634-2 may be replaced by rivers.
- the two phase shift circuits are both configured to include the CR circuit, but one of the phase shift circuits that does not include the condenser microphone can be replaced with a phase shift circuit that includes an LR circuit. .
- FIG. 23 is a circuit diagram showing a configuration of a phase shift circuit 610 L that can be replaced with the preceding phase shift circuit 610 C included in the generator & generator 5 shown in FIG.
- the phase shift circuit 610 L shown in FIG. 23 is a CR circuit consisting of the capacitor 614 and the resistor 616 in the phase shift circuit 610 C shown in FIG. It has a configuration in which it is replaced with an LR circuit consisting of an inductor 617. Connect the voltage across resistor 6 16 to VR1, Assuming that the pressure at both ends of the inductor 6 17 is VL1, as shown in FIG.
- the oblique side is twice the voltage E i, the r end of the resistor 6 16 ⁇ pressure VR1 and both ends of the inductor 6 17
- the voltage VL1 forms a quotient triangle forming two sides that are orthogonal to each other.
- this output voltage E o is the center of the half circle shown in Fig. 24. It can be expressed as a vector with the point as the starting point and the end point at the-point of the circumference I: where the voltage VR1 and the voltage VL1 intersect.
- the amplitude of the output signal is constant regardless of the frequency, and the phase shift It can be seen that the data amount is represented by 07 shown in FIG.
- phase shift circuit 6110C and 610L are both a (1 ⁇ Ts) / (1 + Ts).
- s jw
- a is the gain of each phase shift circuit.
- phase shift circuit 610L is equivalent to the phase shift circuit 610C, and replacing the phase shift circuit 610C with the phase shift circuit 610L is an ": I function.” Therefore, in the oscillator 5E shown in FIG. 19, the phase shift circuit 6100 of the preceding stage is replaced with the phase shift circuit 6100L shown in FIG. By replacing 0 C with the phase shift circuit 730 C shown in Fig. 22 (B) that includes a capacitor mark, each of the two phase shift circuits becomes LR
- An FM modulator including the IiiJ path can be configured.
- FIG. 25 is a circuit diagram showing a configuration of a phase shift circuit 630 L which can be replaced with a subsequent phase shift circuit 630 C included in the oscillator 5 E shown in FIG.
- the phase shift circuit 63 0 L shown in Fig. 25 is connected to the CR circuit consisting of the resistor 6 36 and the capacitor 63 4 in the phase shift circuit 63 0 C shown in Fig. It has a configuration in which it is replaced with an LR circuit consisting of 3 7 and resistor 6 3 6. Assuming that the voltage between both ends of the inductor 6 3 7 is VL2 and the voltage between both ends of the resistor 6 3 6 is VR2, as shown in Fig.
- phase shift circuit 63 0 C shown in FIG. 19
- time constant of the LR circuit in the phase shift circuit 63 0 L shown in FIG.
- the transfer functions of these phase shift circuits 63 0 C and 63 0 L are both 1 a (1-T s) / (1 + T s).
- phase shift circuit 630L is equivalent to the phase shift circuit 630C, and the phase shift circuit 630C can be replaced with the phase shift circuit 630L. Therefore, in the oscillator 5E shown in FIG. 19, the subsequent phase shift circuit 63 0 C is replaced with the phase shift circuit 63 0 L shown in FIG. By replacing 0 C with the phase shift circuit 710 C shown in Fig. 22 (A), which includes a capacitor microphone, each of the two shift circuits includes an LR circuit or a CR circuit. An FM modulator can be configured.
- the oscillation operation is performed at a wave number where the total phase shift amount of the two phase-shifted zero paths is 360 °.
- the oscillation operation may be performed at a frequency at which the sum of the phase shift views by the two phase shift circuits is 180 °.
- FIG. 27 is a circuit diagram of an oscillator configured using two phase shift circuits and a phase inversion circuit.
- the oscillator 5F shown in the figure is composed of a cascade connection of two stages of the phase shift circuit 610C at the preceding stage in the oscillator 5 1 shown in Fig. 19, and the FET 682 and the resistor 6884 and A phase inverting circuit 680 consisting of 686 is connected, and the output of the phase inverting circuit 680 is fed back to the human side of the previous stage via the resistor 670 through the phase shifter [J path 610 C]. You.
- the phase inversion circuit 680 Since the signal is inverted by the phase inversion circuit 680, when the total phase shift amount of the two phase shift circuits 610C is 180 °, the phase shift when the circuit goes through a closed loop is completed. In this case, the feedback amount becomes 360 °, and the loop gain of the feedback loop at this time * 1 By setting, a predetermined swing operation is performed.
- the sound was collected by the condenser microphone. It is possible to configure an FM modulator using sound ⁇ for an FM modulation signal.
- one of the two phase shift circuits 6 10 C included in the oscillator 5 F is replaced with the phase shift circuit 7 10 C shown in FIG. 22 (A), and the other is replaced with the phase shift circuit 7 10 C shown in FIG.
- the FM modulator may be configured in place of the phase shift path 6101L shown.
- FIG. 28 is a circuit diagram of another oscillator configured by cascade-connecting two phase shift circuits and a phase inversion circuit.
- the oscillator 5G shown in the figure has a two-stage cascade connection of the subsequent phase shift circuit 63 0 C in the oscillator 5 E shown in FIG. 19, and a phase inversion circuit 680 connected to the subsequent stage. Then, the output of the phase inversion circuit 680 is fed back to the input side of the preceding phase shift circuit 630 C via the resistor 670.
- phase shift circuit 680 Since the signal is inverted by the phase shift circuit 680, when the total phase shift amount of the two phase shift circuits 63 ° C reaches 180 °, the phase shift when the circuit goes through a closed loop is completed. 3600.
- a predetermined oscillation operation is performed by setting the loop gain of the feedback loop at this time to 1 or less.
- the sound collected by the condenser microphone can be obtained.
- one of the two phase shift circuits 630C included in the oscillator 5G is replaced with a phase shift circuit 730C shown in FIG. 22 (B), and the other is replaced with the phase shift circuit shown in FIG.
- the FM modulator may be configured by replacing the phase shift circuit 630 L shown in FIG.
- FIG. 29 is a circuit diagram showing the detailed configuration of the tenth embodiment of the FM modulator c .
- the FM modulator 1C shown in FIG. 29 does not change the phase of the input AC signal. Performs a total of 360 ° phase shift at the specified frequency with the non-inverting circuit 850 output It is configured to include two phase shift circuits 910C and 830C, and a feedback resistor 870.
- the non-inverting circuit 850 functions as a buffer circuit, and includes, for example, an emitter follower circuit, a source follower circuit, and the like.
- the non-inverting circuit 850 is omitted and the FM modulator is omitted. 1 C may be configured.
- the FM modulator 1C shown in the 29th is configured to include a condenser microphone (details will be described later), and FM-modulates the sound collected by the condenser microphone and outputs it.
- amplifier 2 and antenna 3 are connected after FM modulator 1C, and the output of FM modulator 1 is amplified by amplifier 2 and sent out from antenna 3 to the air. It would be an FM wireless microphone.
- the signal may be transmitted to the transmission line 400 via the transmitting driver 4 as shown in FIG.
- FIG. 30 shows the condenser microphone 8 14-1 and the capacity 8 1 4-2 included in the FM modulator i3 ⁇ 4: 1 C shown in FIG.
- FIG. 3 is a circuit diagram showing a configuration of an oscillator 5H simplified in place of FIG.
- the phase shift circuit 810C at the front stage shown in the figure includes a differential amplifier 812 that amplifies the differential voltage of the two inputs with a predetermined amplification and outputs the amplified signal, and a phase II that converts the phase of the input AC signal to a predetermined II.
- the capacitor 814 and the resistor 816 which are shifted to the non-inverting input terminal of the differential amplifier 812 and the voltage level of about 1 Z2 without changing the phase of the input AC signal.
- a resistor 818 and 820 which are input to the inverting input terminal of the differential amplifier 812.
- FIG. 31 is a vector diagram showing the relationship between the input / output voltage of the phase shift circuit 8100 C shown in FIG. 30 and the voltage appearing in the capacity and the like.
- the voltage VR1 appearing across the resistor 8 16 and the voltage The voltage VC1 appearing at both ends is 90 ° out of phase with each other, and the vector sum of these is equivalent to the input voltage Ei of the phase shift circuit 8100C. Accordingly, when the amplitude of the input voltage Ei is constant and only the frequency changes, the voltage VR1 across the resistor 8 16 and the voltage VC1 across the capacitor 8 14 along the circumference of the semicircle shown in FIG. Changes.
- the voltage applied to the non-inverting input terminal of the differential amplifier 812 (the voltage VC1 across the capacitor 814) to the voltage applied to the inverting input terminal (the voltage Ei / 2 across the resistor 820) is calculated.
- the difference obtained by the torque is the difference voltage Eo '.
- This differential voltage Eo ' can be represented by a vector whose center point is the starting point and whose end point is a point on the circumference where voltage VC1 and voltage VR1 intersect in the semicircle shown in Fig. 31. And its size is equal to the radius of the semicircle Ei / 2.
- the output voltage Eo of the differential amplifier 812 is obtained by amplifying the differential voltage Eo ′ with a predetermined amplification factor. Therefore, the above-described phase shift circuit 8100C operates as an all-pass circuit, in which the output voltage Eo is constant regardless of the frequency of the input voltage Ei. Further, as is clear from FIG. 31, since the voltage VC1 and the voltage VR1 intersect at right angles on the circumference, the phase difference between the input pressure Ei and the voltage VC1 varies from a frequency ⁇ of 0 to ⁇ . From 90 ° in the clockwise direction (phase lag direction) based on the human-power voltage Ei. To change. Then, the phase shift amount 09 of the entire phase shift circuit 8 10 C changes from 0 ° to 180 ° according to the frequency.
- the subsequent phase shift circuit 830C shown in FIG. 30 includes a differential amplifier 832 that amplifies the differential voltage of the two inputs with a predetermined amplification and outputs the amplified voltage, and shifts the phase of the input AC signal by a predetermined amount. And a capacitor 834 and a resistor 836 that are input to the non-inverting input terminal of the differential amplifier 832, and the voltage level is divided into about 1/2 without changing the phase of the input AC signal, and the differential amplifier is divided. It is configured to include resistors 838 and 840 input to the inverting input terminal of 812.
- FIG. 32 is a vector diagram showing the relationship between the input / output voltage of the phase shift circuit 830C shown in FIG. 30 and the voltage appearing in the capacity and the like.
- the voltage VC2 appearing at both ends of the capacitor 834 and the voltage VR2 appearing at both ends of the resistor 836 are out of phase with each other by 90 °. Is the input voltage E i. Therefore, when the amplitude of the input signal is constant and only the frequency changes, the voltage VC2 across the capacitor 834 and the voltage VR2 across the resistor 836 along the circumference of the semicircle shown in FIG. Changes.
- the voltage applied to the non-inverting input terminal of the differential amplifier 832 (the voltage VR2 across the resistor 836) and the voltage applied to the inverting input terminal (the voltage E i / 2 ) Is the difference voltage E o '.
- This difference voltage E o ' is represented by a vector whose center point is the start point and whose end point is a point on the circumference where voltage VR2 and voltage VC2 intersect in the semicircle shown in Fig. 32. Whose size is equal to the radius of the semicircle E i / 2.
- the output voltage E o of the differential amplifier 832 is obtained by amplifying the difference voltage E o 'with a predetermined amplification factor. Therefore, the above-described phase shift circuit 830C operates as an all-pass circuit, since the output voltage E o is constant regardless of the frequency of the input signal.
- the phase difference between the input voltage Ei and the voltage VR2 is 180 as it changes. To 270 °. Then, the phase shift amount 010 of the entire phase shift circuit 830C changes from 180 ° to 360 ° according to the frequency. In this way, the phase is shifted by a predetermined amount at each of the two phase shift circuits 8100C and 8300C, and the two phase shift circuits 8100C and 8300C are shifted at a predetermined frequency. A signal is output in which the sum of the phase shifts -M is 360 °.
- the FM modulator 1 C shown in FIG. 29 includes a phase shift circuit 8 10 C in the preceding stage included in the oscillator 5 H shown in FIG.
- the FM modulator 6 having such a configuration will be described.
- FIG. 33 is a circuit diagram showing a configuration of a phase shift circuit including a condenser microphone.
- FIG. 33A shows a configuration of a phase shift circuit 910C of a preceding stage included in the FM modulator 1C. It is shown.
- This phase shift circuit 910C is a CR circuit comprising a capacitor 814 and a resistor 816 in the previous phase shift circuit 8100C included in the oscillator 5H shown in FIG. Is replaced by a CR circuit consisting of a condenser microphone 8 14-1 and a capacitor 8 14-2 and a resistor 8 16.
- a condenser microphone is provided in the preceding phase shift circuit, but a condenser microphone may be included in the subsequent phase shift circuit. That is, the subsequent phase shift circuit 830 C constituting the oscillator 5 shown in FIG. 30 is connected to the phase shift circuit 930 C shown in FIG. 3311 ( ⁇ ) (the capacity of the capacitor 834 in the phase shift circuit 830 C). Alternatively, a condenser microphone 834-1 and a capacitor 834-2 may be used.
- the above-mentioned FM modulator 1C has the two phase shift circuits both including the CR circuit, it is also possible to replace the phase shift circuit not including the condenser microphone with the LR [phase shift circuit including the port I path]. it can.
- FIG. 34 is a circuit diagram showing a configuration of a phase shift circuit 8110L replaceable with the preceding phase shift circuit 8100C included in the oscillator 5 shown in FIG.
- the phase shift circuit 810 L shown in FIG. 34 is different from the phase shift circuit 810 C shown in FIG. 30 in that a CR circuit consisting of a capacitor 814 and a resistor 8 16
- the configuration is such that it is replaced with an LR circuit consisting of an inductor 8 17.
- phase circuits 810C and 810L are both a (1—Ts) / (1 + Ts).
- s jw
- a is the gain of each phase shift circuit.
- phase shift circuit 8101L is equivalent to the phase shift circuit 8110C, and the phase shift circuit 810C can be replaced with the phase shift circuit 8101L. Therefore, in the oscillator 5H shown in FIG. 30, the phase shift circuit 810C in the preceding stage is replaced with the phase shift circuit 810C shown in FIG.
- phase shift circuit 930C shown in Fig. 33 (B) which includes the phase shifter, each of the two phase shift circuits can constitute an FM modulator including an LR circuit or a CR circuit. it can.
- FIG. 35 is a circuit diagram for reducing the configuration of the phase shift circuit 830 L which can be replaced with the subsequent phase shift circuit 830 C included in the oscillator 5 H shown in FIG.
- the phase shift circuit 830 L shown in FIG. 35 is different from the phase shift circuit 830 C shown in FIG. 30 in that a CR circuit consisting of a resistor 836 and a capacitor 834 is connected to the inductor circuit. It has a configuration in which it is replaced by an LR circuit consisting of 837 and a resistor 836.
- phase shift circuit 830 C shown in FIG. 30
- time constant of the LR circuit in the phase shift circuit 830 L shown in FIG. the transfer functions of these phase shift circuits 830C and 830L are both -a (1-Ts) / (1 + Ts).
- the phase shift circuit 830 L is equivalent to the phase shift circuit 830 C, and it is possible to replace the phase shift circuit 830 C with the phase shift circuit 830 L. Therefore, in the oscillator 5H shown in FIG. 30, the subsequent phase shift circuit 830C is replaced with the phase shift circuit 830L shown in FIG. By replacing 0 C with the phase shift circuit 910 C shown in Fig. 33 (A) including a capacitor microphone, the phase shift circuit including the LR circuit and the phase shift circuit including the CR circuit are connected vertically. This makes it possible to construct a modified FM modulation device.
- the oscillation operation is performed at a frequency at which the sum of the phase shifts by the two phase shift circuits becomes 360 °.
- the oscillation operation may be performed at a frequency at which the sum of the phase shift amounts of the two phase shift circuits is 180 °.
- FIG. 36 is a circuit diagram showing a configuration of an oscillator configured using two phase shift circuits and a phase inversion circuit.
- the oscillator 5J shown in the figure has a phase inverting circuit 880 for inverting the phase of an input AC signal and outputting the inverted signal, and a total of 180 at a predetermined frequency. It is configured to include two phase shift circuits 8100 C for performing the phase shift of the above and a feedback resistor 870.
- the phase relationship between the input and output signals of the two phase shift circuits 8100C is as described with reference to FIG. 31.
- the phase shift by the entire two phase shift circuits 8100C is performed.
- the sum of fi is 180 °.
- phase inversion circuit 880 connected in front of the two phase shift circuits 810C inverts the phase of the input AC signal.
- an emitter ground circuit and a source ground c thus realized by a circuit that combines a circuit or an operational amplifier resistor, the phase inversion circuit 8 8 0 for the phase of the signal is inverted by the phase shift amount of the interleaf ten by two phase shifting circuits 8 1 0 C Is 180 °, the phase shift amount when the circuit goes through the closed loop becomes 360 °, and the specified oscillation operation is performed by setting the loop gain of the feedback loop to 1 or more at this time. .
- the signal is collected by the condenser microphone. It is possible to configure an FM modulator that uses the sound that has been emitted for the FM modulation signal.
- one of the two phase-shift circuits 8100C included in the oscillator 5J is replaced with the phase-shift circuit 9110C shown in FIG. 33 (A) [H] and the other is shifted to the third
- the FM modulator may be configured by replacing the phase shift circuit 8101L shown in FIG.
- FIG. 37 is a circuit diagram of another oscillator configured using two phase shift circuits [nl path and phase inversion path].
- the oscillator 5K shown in the same figure is connected in cascade using two phase shift circuits 830C at the subsequent stage in the oscillator 5H shown in FIG.
- a phase inversion circuit 880 is connected to the input side, and the output of the subsequent phase shift circuit 830C is fed back to the input side of the phase inversion circuit 880 via the feedback resistor 870.
- the phase inverting circuit 880 Since the signal is inverted by the phase inverting circuit 880, when the total phase shift amount of the two phase shift circuits 830C is 180 °, the phase shift when the circuit goes through a closed loop is completed. 3600. By setting the loop gain of the feedback loop to 1 or more h at this time, a predetermined generating operation is performed.
- the oscillators 5 C, 5 D, 5 E, 5 F, 5 G, 5 H, 5 J, 5 K, etc. are a non-inverting circuit and two phase shifting circuits or a phase inverting circuit and two phase shifting circuits. And a predetermined tuning operation by setting the total phase shift to 360 ° at a predetermined frequency by all three connected circuits. . Therefore, focusing only on the amount of phase shift, there is a certain degree of freedom as to which of the two phase shift circuits is used in the preceding stage or in which order the three circuits described above are connected. The connection order can be determined as needed.
- the capacity connected to the condenser microphone in a row may be omitted, the condenser microphone and the capacity may be connected in parallel, or the condenser microphone and a plurality of capacity may be combined in parallel and in series. .
- the capacitance of the connected capacitor is the latter.
- the FM constants of the FM modulator iS with a fixed carrier frequency are realized by fixing the element constant of each element in each phase shift path.
- the frequency may be arbitrarily changed.
- the phase shift circuit 110 By changing at least one of 6, 6 and 6 with a variable resistor to vary this resistance value, or by changing at least one of the capacity 14-2 or 34 in the phase shift circuit 110 C or 30 C.
- this capacitance By changing this capacitance by replacing it with a variable capacitance element, the frequency of the signal output from the FM modulator 1 can be changed by changing the amount of phase shift by each phase shift circuit. it can.
- variable resistor can be formed by using a channel resistance of a FET whose gate voltage can be changed, and a reverse bias voltage for applying a variable capacitance element between an anode and a cathode can be used.
- the inductor in the LR circuit may be configured using a variable inductor or a variable resistor.
- a voltage divider is connected to the output side of the phase shifter at the subsequent stage, and the manpower of the circuit is extracted as the oscillation output, and the divided output is fed back to the feedback resistor. It may be connected to the input side of the preceding phase shift circuit via 0 or the like.
- high stability is realized by configuring an FM modulator using phase shift circuits 110 C, 30 C, etc. using operational amplifiers.
- the phase shift circuit is used in 110 C, 30 C, etc.
- the offset voltage and the voltage gain are not required to be so high, so the differential with the specified gain is required.
- An amplifier may be used instead of the operational amplifier in the phase shift circuit.
- FIG. 38 is a circuit diagram in which a part necessary for the operation of the phase shift circuit in the configuration of the operational amplifier is extracted, and the whole operates as a differential amplifier having a predetermined gain.
- the differential amplifier shown includes a differential input stage 100 composed of FETs, a constant current circuit 102 for supplying a constant current to the differential input stage 100, and a predetermined current supplied to the constant current circuit 102. It comprises a bias circuit 104 for applying a bias voltage and an output amplifier 106 connected to the differential input stage 100.
- the multistage amplifier circuit for gaining the voltage gain included in the actual operational amplifier is omitted, so that the configuration of the differential amplifier can be simplified and a wider band can be achieved. In this way, by simplifying the circuit, the upper limit of the operating frequency can be reduced to 0 °, and accordingly, the upper limit of the output frequency of the FM modulator configured using this differential amplifier is increased. can do. Availability on litter
- a condenser microphone is installed in one of the two phase-shift circuits connected vertically, and the sound collected by the condenser microphone is directly FM-modulated and output, so that changes in the capacitance of the condenser microphone are converted to a voltage. This eliminates the need for additional circuits and the like, and can simplify the circuit configuration of the entire FM modulator.
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Abstract
An FM modulator (1) includes two phase shifting circuits (110C and 30C) and a capacitor microphone (14-1) is provided in the circuit (110C). The FM modulator (1) oscillates at the frequency at which the sum of the amounts of phase shift of the two circuits (110C and 30C) is 360°. When the capacitance of the microphone (14-1) is slightly changed by the sound pressure, the modulator (1) outputs FM signals from an output terminal (92) by directly utilizing the change for the frequency modulation.
Description
叨 細 書 Edifice
F M変調装置 技術分野 FM modulator Technical field
本発明は、 音声を F M変調して送出する F M変調装置に関する。 背景技術 TECHNICAL FIELD The present invention relates to an FM modulator for FM-modulating a sound and sending it out. Background art
F Mワイヤレスマイクは、 マイクロホンで集めた音声を増幅した後、 F M変調 を行い、 この F M変調された信号を増幅してアンテナから送信するものであり、 そのために、 マイクロホン、 低周波增幅回路、 F M変調回路、 高周波増幅回路、 アンテナを含んで構成されている。 The FM wireless microphone amplifies the sound collected by the microphone, performs FM modulation, amplifies the FM-modulated signal, and transmits it from the antenna. For this purpose, the microphone, low-frequency bandwidth circuit, and FM modulation It includes a circuit, a high-frequency amplifier circuit, and an antenna.
F M変調回路は一般に、 歪みの少ない正弦波を発生する L C発振器を用いて構 成されている。 この L C発振器としては、 例えばコルピッツ型等の各種の允振器 があり、 例えばその内部に含まれる L C共振回路のキャパシ夕をバリキヤップ (可変容量ダイオード) で構成し、 そのキャパシタンスを F M変調信号の電圧レ ベルの変動に応じて変化させることにより、 F M変調を行っている。 The FM modulation circuit is generally configured using an LC oscillator that generates a sine wave with little distortion. There are various types of LC oscillators such as Colpitts type LC oscillators. For example, the capacity of the LC resonance circuit included in the LC oscillator is composed of a varicap (variable capacitance diode), and the capacitance is the voltage of the FM modulation signal. FM modulation is performed by changing the level according to the fluctuation of the level.
ところが、 この種の L C発振器は、 発振周波数を大きく変えると発振出力の電 圧レベルも変化するという問題があり、 そのままでは実用的でない。 そのため、 この種の L C発 器を用いる場合には、 F Mキヤリァの振幅を一定にする回路が 必要となり、 冋路構成が複雑になる。 発明の開示 However, this type of LC oscillator has the problem that if the oscillation frequency is greatly changed, the voltage level of the oscillation output also changes, and it is not practical as it is. Therefore, when using this type of LC generator, a circuit for keeping the amplitude of the FM carrier constant is required, and the circuit configuration becomes complicated. Disclosure of the invention
本発明は、 このような課題を解決するために考えられたものであり、 その S的 は回路構成が簡単な F M変調装置を提供することにある。 The present invention has been conceived to solve such a problem, and an object thereof is to provide an FM modulator having a simple circuit configuration.
本発明の F M変調装置は、 差動増幅器と C R冋路とを含む全域通過型の 2つの 移相回路を備え、 これら 2つの移相回路を縦続接続して後段の前記移相回路の出 力を前段の前記移相回路の入力側に帰還させるとともに、 前記 2つの移相回路の いずれか一方に含まれる前記 C R冋路内のキャパシ夕としてコンデンサマイクを
用いることにより、 前記 2つの移相回路のいずれかから F M変調された信号を出 力する。 図面の簡単な説明 The FM modulator of the present invention includes two all-pass type phase shift circuits including a differential amplifier and a CR circuit, and cascade-connects these two phase shift circuits to output the output of the phase shift circuit at a subsequent stage. To the input side of the previous phase shift circuit, and a condenser microphone as a capacitor in the CR circuit included in one of the two phase shift circuits. By using this, an FM-modulated signal is output from one of the two phase shift circuits. BRIEF DESCRIPTION OF THE FIGURES
第 1図は、 第 1の実施形態の F M変調装置の構成を示す回路図、 FIG. 1 is a circuit diagram showing a configuration of an FM modulator of a first embodiment,
第 2図は、 F M変調装置の他の構成を示す図、 FIG. 2 is a diagram showing another configuration of the FM modulator,
第 3図は、 第 1図に示した F M変調装置に含まれるコンデンサマイク等を静電 容量が固定のキャパシタに置き換えた発振器の構成を示す回路図、 FIG. 3 is a circuit diagram showing a configuration of an oscillator in which a capacitor microphone or the like included in the FM modulator shown in FIG. 1 is replaced with a capacitor having a fixed capacitance;
第 4図は、 3図に す前段の移相回路の入出力電圧等に関するべク トル図、 第 5図は、 第 3図に示す後段の移相问路の入出力電圧等に関するべク トル図、 第 6図は、 コンデンサマイクを含む移相回路の構成を示す回路図、 Fig. 4 is a vector diagram related to the input / output voltage etc. of the preceding phase shift circuit shown in Fig. 3, and Fig. 5 is a vector related to the input / output voltage etc. of the subsequent phase shifting circuit shown in Fig. 3. Fig. 6 is a circuit diagram showing the configuration of the phase shift circuit including the condenser microphone.
第 7図は、 L R回路を含む移相回路の構成を示す回路図、 FIG. 7 is a circuit diagram showing a configuration of a phase shift circuit including an LR circuit,
第 8図は、 第 7図に す移相回路の入出力電圧等に関するべク トル図、 第 9図は、 L R回路を含む移相回路の他の構成を示す回路図、 FIG. 8 is a vector diagram relating to the input / output voltage of the phase shift circuit shown in FIG. 7, and FIG. 9 is a circuit diagram showing another configuration of the phase shift circuit including the LR circuit.
第 1 0図は、 第 9図に示す移相回路の入出力電圧等に関するべク トル図、 第 1 1図は、 移相回路内に分圧回路を設けた発振器の構成を示す回路図、 第 1 2図は、 発振器の他の構成を示す冋路図、 FIG. 10 is a vector diagram relating to the input / output voltage of the phase shift circuit shown in FIG. 9, and FIG. 11 is a circuit diagram showing a configuration of an oscillator having a voltage divider circuit in the phase shift circuit. FIG. 12 is a circuit diagram showing another configuration of the oscillator,
第 1 3図は、 第 1 2図に示した前段の移相回路と置き換え可能な移相回路の構 成を示す回路図、 FIG. 13 is a circuit diagram showing a configuration of a phase shift circuit which can be replaced with the preceding phase shift circuit shown in FIG. 12,
第 1 4図は、 第 1 2図に示した後段の移相回路と置き換え "J能な移相回路の構 成を示す回路図、 FIG. 14 is a circuit diagram showing the configuration of a phase shift circuit that can be replaced with the subsequent phase shift circuit shown in FIG.
第 1 5図は、 第 4の実施形態の F M変調装置の詳細構成を示す回路図、 第 1 6図は、 位相反転回路を含む発振器の構成を示す回路図、 FIG. 15 is a circuit diagram showing a detailed configuration of an FM modulator according to a fourth embodiment. FIG. 16 is a circuit diagram showing a configuration of an oscillator including a phase inversion circuit.
第 1 7図は、 位相反転回路を含む発振器の他の構成を示す回路図、 FIG. 17 is a circuit diagram showing another configuration of an oscillator including a phase inversion circuit,
第 1 8図は、 第 7の実施形態の F M変調装置の詳細構成を小す回路図、 第 1 9図は、 第 1 8図に示した F M変調装置に含まれるコンデンサマイク等を 静電容量が固定のキャパシ夕に置き換えた発振器の構成を示す回路図、 FIG. 18 is a circuit diagram for reducing the detailed configuration of the FM modulator according to the seventh embodiment, and FIG. 19 is a diagram illustrating a capacitor microphone and the like included in the FM modulator shown in FIG. Is a circuit diagram showing a configuration of an oscillator replaced with a fixed capacity,
第 2 0図は、 第 1 9図に示す前段の移相回路の入出力電圧等に関するべク トル 図、
第 2 1 Ι¾Ίは、 第 1 9図に示す後段の移相回路の入出力窑圧等に関するべク トル 図、 FIG. 20 is a vector diagram relating to input / output voltages and the like of the preceding phase shift circuit shown in FIG. Figure 21 is a vector diagram related to input and output pressure of the subsequent phase shift circuit shown in Fig. 19,
第 2 2図は、 コンデンサマイクを含む移相回路の構成を示す回路図、 第 2 3図は、 第 1 9図に示した前段の移相回路と置き換え可能な移相回路の構 成を示す问路図、 FIG. 22 is a circuit diagram showing the configuration of a phase shift circuit including a condenser microphone, and FIG. 23 is a diagram showing the configuration of a phase shift circuit that can replace the preceding phase shift circuit shown in FIG.问 Road map,
第 2 4図は、 第 2 3図に示す移相回路の入出力電圧等に関するべク トル図、 第 2 5図は、 第 1 9図に示した後段の移相回路と置き換え可能な移相问路の構 成を示す冋路図、 Fig. 24 is a vector diagram of the input / output voltage etc. of the phase shift circuit shown in Fig. 23, and Fig. 25 is a phase shifter that can be replaced with the subsequent phase shift circuit shown in Fig. 19 A circuit diagram showing the structure of the circuit,
第 2 6図は、 第 2 5図に示す移相 [口:!路の入出力亀圧等に関するべク トル図、 第 2 7図は、 位相反転回路を含む発振器の構成を示す回路図、 FIG. 26 is a vector diagram of the phase shift shown in FIG. 25 [port: input / output turret pressure of the road, etc.] FIG. 27 is a circuit diagram showing a configuration of an oscillator including a phase inversion circuit,
第 2 8図は、 位相反転回路を含む発振器の他の構成を示す回路図、 FIG. 28 is a circuit diagram showing another configuration of the oscillator including the phase inversion circuit,
第 2 9図は、 第 1 0の実施形態の F M変調装置の構成を示す回路図、 第 3 0図は、 第 2 9図に示した F M変調装置に含まれる F E Tとその周辺回路 を抵抗 ίιΰが固定の抵抗に置き換えた発振器の構成を示す回路図、 FIG. 29 is a circuit diagram showing a configuration of the FM modulator according to the tenth embodiment. FIG. 30 is a diagram showing the FET and its peripheral circuits included in the FM modulator shown in FIG. Is a circuit diagram showing a configuration of an oscillator replaced with a fixed resistor,
第 3 1図は、 第 3 0図に示す前段の移相回路の入出力 IS圧等に関するべク 卜ル 図、 FIG. 31 is a vector diagram relating to the input / output IS pressure and the like of the preceding phase shift circuit shown in FIG. 30;
第 3 2図は、 第 3 0図に示す後段の移相问路の人出力電圧等に関するべク トル 図、 FIG. 32 is a vector diagram relating to the human output voltage and the like of the subsequent phase shift circuit shown in FIG. 30,
第 3 3図は、 コンデンサマイクを含む移相回路の構成を示す回路図、 第 3 4図は、 L R回路を含む移相回路の構成を示す回路図、 FIG. 33 is a circuit diagram showing a configuration of a phase shift circuit including a condenser microphone. FIG. 34 is a circuit diagram showing a configuration of a phase shift circuit including an LR circuit.
第 3 5図は、 L R回路を含む移相回路の他の構成を示す回路図、 FIG. 35 is a circuit diagram showing another configuration of the phase shift circuit including the LR circuit,
第 3 6図は、 位相反転回路を含む発振器の構成を示す回路図、 FIG. 36 is a circuit diagram showing a configuration of an oscillator including a phase inversion circuit,
第 3 7図は、 位相反転回路を含む発振器の他の構成を示す冋路図、 FIG. 37 is a circuit diagram showing another configuration of the oscillator including the phase inversion circuit,
第 3 8図は、 オペアンプの構成の中で移相回路の動作に必要な部分を抽出した 回路図である。 発明を実施するための最良の形態 FIG. 38 is a circuit diagram in which a portion necessary for the operation of the phase shift circuit in the configuration of the operational amplifier is extracted. BEST MODE FOR CARRYING OUT THE INVENTION
以下、 本発明を適用した 実施形態について、 図面を参照しながら 体的に説 明する。
〔第 1の実施形態〕 Hereinafter, embodiments to which the present invention is applied will be specifically described with reference to the drawings. [First embodiment]
第 1図は、 本発明を適用した第 1の灾施形態の F M変調装 ¾の構成を示す回路 図である。 FIG. 1 is a circuit diagram showing a configuration of an FM modulator according to a first embodiment to which the present invention is applied.
同図に示す F M変調装置 1は、 所定の周波数において合計で 3 6 0 ° の位相シ フ トを行う 2つの移相回路 1 1 0 C、 3 0 Cと、 後段の移相回路 3 0 Cの出力を 前段の移相回路 1 1 0 Cの入力側に帰還させる帰還抵抗 7 0とを含んで構成され、 帰還抵抗 7 0は 0 Ωから有限の抵抗値を有している。 これら 2つの移相回路 1 1 0 C、 3 0 Cと帰 抵抗 7 0は発振器を構成している。 The FM modulator 1 shown in the figure includes two phase shift circuits 110 C and 30 C that perform a total of 360 ° phase shift at a predetermined frequency, and a phase shift circuit 30 C at the subsequent stage. And a feedback resistor 70 that feeds back the output of the first stage to the input side of the phase shift circuit 110 C of the preceding stage. The feedback resistor 70 has a finite resistance value from 0 Ω. These two phase shift circuits 110 C and 30 C and the return resistor 70 constitute an oscillator.
また、 第 1図に示す F M変調装置 1は、 後述するようにコンデンサマイクを備 えており、 このコンデンサマイクで集音した 声を F M変調して出力する。 The FM modulator 1 shown in FIG. 1 is provided with a condenser microphone as described later, and FM-modulates the voice collected by the condenser microphone and outputs it.
したがって、 例えば第 1図に示すように、 F M変調装置 1の後段に増幅器 2お よびアンテナ 3を接続し、 F M変調装^ 1の出力を増幅器 2によって増幅してァ ンテナ 3から空中に送出すれば F Mワイヤレスマイクとなる。 また、 アンテナ 3 から空屮に送出する場合の他、 第 2図に ]くすように送信ドライバ 4を介して伝送 路 4 0 0に送出するようにしてもよい。 Therefore, as shown in FIG. 1, for example, an amplifier 2 and an antenna 3 are connected after the FM modulator 1, and the output of the FM modulator 1 is amplified by the amplifier 2 and transmitted from the antenna 3 to the air. It would be an FM wireless microphone. In addition to the case where the signal is transmitted from the antenna 3 to the multiplex, the signal may be transmitted to the transmission line 400 via the transmission driver 4 as shown in FIG.
第 1図に示した F M変調装置 1の詳細について説明する前に、 その基本となる 発振器の動作について説明する。 Before describing the details of the FM modulator 1 shown in FIG. 1, the basic operation of the oscillator will be described.
第 3図は、 ¾ 1 ^に示した F M変調装置 1に含まれるコンデンサマイク 1 4一 1およびキャパシタ 1 4— 2を、 静電容最が固定のキャパシ夕 1 4に置き換えて 簡略化した究振器 5の構成を示す回路図である。 同図に示す発振器 5は、 所定の 周波数において合計で 3 6 0 ° の位相シフ トを行う 2つの移相回路 1 0 C、 3 0 Cと、 後段の移相回路 3 0 Cの出力を前段の移相回路 1 0 Cの入力側に帰還させ る帰還抵抗 7 0とを含んで構成されている。 Fig. 3 shows a simplified investigation by replacing the condenser microphone 141-1 and capacitor 14-2 included in the FM modulator 1 shown in ¾1 ^ with a fixed capacitance 14 capacitor. 6 is a circuit diagram showing a configuration of a device 5. FIG. The oscillator 5 shown in the figure includes two phase shift circuits 10 C and 30 C that perform a total of 360 ° phase shift at a predetermined frequency, and outputs the output of the subsequent phase shift circuit 30 C in the preceding stage. And a feedback resistor 70 for feeding back to the input side of the phase shift circuit 10C.
第 3図に示す発振器 5を構成する前段の移相回路 1 0 Cは、 差動増幅器の一種 であるオペアンプ (演算増幅器) 1 2と、 この移相回路 1 0 Cに入力された信号 の位相を所定鼂シフ トしてオペアンプ 1 2の非反転入力端子に入力する抵抗 1 6 およびキャパシ夕 1 4と、 この移相回路 1 0 Cの入力端とオペアンプ 1 2の反転 入力端子との問に挿入された抵抗 1 8と、 オペアンプ 1 2の出力端と反転入力端 子との間に挿入された抵抗 2 0とを含んで構成されている。
このような構成を有する移相回路 1 0 Cに交流信号が人力されると、 The phase shift circuit 10 C at the preceding stage constituting the oscillator 5 shown in FIG. 3 is composed of an operational amplifier (operational amplifier) 12, which is a type of differential amplifier, and the phase of a signal input to the phase shift circuit 10 C. Is shifted by a predetermined amount, and the resistor 16 and the capacitor 14 input to the non-inverting input terminal of the operational amplifier 12 and the input terminal of the phase shift circuit 10 C and the inverting input terminal of the operational amplifier 12 It is configured to include a resistor 18 inserted and a resistor 20 inserted between the output terminal of the operational amplifier 12 and the inverting input terminal. When an AC signal is manually input to the phase shift circuit 10 C having such a configuration,
プ 1 2の非反転人力端子にはキャパシ夕 1 4の両端に現れる電圧 VC1が印加され る。 また、 オペアンプ 1 2の 2入力端子間には電位差が生じないので、 オペアン プ 1 2の反転入力端子の電位と、 抵抗 1 6とキャパシ夕 1 4の接統点の電位とは 等しくなる。 したがって、 抵抗 1 8の両端には抵抗 1 6の π¾端電圧 VR1と同じ電 圧 VR1が現れる。 The voltage VC1 appearing at both ends of the capacity 14 is applied to the non-inverting human terminal of the step 12. Also, since there is no potential difference between the two input terminals of the operational amplifier 12, the potential of the inverting input terminal of the operational amplifier 12 is equal to the potential of the connection point between the resistor 16 and the capacitor 14. Therefore, the same voltage VR1 as the π¾ terminal voltage VR1 of the resistor 16 appears at both ends of the resistor 18.
ここで、 抵抗 1 8と抵抗 2 0の各抵抗値が等しいものとすると、 これら 2つの 抵抗 1 8、 2 0に同じ電流が流れるため、 抵抗 2 0の両端にも電圧 VR1が現れる c しかも、 これら 2つの抵抗 1 8、 2 0の各両端に現れる電圧 VR1はベク トル的に 同方向を有している。 したがって、 オペアンプ 1 2の反転入力端子 (電圧 V C1) を ¾準にして考えると、 第 4図に示すように、 抵抗 1 8の両端電圧 VR1をべク 卜 ル的に加算したものが入力電圧 E i に、 抵抗 2 0の電圧 VR1をベク トル的に減算 したものが出力電圧 E o になる。 また、 入出力電圧の大きさと位相の関係は、 入 力電圧 E i および出力電圧 E o を斜辺とし、 電圧 VR1の 2倍を底辺とする二等辺 三角形で表すことができ、 出力信号の振幅は周波数に関係なく入力信号の振幅と 同じであって、 位相シフ ト量は第 4図に示す 0 1 で表されることがわかる。 Here, when the resistor 1 8 as the resistance of the resistor 2 0 are equal, these two resistors 1 8, 2 0 for the same current flows through the resistor 2 0 both ends c moreover the voltage VR1 appears also show, The voltage VR1 appearing at both ends of these two resistors 18 and 20 has the same direction in vector. Therefore, considering the inverting input terminal (voltage V C1) of the operational amplifier 12 as a standard, as shown in FIG. 4, the sum of the voltage VR1 across the resistor 18 in a vector manner is the input voltage The output voltage Eo is obtained by vectorially subtracting the voltage VR1 of the resistor 20 from Ei. The relationship between the magnitude and the phase of the input / output voltage can be expressed by an isosceles triangle with the input voltage E i and the output voltage E o as the hypotenuse and the base as twice the voltage VR1. It can be seen that the amplitude is the same as the amplitude of the input signal regardless of the frequency, and the phase shift amount is represented by 01 shown in FIG.
また、 第 3図に示す発 ¾器 5を構成する後段の移相 [nj路 3 0 Cは、 差動増幅器 の一種であるオペアンプ 3 2と、 この移相 I ]路 3 0 Cに入力された信号の位相を 所定量シフ トしてオペアンプ 3 2の非反転入力端子に入力するキャパシ夕 3 4お よび抵抗 3 6と、 この移相回路 3 0 Cの入力端とオペアンプ 3 2の反転人力端子 との間に挿入された抵抗 3 8と、 オペアンプ 3 2の出力端と反転人力端子との間 に挿入された抵抗 4 0とを含んで構成されている。 この移相问路 3 0 Cは、 基本 的な構成は前段の移相回路 1 0 Cと同じであり、 移相回路 3 0 C内の C R回路を 構成するキャパシ夕 3 4と抵抗 3 6との接続順序は移相回路 1 0 C内の C R回路 を構成するキャパシ夕 1 4と抵抗 1 6との接続順序と反対である。 The phase shift [nj path 30 C] of the subsequent stage constituting the generator 5 shown in FIG. 3 is input to an operational amplifier 32 which is a type of differential amplifier and to this phase shift I] path 30 C. And a resistor 36 to shift the phase of the input signal by a predetermined amount to the non-inverting input terminal of the operational amplifier 32, and the input terminal of the phase shift circuit 30C and the inverting power of the operational amplifier 32. A resistor 38 is inserted between the output terminal of the operational amplifier 32 and the inverted terminal, and a resistor 40 is inserted between the output terminal of the operational amplifier 32 and the inverted human terminal. The basic configuration of the phase shift circuit 30 C is the same as that of the preceding phase shift circuit 10 C, and the capacity 3 4 and the resistor 3 6 which constitute the CR circuit in the phase shift circuit 30 C are connected. The order of connection is opposite to the order of connection between the capacity 14 and the resistor 16 constituting the CR circuit in the phase shift circuit 10C.
したがって、 キャパシ夕 3 4の両端電圧を VC2、 抵抗 3 6の両端電圧を VR2と して入出力 ¾圧の大きさと位相の関係を考えると、 第 5図に示すように、 入力電 圧 E i および出力電圧 E o を斜辺とし、 電 K VC2の 2倍を底辺とする二等辺三角 形で表すことができ、 出力信号の娠幅は周波数に関係なく入力 β号の振幅と同じ
であって、 位相シフ ト量は第 5図に示す 02 で表されることがわかる。 Therefore, considering the relationship between the magnitude and phase of the input / output voltage, assuming that the voltage across the capacitor 34 is VC2 and the voltage across the resistor 36 is VR2, as shown in Fig. 5, the input voltage E i And the output voltage E o is defined as the hypotenuse, and the base of the power K VC2 can be represented by an isosceles triangle. It can be seen that the phase shift amount is represented by 02 shown in FIG.
このようにして、 2つの移相回路 10 C、 30 Cのそれそれにおいて位相が所 定量シフ トされ、 所定の周波数において 2つの移相回路 10 C、 30 Cの全体に より位相シフ ト量の合計が 360 ° となる信号が出力される。 しかも、 後段の移 相回路 30 Cの出力は、 帰還抵抗 70を介して移相回路 10 Cの入力側に帰還さ れており、 帰還ループのループゲインを 1以上に設定することにより、 一巡した ときに位相シフ ト景の合計が 360 ° となるような周波数で正弦波発振が行われ る。 なお、 抵抗 1 8と 20の抵抗値を同じにするとともに抵抗 38と 40の抵抗 値を同じにした場合には、 移相回路 10 C、 30 Cの利得が 1になり、 しかも帰 還ループで生じる損尖もあるため、 ループゲインが 1より小さくなる。 このため、 ル一プゲインを 1以上にするためには、 抵抗 1 8よりも抵抗 20の抵抗値を ¾く、 あるいは抵抗 38よりも抵抗 40の抵抗値を高くする必要がある。 In this way, the phase of each of the two phase shift circuits 10 C and 30 C is shifted by a predetermined amount, and the amount of phase shift is determined by the entire two phase shift circuits 10 C and 30 C at a predetermined frequency. A signal totaling 360 ° is output. In addition, the output of the subsequent phase shift circuit 30 C is fed back to the input side of the phase shift circuit 10 C via the feedback resistor 70, and the loop has been completed by setting the loop gain of the feedback loop to 1 or more. Sometimes a sine wave oscillation occurs at a frequency such that the sum of the phase shift views is 360 °. If the resistors 18 and 20 have the same resistance value and the resistors 38 and 40 have the same resistance value, the gains of the phase shift circuits 10 C and 30 C become 1, and furthermore, in the feedback loop, The loop gain will be less than 1 because some peaks will occur. For this reason, in order to make the loop gain 1 or more, it is necessary to make the resistance value of the resistance 20 larger than the resistance 18 or make the resistance value of the resistance 40 larger than the resistance 38.
ところで、 第 1図に示した FM変調装置 1は、 ヒ述した発振器 5に含まれる前 段の移相回路 1 0 Cをコンデンサマイクを含む移相回路 1 1 0 Cに置き換えた構 成を有しており、 次にこのような構成を有する FM変調装置 1について説明する c 第 6図は、 コンデンサマイクを む移相回路の構成を示す回路図であり、 同図 (A) には FM変調装置 1に含まれる前段の移相回路 1 10 Cの構成が示されて いる。 この移相回路 1 1 0 Cは、 第 3図に した発^器 5に まれる前段の移相 回路 10 Cにおいて、 抵抗 1 6とキャパシタ 14からなる CR回路を抵抗 1 6と コンデンサマイク 14— 1およびキャパシ夕 14一 2からなる CR回路に置き換 えた構成を右している。 By the way, the FM modulator 1 shown in FIG. 1 has a configuration in which the phase shift circuit 10 C of the preceding stage included in the oscillator 5 described above is replaced with a phase shift circuit 110 C including a condenser microphone. Next, the FM modulator 1 having such a configuration will be described.c FIG. 6 is a circuit diagram showing the configuration of a phase shift circuit including a condenser microphone, and FIG. The configuration of the preceding phase shift circuit 110C included in the device 1 is shown. This phase-shift circuit 110 C is the same as the phase-shift circuit 10 C in the preceding stage included in the generator 5 shown in FIG. 3 except that a CR circuit comprising a resistor 16 and a capacitor 14 is connected to a resistor 16 and a capacitor microphone The right side shows the configuration in which a CR circuit consisting of 1 and 14 is replaced.
このように、 コンデンサマイク 14— 1を含んで CR回路を構成した場合には、 この CR回路の時定数 T ( = CR) がコンデンサマイク 14— 1が有する静電容 量に応じて微小変化する。 この静電容量の変化は、 コンデンサマイク 14一 1が 拾う音圧変化を反映しており、 音圧に応じて一方の移相回路 1 10 Cに含まれる CR回路の時定数が変化することから発振出力の周波数も変化する。 すなわち、 コンデンサマイク 14— 1を CR回路の -部に使用することにより、 簡単に FM 変調された信 を得ることができる。 したがって、 FM変調装置 1の回路構成自 体を簡略化することができる。
また、 F M変調装置 1を構成する 2つの移相回路 1 1 0 C、 3 0 Cのそれそれ は全域通過 问路であって、 F M変調されたキヤリァの周波数を変更した場合で あっても振幅がほぼ 定であり、 振幅変動を防止するための他の構成が不要とな る。 As described above, when the CR circuit is configured to include the condenser microphone 14-1, the time constant T (= CR) of the CR circuit slightly changes according to the capacitance of the condenser microphone 14-1. This change in capacitance reflects the change in sound pressure picked up by the condenser microphones 14-11, and the time constant of the CR circuit included in one phase shift circuit 110C changes according to the sound pressure. The frequency of the oscillation output also changes. In other words, FM-modulated signals can be easily obtained by using the condenser microphone 14-1 in the-part of the CR circuit. Therefore, the circuit configuration itself of the FM modulator 1 can be simplified. Further, each of the two phase shift circuits 110 C and 30 C constituting the FM modulator 1 is an all-pass circuit, and even if the frequency of the FM-modulated carrier is changed, the amplitude is changed. Is almost constant, and another configuration for preventing amplitude fluctuation is not required.
なお、 般のコンデンサマイクは、 平行板コンデンサの静電容量の変化を利用 したマイクロホンであり、 平行電極間に数十 Μ Ωの抵抗を介して所定の直流電圧 を印加し、 音圧による静電容量の変化に応じた電圧変化を電気信号として取り出 している。 したがって、 通常の使い方をする場合には所定の直流電圧を印加する 必要があるが、 本実施形態では音圧に応じて静電容量が変化するキャパシ夕とし てコンデンサマイク 1 4一 1を使用しているため、 所定の直流電圧を印加する必 要はない。 A general condenser microphone is a microphone that uses the change in capacitance of a parallel plate capacitor, applies a predetermined DC voltage between parallel electrodes through a resistance of several tens of Ω, and applies electrostatic capacitance due to sound pressure. The change in voltage according to the change in capacitance is extracted as an electrical signal. Therefore, it is necessary to apply a predetermined DC voltage for normal use, but in the present embodiment, the condenser microphone 1411 is used as a capacitor whose capacitance changes according to the sound pressure. Therefore, it is not necessary to apply a predetermined DC voltage.
また、 第 1図に した F M変調装置 1では、 前段の移相回路にコンデンサマイ クを含ませたが、 後段の移相回路にコンデンサマイクを含ませるようにしてもよ レ、。 すなわち、 第 3図に示した発振器 5を構成する後段の移相回路 3 0 Cを、 第 6図 (B ) に示す移相回路 1 3 0 C (移相回路 3 0 C内のキャパシ夕 3 4の代わ りにコンデンサマイク 3 4— 1とキャパシ夕 3 4— 2を用いたもの) に置き換え るようにしてもよい。 Further, in the FM modulator 1 shown in FIG. 1, a condenser microphone is included in the preceding phase shift circuit, but a condenser microphone may be included in the subsequent phase shift circuit. That is, the phase shift circuit 30 C of the subsequent stage constituting the oscillator 5 shown in FIG. 3 is replaced with the phase shift circuit 130 C shown in FIG. 6 (B) (capacity signal in the phase shift circuit 30 C). (The one using condenser microphone 34-1 and capacity 34-2 instead of 4) may be used.
また、 上述した F M変調装置 1は、 2つの移相回路をともに C R回路を含んで 構成したが、 コンデンサマイクを含まない一方の移相 ["|路を L R tol路を含む移相 回路に置き換えることもできる。 In the FM modulator 1 described above, the two phase shift circuits are both configured to include the CR circuit, but one of the phase shift circuits not including the condenser microphone is replaced with a phase shift circuit including the LR tol path. You can also.
第 7図は、 第 3図に示した発振器 5に含まれる前段の移相回路 1 0 Cと置き換 え可能な移相回路 1 0 Lの構成を示す回路図である。 第 7図に示す移相回路 1 0 Lは、 第 3図に示した移相回路 1 0 Cに対して、 抵抗 1 6とキャパシ夕 1 4から なる C R回路をインダクタ 1 7と抵抗 1 6からなる L R回路に置き換えた構成を 有している。 インダク夕 1 7の両端電圧を VL1、 抵抗 1 6の両端電圧を VR1とし て入出力電圧の大きさと位相の関係を考えると、 第 8図に示すように、 入力電圧 E i および出力電圧 E o を斜辺とし、 電圧 VL1の 2倍を底辺とする二等辺三角形 で表すことができ、 出力信号の振幅は周波数に関係なく人力信号の振幅と同じで あって、 位相シフ ト量は第 8図に示す 03 で表されることがわかる。
また、 第 3図に示した移相回路 1 0 C内の CR回路の時定数と第 7図に示した 移相回路 1 0 L内の LR回路の時定数をともに Tとすると、 これらの移相回路 1 0 C、 1 0 Lの伝達関数はともに ( 1— T s) / ( 1 +T s) となる。 ここで、 s = j ωである。 FIG. 7 is a circuit diagram showing a configuration of a phase shift circuit 10L which can be replaced with the preceding phase shift circuit 10C included in the oscillator 5 shown in FIG. The phase shift circuit 10 L shown in FIG. 7 is different from the phase shift circuit 10 C shown in FIG. 3 in that a CR circuit consisting of the resistor 16 and the capacitor 14 is connected to the inductor 17 and the resistor 16. LR circuit. Considering the relationship between the magnitude and phase of the input / output voltage assuming that the voltage across the inductor 17 is VL1 and the voltage across the resistor 16 is VR1, as shown in Fig. 8, the input voltage E i and the output voltage E o Can be represented by an isosceles triangle whose base is twice the voltage VL1.The amplitude of the output signal is the same as the amplitude of the human-powered signal regardless of the frequency, and the amount of phase shift is shown in Fig. 8. It can be seen that it is represented by 03 as shown. If the time constant of the CR circuit in the phase shift circuit 10C shown in FIG. 3 and the time constant of the LR circuit in the phase shift circuit 10L shown in FIG. The transfer functions of the phase circuits 10C and 10L are both (1-Ts) / (1 + Ts). Here, s = jω.
このように、 移相回路 1 0 Lは移相回路 1 0 Cと等価であり、 移相回路 1 0 C を移相回路 1 0 Lに置き換えることができる。 したがって、 第 3図に示した発振 器 5において、 前段の移相回路 1 0 Cを第 7図に示した移相回路 1 0 Lに置き換 えるとともに、 後段の移相回路 30 Cをコンデンサマイクを含む第 6図 (Β) に 示した移相 |π|路 1 3 0 Cに置き換えることにより、 2つの移相回路のそれそれが LR回路あるいは CR回路を含んだ FM変調装置を構成することができる。 Thus, the phase shift circuit 10L is equivalent to the phase shift circuit 10C, and the phase shift circuit 10C can be replaced with the phase shift circuit 10L. Therefore, in the oscillator 5 shown in FIG. 3, the phase shift circuit 10 C in the preceding stage is replaced with the phase shift circuit 10 L shown in FIG. 7, and the phase shift circuit 30 C in the latter stage is replaced by a condenser microphone. By replacing the phase shift | π | path 130 C shown in Fig. 6 (Β), each of the two phase shift circuits constitutes an FM modulator including an LR circuit or a CR circuit. Can be.
また、 第 9図は第 3図に^した発振器 5に含まれる後段の移相回路 3 0 Cと置 き換え可能な移相回路 30 Lの構成を示す回路冈である。 第 9図に示す移相回路 30 Lは、 第 3図に示した移相回路 30 Cに対して、 キャパシ夕 34と抵抗 36 からなる CRIHJ路を抵抗 36とインダク夕 37からなる LR回路に置き換えた構 成を ίしている。 抵抗 36の両端電圧を VR2、 インダク夕 37の両端電圧を VL2 として人出力電圧の大きさと位相の関係を考えると、 第 10図に示すように、 入 力電圧 Ei および出力電圧 Eo を斜辺とし、 電圧 VR2の 2倍を底辺とする二等辺 二角形で表すことができ、 出力信 の振幅は周波数に関係なく入力 ^号の振幅と 同じであって、 位 fflシフ ト は第 1 0図に示す 04 で表されることがわかる。 また、 第 3図に示した移相回路 30 C内の CR回路の時定数と第 9図に示した 移相回路 30 L内の LR冋路の時定数をともに Tとすると、 これらの移相回路 3 0 C、 30 Lの伝達関数はともに一 ( 1— T s) / ( 1 +T s) となる。 FIG. 9 is a circuit diagram showing a configuration of a phase shift circuit 30L that can be replaced with a subsequent phase shift circuit 30C included in the oscillator 5 shown in FIG. The phase shift circuit 30L shown in Fig. 9 is different from the phase shift circuit 30C shown in Fig. 3 in that the CRIHJ path consisting of the capacitor 34 and the resistor 36 is replaced by an LR circuit consisting of the resistor 36 and the inductor 37. It has a different configuration. Considering the relationship between the magnitude and phase of the human output voltage with the voltage across the resistor 36 as VR2 and the voltage across the inductor 37 as VL2, as shown in Fig. 10, the input voltage Ei and the output voltage Eo are The amplitude of the output signal is the same as the amplitude of the input ^ signal regardless of the frequency, and the position ffl shift is shown in Fig. 10. You can see that it is represented by 04. Assuming that the time constant of the CR circuit in the phase shift circuit 30C shown in FIG. 3 and the time constant of the LR circuit in the phase shift circuit 30L shown in FIG. The transfer functions of the circuits 30 C and 30 L are both 1 (1-T s) / (1 + T s).
このように、 移相回路 30 Lは移相回路 30 Cと等価であり、 移相回路 30 C を移相回路 30 Lに置き換えることができる。 したがって、 第 3図に示した発 器 5において、 後段の移相回路 30 Cを第 9図に示した移相回路 30 Lに置き換 えるとともに、 前段の移相回路 10 Cをコンデンサマイクを含む第 6図 (A) に 示した移相回路 1 1 0 Cに置き換えることにより、 2つの移相回路のそれそれが LR冋路あるいは CR问路を含んだ FM変調装置を構成することができる。 As described above, the phase shift circuit 30L is equivalent to the phase shift circuit 30C, and the phase shift circuit 30C can be replaced with the phase shift circuit 30L. Therefore, in the generator 5 shown in FIG. 3, the subsequent phase shift circuit 30C is replaced with the phase shift circuit 30L shown in FIG. 9, and the preceding phase shift circuit 10C includes a condenser microphone. By substituting the phase shift circuit 110C shown in FIG. 6 (A), each of the two phase shift circuits can constitute an FM modulator including an LR line or a CR line.
〔第 2の実施形態〕
第 3図に示した各移相回路は、 オペアンプの出力を抵抗を介して直接オペアン プの人力側に帰還させているが、 各オペアンプの出力端了-に分圧回路を接続して 分圧出力をオペアンプの入力側に帰還させてもよい。 [Second embodiment] In each phase shift circuit shown in Fig. 3, the output of the operational amplifier is directly fed back to the human side of the operational amplifier via a resistor.However, a voltage divider is connected to the output terminal of each operational amplifier to divide the voltage. The output may be fed back to the input side of the operational amplifier.
第 1 1図は移相回路内に分圧回路を設けた発振器の詳細構成を示す回路図であ る。 同図に示す移相 M路 2 1 0 C内のオペアンプ 1 2の出力端には抵抗 2 1およ び抵抗 2 3からなる分圧回路が接続され、 この分圧回路の分圧出力端は抵抗 2 0 を介してオペアンプ 1 2の反 fc人力端子と接^されている。 同様に、 移相回路 2 3 0 C内のオペアンプ 3 2の出力端には抵抗 4 1および抵抗 4 3からなる分圧回 路が接続され、 この分圧出力端は抵抗 4 0を介してオペアンプ 3 2の反転入力端 子と接続されている。 FIG. 11 is a circuit diagram showing a detailed configuration of an oscillator provided with a voltage dividing circuit in a phase shift circuit. The voltage dividing circuit composed of the resistors 21 and 23 is connected to the output terminal of the operational amplifier 12 in the phase shift M path 210C shown in FIG. It is connected to the anti-fc human input terminal of the operational amplifier 12 via the resistor 20. Similarly, the output terminal of the operational amplifier 32 in the phase shift circuit 230 C is connected to a voltage dividing circuit composed of the resistors 41 and 43, and the voltage dividing output terminal is connected to the operational amplifier via the resistor 40. 32 Connected to 2 inverting input terminal.
第 1 1図において、 抵抗 2 1の抵抗値を R 21、 抵抗 2 3の抵抗値を R 23とする と、 オペアンプ 1 2の出力電圧 E o と抵抗 2 1および抵抗 2 3からなる分圧回路 の分圧出力 E o ' との問には、 抵抗 2 0の抵抗値に対して R 21、 R 23が十分小さ いときは E o = ( 1 + R 21/R 23) E o ' の関係がある。 したがって、 R 21およ び R 23の値を調整することにより 1より大きなゲインが得られ、 しかもベク トル 図は第 4図に示した電圧 E o を分 出力 E o ' に置き換えればよいため、 周波数 が変化しても出力電圧 E o の &幅が一定であり、 位相のみを所定量シフ トするこ とができる。 移相回路 2 3 0 Cについても同様であり、 周波数が変化しても出力 亀圧 E o の抿幅を一定にしたまま位相のみを所定 fiシフ 卜することができる。 このように、 2つの移相回路内にそれそれ分圧回路を設けることにより、 抵抗 1 8と 2 0の抵抗値を同じにするとともに抵抗 3 8と 4 0の抵抗値を同じにした 場合であっても、 2つの移相回路を縦続接続して形成される帰還ループのループ ゲインを確実に 1以上にすることができ、 発振動作を安定化させることができる。 また、 第 1 1図に示す 2つの移相回路 2 1 0 C、 2 3 0 Cのいずれか -方が有 する C R回路内のキャパシ夕をコンデンサマイクを用いて構成することにより、 第 1図と同様に F M変調装置を構成することができる。 In Fig. 11, assuming that the resistance value of resistor 21 is R21 and the resistance value of resistor 23 is R23, the voltage dividing circuit consisting of the output voltage Eo of the operational amplifier 12 and the resistors 21 and 23 When R21 and R23 are sufficiently small with respect to the resistance value of the resistor 20, the relation of Eo = (1 + R21 / R23) Eo ' There is. Therefore, a gain greater than 1 can be obtained by adjusting the values of R21 and R23, and the vector diagram can be obtained by replacing the voltage Eo shown in Fig. 4 with the partial output Eo '. Even when the frequency changes, the & width of the output voltage E o is constant, and only the phase can be shifted by a predetermined amount. The same applies to the phase shift circuit 230 C. Even if the frequency changes, it is possible to shift only the phase by a predetermined fidelity while keeping the width of the output pressure E o constant. Thus, by providing a voltage dividing circuit in each of the two phase shift circuits, when the resistance values of the resistors 18 and 20 are the same and the resistance values of the resistors 38 and 40 are the same. Even if there is, the loop gain of the feedback loop formed by cascading the two phase shift circuits can be reliably set to 1 or more, and the oscillation operation can be stabilized. In addition, by configuring the capacity in the CR circuit of either of the two phase shift circuits 210 C and 230 C shown in FIG. 11 by using a condenser microphone, FIG. The FM modulator can be configured in the same manner as.
なお、 第 1 1図では、 第 3図に示した移相回路 1 0 C、 3 0 C内のオペアンプ 1 2、 3 2の出力端に分圧回路を接続する例を示したが、 第 7図および第 9図に 示す移相回路 1 0 L、 3 0 L内のオペアンプの出力端に分圧回路を接続して分圧
出力端を各オペアンプ 1 2、 3 2の人力側に帰還させる場合も、 第 1 1図に示し た発振器 5 Aと同様の安定した発振動作が行われる。 FIG. 11 shows an example in which a voltage dividing circuit is connected to the output terminals of the operational amplifiers 12 and 32 in the phase shift circuits 10 C and 30 C shown in FIG. A voltage divider is connected to the output terminals of the operational amplifiers in the phase shift circuits 10 L and 30 L shown in the figures and Fig. When the output terminal is fed back to the human side of each of the operational amplifiers 12 and 32, a stable oscillation operation similar to that of the oscillator 5A shown in FIG. 11 is performed.
〔第 3の実施形態〕 [Third embodiment]
第 1 2図は、 発振器の他の構成を示す回路図である。 同図に示す発振器 5 Bは、 所定の周波数において合計で 3 6 0 ° の位相シフ トを行う 2つの移相回路 4 1 0 C、 4 3 0 Cを含んで構成されている。 FIG. 12 is a circuit diagram showing another configuration of the oscillator. The oscillator 5B shown in the figure is configured to include two phase shift circuits 410C and 430C that perform a total of 360 ° phase shift at a predetermined frequency.
第 1 1図に示した発振器 5 Aは、 前段の移相回路 1 0 C内の抵抗 1 8と抵抗 2 0の各抵抗倘を同じに設定することで、 入力される交流信号の周波数が変わった ときの振幅変化を抑えている。 これに対し、 第 1 2図に示す発振器 5 Bに含まれ る前段の移相问路 4 1 0 Cは、 移相回路内に分 路を設けずに、 抵抗 1 8 ' の 抵抗値よりも抵抗 2 0 ' の抵抗値を大きく設定することにより、 移相回路 4 1 0 Cの利捋を 1より人きな値に設定している。 The oscillator 5A shown in Fig. 11 changes the frequency of the input AC signal by setting the resistances 倘 of the resistors 18 and 20 in the preceding phase shift circuit 10C to the same value. The amplitude change at the time of is suppressed. On the other hand, the phase shift circuit 4100 C in the preceding stage included in the oscillator 5 B shown in FIG. 12 does not have a shunt in the phase shift circuit and has a higher resistance than the resistance of the resistor 18 ′. By setting the resistance value of the resistor 20 'to a large value, the use of the phase shift circuit 410C is set to a value more unique than 1.
後段の移相回路 4 3 0 Cについても同様であり、 抵抗 3 8 ' の抵抗値よりも抵 抗 4 0 ' の抵抗値を大きく設定することで、 移相回路 4 3 0 Cの利得を 1より大 きな値に設定している。 The same applies to the subsequent phase shift circuit 430C, and by setting the resistance value of the resistor 40 'to be larger than the resistance value of the resistor 38', the gain of the phase shift circuit 430C is set to 1 It is set to a larger value.
ところで、 上述したように、 移相问路の利得が 1より人きな値になるように各 抵抗の値を設定すると、 入力される信号の周波数に応じて利得変動が生じるおそ れがある。 例えば、 前段の移相回路 4 1 0 Cについて考えると、 入力信号の周波 数が低い場合には移相回路 4 1 0 Cはボルテージホロヮ冋路となるためこのとき の利得は 1倍となるのに対し、 周波数が高い場合には移相回路 4 1 0 ( は反' 増 幅器となるためこのときの利得は— m倍 (mは抵抗 2 0 ' と抵抗 1 8 ' の抵抗比) となり、 入力信号の周波数が変化したときに移相回路 4 1 0 Cの利得も変化して 出力信 の振幅変動が生じる。 By the way, as described above, if the value of each resistor is set so that the gain of the phase shift circuit becomes a value more unique than 1, gain fluctuation may occur depending on the frequency of the input signal. For example, considering the preceding phase shift circuit 410C, when the frequency of the input signal is low, the phase shift circuit 410C becomes a voltage hollow path, and the gain at this time becomes 1 times. On the other hand, when the frequency is high, the phase shift circuit 4 10 (is an anti-amplifier, so the gain at this time is -m times (m is the resistance ratio between the resistance 20 'and the resistance 18'). When the frequency of the input signal changes, the gain of the phase shift circuit 410C also changes, and the amplitude of the output signal fluctuates.
このような振幅変動は、 オペアンプ 1 2の反転入力端子に抵抗 1 9を接続して、 入力信号の周波数が低い場合と高い ¾合の利得を一致させることにより抑えるこ とができる。 具体的には、 抵抗 1 8 ' の抵抗値を r、 抵抗 2 0 ' の抵抗値を m r とすると、 抵抗 1 9の抵抗値を m r / ( m— 1 ) に設定することにより、 入力信 号の周波数が 0と無限大のときの移相 0路 4 1 0 Cの各利捋を一致させることが できる。 同様に、 移相回路 4 3 0 Cについてもオペアンプ 3 2の反転入力端子に
所定の抵抗値を有する抵抗 3 9を接続することにより、 出力信号の振幅変動を抑 えることができる。 なお、 抵抗 1 9および抵抗 3 9の一方端はグランドレベル以 外の固定電位に接続してもよい。 Such amplitude fluctuation can be suppressed by connecting the resistor 19 to the inverting input terminal of the operational amplifier 12 and matching the gain when the frequency of the input signal is low with that when the frequency of the input signal is high. Specifically, assuming that the resistance of the resistor 18 'is r and the resistance of the resistor 20' is mr, the input signal is set by setting the resistance of the resistor 19 to mr / (m-1). When the frequency is 0 and infinity, the respective benefits of the phase shift 0 path 410C can be matched. Similarly, the phase shift circuit 4300C is also connected to the inverting input terminal of the operational amplifier 32. By connecting the resistor 39 having a predetermined resistance value, the amplitude fluctuation of the output signal can be suppressed. Note that one ends of the resistors 19 and 39 may be connected to a fixed potential other than the ground level.
また、 第 1 2図に示す 2つの移相回路 4 1 ◦ C、 4 3 0 Cのいずれか一方が有 する C R iP]路内のキャパシ夕をコンデンサマイクを用いて構成することにより、 第 1図と同様に F M変調装置を構成することができる。 In addition, the capacity in the CR iP] circuit, which is provided by either of the two phase shift circuits 41 C or 43 C shown in FIG. An FM modulator can be configured in the same manner as in the figure.
なお、 第 1 2図に示す発振器 5 Bは、 C R回路を含む移相回路 4 1 0 C、 4 3 0 Cを縱続接続しているが、 C R回路を L R回路に置き換えることも可能である c 例えば、 第 1 3図に示す移相回路 4 1 0 Lは第 1 2図に示した前段の移相回路 4 1 0 Cと等 itffiであり、 移相回路 4 1 0 Cを移相回路 4 1 0 Lに置き換えることが できる。 同様に、 第 1 4図に/ す移相回路 4 3 0 Lは第 1 2図に した後段の移 相回路 4 3 0 Cと等価であり、 移相回路 4 3 0 Cを移相回路 4 3 0 Lに置き換え ることができる。 第 1 2図に示した前段の移相回路 4 1 0 Cを第 1 3図に示す移 相回路 4 1 0 Lに置き換える場合には、 第 1 2図に示した後段の移相回路 4 3 0 C内のキャパシ夕 3 4をコンデンサマイクを用いて構成すればよい。 Although the oscillator 5B shown in FIG. 12 has a cascade connection of phase shift circuits 4110C and 4330C including a CR circuit, the CR circuit can be replaced with an LR circuit. c For example, the phase shift circuit 4 10 L shown in FIG. 13 is the same as the previous phase shift circuit 4 10 C shown in FIG. 12, and the phase shift circuit 4 10 C is a phase shift circuit. It can be replaced with 410 L. Similarly, the phase shift circuit 430 L shown in FIG. 14 is equivalent to the phase shift circuit 430 C of the subsequent stage shown in FIG. It can be replaced by 30 L. When replacing the preceding phase shift circuit 4100C shown in FIG. 12 with the phase shift circuit 410L shown in FIG. 13, the subsequent phase shift circuit 43 shown in FIG. The capacity 34 in 0 C may be configured using a condenser microphone.
〔第 4の実施形態〕 (Fourth embodiment)
第 1図に^した F M変調装^では、 2つの移相回路を合わせた位相シフ ト量が 所定の周波数において 3 6 0 ° となるようにしているが、 2つの移相问路を縦続 接続して形成される帰還ループの一部に、 位相を変化させない非反転回路を接続 して F M変調装置を構成してもよい。 In the FM modulator shown in Fig. 1, the combined phase shift of the two phase shift circuits is set to 360 ° at a predetermined frequency, but the two phase shift circuits are cascaded. The FM modulator may be configured by connecting a non-inverting circuit that does not change the phase to a part of the feedback loop formed as described above.
第 1 5図は、 F M変調装置の第 4の実施形態の詳細構成を示す回路図である。 同図に示す F M変調装置 1 Aは、 移相回路 1 1 0 Cと移相回路 3 0 Cを縦続接続 する点では第 1図に示した F M変調装置 1と同じであり、 後段の移相回路 3 0 C の出力側に非反転回路 5 0を接続する点で^ 1図に示した: P M変調装置 1と異な る。 FIG. 15 is a circuit diagram showing a detailed configuration of a fourth embodiment of the FM modulator. The FM modulator 1A shown in the figure is the same as the FM modulator 1 shown in FIG. 1 in that the phase shift circuit 110C and the phase shift circuit 30C are connected in cascade, and the subsequent phase shifter is used. FIG. 1 shows that the non-inverting circuit 50 is connected to the output side of the circuit 30 C. FIG.
この非反転回路 5 0は、 オペアンプ 5 2と抵抗 5 4および 5 6によって構成さ れており、 2つの抵抗 5 4、 5 6の抵抗比に応じた所定の利得を有している。 し たがって、 閉ループを形成した際の損失をこの利得で補うことができ、 帰還ル一 プのループゲインを容易に 1以上に設定することができる。 また、 非反転问路 5
0に^力増幅段としての機能を持たせることもできる。 The non-inverting circuit 50 includes an operational amplifier 52 and resistors 54 and 56, and has a predetermined gain according to a resistance ratio of the two resistors 54 and 56. Therefore, the loss at the time of forming the closed loop can be compensated by this gain, and the loop gain of the feedback loop can be easily set to 1 or more. In addition, the non-inverting path 5 It is also possible for 0 to have a function as a power amplification stage.
なお、 第 1 5図では、 一例として第 1図に示した F M変調装置 1に非反転回路 5 0を接続した構成を説明したが、 上述した各種の移相回路を任意の順序で縦続 接統して構成される各種の F M変調装置に第 1 5図に示す非反転回路 5 0を接続 してもよい。 In FIG. 15, as an example, the configuration in which the non-inverting circuit 50 is connected to the FM modulator 1 shown in FIG. 1 has been described, but the above-described various phase shift circuits are cascaded in an arbitrary order. The non-inverting circuit 50 shown in FIG. 15 may be connected to various FM modulators configured as described above.
〔第 5の実施形態〕 (Fifth embodiment)
上述した各種の F M変調装置においては、 2つの移相回路による位相シフ ト量 の^,汁が3 6 0 ° となる周波数の允^動作を行っていたが、 閉ループ内に位相反 転回路を接絞することにより、 2つの移相回路による位相シフ ト量の合計が 1 8 0。 となる周波数で ¾振動作を行わせるようにしてもよい。 In the various FM modulators described above, the phase shift amount is controlled by two phase shift circuits, and the operation of adjusting the frequency at which the juice becomes 360 ° is performed.However, a phase inversion circuit is provided in a closed loop. By closing the aperture, the total amount of phase shift by the two phase shift circuits is 180. The vibration operation may be performed at a frequency such that:
第 1 6図は、 2つの移相回路と位相反転 Lnj路とを縦続接絞して構成した発振器 の In]路囟である。 同図に小す発振器 5 Cは、 第 3図に示した発 ¾器 5内の前段の 移相回路 1 0 Cを 2段縦続接続するとともに、 その後段にオペアンプ 8 2および 抵抗 8 4、 8 6からなる位相反転冋路 8 0を接続し、 この位相反転回路 8 0の出 力を' Ηϋ抵抗 7 0を介して前段の移相回路 1 0 Cの人力側に帰還させている。 位相反 fc0路 8 0によって ϋ が反転するため、 2つの移相回路 1 0 Cによる 位相シフ ト量の ΠΙ·が 1 8 0 ° となるときに、 閉ループを一巡したときの位相シ フ ト罱が 3 6 0 ° となり、 このときの' ilループのループゲインを 1以上に設定 することにより所定の発振動作が行われる。 FIG. 16 shows an In] path of an oscillator configured by cascading two phase shift circuits and a phase inversion Lnj path. The oscillator 5C shown in the figure is composed of a cascade connection of two stages of the preceding phase shift circuit 10C in the generator 5 shown in FIG. 3, and an operational amplifier 82 and resistors 84, 8 A phase inversion circuit 80 composed of 6 is connected, and the output of the phase inversion circuit 80 is fed back to the human-powered side of the preceding phase shift circuit 10 C via a resistance 70. Since 反 is inverted by the phase reversal fc0 path 80, when the phase shift amount 2 · by the two phase shift circuits 10 C becomes 180 °, the phase shift when the circuit goes through a closed loop Is 360 °, and a predetermined oscillation operation is performed by setting the loop gain of the loop at this time to 1 or more.
したがって、 発振器 5 Cに含まれる 2つの移相回路 1 0 Cのいずれか 方を第 6 f¾| ( A ) に示した移相回路 1 1 0 Cに ISき換えることにより、 コンデンサマイ クによって集音した音; i 'を F M変調信 に用いた F M変調装置を構成することが できる。 あるいは、 発振器 5 Cに含まれる 2つの移相回路 1 0 Cのいずれか一方 を第 6 ( A ) に示した移相回路 1 1 0 Cに置き換えるとともに、 他方を第 7図 に示した移相回路 1 0 Lに置き換えて F M変調装置を構成してもよい。 Therefore, by replacing one of the two phase shift circuits 10 C included in the oscillator 5 C with the phase shift circuit 110 C shown in the sixth f¾ | It is possible to construct an FM modulator using the sound that has been sounded; i ′ for the FM modulation signal. Alternatively, one of the two phase shift circuits 10 C included in the oscillator 5 C is replaced with the phase shift circuit 110 C shown in FIG. 6 (A), and the other is shifted as shown in FIG. The FM modulator may be configured in place of the circuit 10L.
〔第 6の実施形態〕 (Sixth embodiment)
第 1 6図に示した発振器 5 Cは、 移相回路 1 0 Cを縦続接続する例を示したが、 第 3図に示す後段の移相回路 3 0 Cを縦続接続してもよい。 Although the oscillator 5C shown in FIG. 16 shows an example in which the phase shift circuit 10C is cascaded, the oscillator 5C shown in FIG. 3 may be cascaded with the subsequent phase shift circuit 30C.
1 7図は、 位相反転 IHJ路を内部に含む発振器の他の構成を示す回路図である。
同図に示す発振器 5 Dは、 第 3図に小した発振器 5内の後段の移相回路 3 0 Cを 2段縦^接続するとともに、 その後段に位相反転回路 8 0を接続し、 この位相反 転回路 8 0の出力を^還抵抗 7 0を介して前段の移相回路 3 0 Cの人力側に帰還 させている。 FIG. 17 is a circuit diagram showing another configuration of the oscillator including the phase inversion IHJ path therein. In the oscillator 5D shown in FIG. 3, a phase shift circuit 30C at the subsequent stage in the oscillator 5 shown in FIG. 3 is vertically connected by two stages, and a phase inverting circuit 80 is connected to the subsequent stage. The output of the inversion circuit 80 is fed back to the human side of the preceding phase shift circuit 30 C via the feedback resistor 70.
位相反転回路 8 0によって 号が反転するため、 2つの移相回路 3 0 Cによる 位相シフ ト量の合計が 1 8 0 ° となるときに、 閉ループを一巡したときの位相シ フ ト量が 3 6 0 ° となり、 このときの帰還ループのループゲインを 1以上に設定 することにより所定の発振動作が行われる。 Since the signal is inverted by the phase inverting circuit 80, when the total phase shift amount of the two phase shift circuits 30 C is 180 °, the phase shift amount when making a round of the closed loop is 3 60 °, and a predetermined oscillation operation is performed by setting the loop gain of the feedback loop at this time to 1 or more.
したがって、 発振器 5 Dに含まれる 2つの移相回路 3 0 Cのいずれか -方を第 6冈 (B ) に示した移相冋路 1 3 0 Cに置き換えることにより、 コンデンサマイ クによって集音した音声を F M変調信号に用いた F M変調装置を構成することが できる。 あるいは、 発振器 5 Bに含まれる 2つの移相回路 3 0 Cのいずれか一方 を第 6図 (B ) に/下した移相回路 1 3 0 Cに置き換えるとともに、 他方を第 9図 に示した移相回路 3 0 Lに匿き換えて F M変調装置を構成してもよい。 Therefore, by replacing one of the two phase-shift circuits 30 C included in the oscillator 5 D with the phase-shift circuit 130 C shown in FIG. 6B, sound is collected by the condenser microphone. It is possible to configure an FM modulator using the converted voice for the FM modulation signal. Alternatively, one of the two phase shift circuits 30 C included in the oscillator 5 B is replaced with a phase shift circuit 130 C shown in FIG. 6B and the other is shown in FIG. The FM modulator may be configured by replacing the phase shift circuit with 30 L.
〔第 7の実施形態〕 (Seventh embodiment)
1 8図は、 F M変調装置の第 7の灾施形態の詳細構成を示す回路図である。 同図に示す F M変調装置 1 Bは、 所定の周波数において合計で 3 6 0 ° の位相 シフ トを行う 2つの移相回路 7 1 0 C、 6 3 0 Cと、 後段の移相问路 6 3 0 Cの 出力 の位相を変えずに所定の増幅度で増幅して出力する非反転回路 6 5 0と、 非反転回路 6 5 0の出力を前段の移相回路 7 1 0 Cの入力側に帰還させる抵抗 6 7 0とを含んで構成されている。 この抵抗 6 7 0は 0 Ωから有限の抵抗値を有し ている。 また、 抵抗 6 7 0と直列に接続されたキャパシ夕 6 7 2は直流電流を阻 止するためのものであり、 そのィンピーダンスは動作周波数において極めて小さ く、 すなわち大きな静電容量を有している。 FIG. 18 is a circuit diagram showing a detailed configuration of the seventh embodiment of the FM modulator. The FM modulator 1B shown in the figure includes two phase shift circuits 710C and 630C that perform a total of 360 ° phase shift at a predetermined frequency, and a phase shift circuit 6 in the subsequent stage. A non-inverting circuit 650 that amplifies and outputs the output of the 3 0 C at a predetermined amplification level without changing the phase, and the output of the non-inverting circuit 65 0 is the input side of the previous phase shift circuit 7 10 C And a resistor 670 that feeds back the current. This resistor 670 has a finite resistance value from 0 Ω. The capacitor 672 connected in series with the resistor 670 is for blocking DC current, and its impedance is extremely small at the operating frequency, that is, it has a large capacitance. I have.
この F M変調装置 1 Bは、 コンデンサマイクを含んで構成されており (詳細は 後述する) 、 このコンデンサマイクで柒音した音声を F M変調信号として用い、 発振出力として F M変調された信号を出力している。 The FM modulator 1B is configured to include a condenser microphone (details will be described later), and uses the sound obtained by the condenser microphone as an FM modulation signal, and outputs an FM-modulated signal as an oscillation output. ing.
例えば、 第 1 8図に示すように、 F M変調装置 1 Bの後段に増幅器 2およびァ ンテナ 3を接続し、 F M変調装置 6の出力を増幅器 2によって増幅してアンテナ
3から空中に送出すれば FMワイヤレスマイクとなる。 また、 アンテナ 3から空 中に送出する場合の他、 第 2図に したような送信ドライバ 4を介して伝送路 4 00に送出するようにしてもよい。 For example, as shown in Fig. 18, an amplifier 2 and an antenna 3 are connected after the FM modulator 1B, and the output of the FM modulator 6 is amplified by the amplifier 2 and the antenna If you send it out of the air from 3, it becomes an FM wireless microphone. In addition to transmitting the signal from the antenna 3 to the air, the signal may be transmitted to the transmission line 400 via the transmitting driver 4 as shown in FIG.
第 1の '施形態と同様に、 第 1 8図に示した FM変調装置 1 Bの詳細について 説明する前に、 その基本となる発振器の動作について説明する。 As in the case of the first embodiment, before describing the details of the FM modulator 1B shown in FIG. 18, the basic operation of the oscillator will be described.
第 1 9図は、 第 1 8図に示した FM変調装置 1 Bに含まれるコンデンサマイク 6 14 - 1およびキャパシ夕 6 14— 2を、 静電容量が固定のキャパシ夕 6 14 に置き換えた場合の 振器の構成を示す回路図である。 同^に示す発振器 5 Eは、 所定の周波数において台計で 360 ° の位相シフ トを行う 2つの移相回路 6 10 C、 630 Cと、 後段の移相回路 630 Cの出力^号の位相を変えずに所定の増 幅度で増幅して出力する非反転回路 650と、 非反転回路 650の出力を前段の 移相回路 6 1 0 Cの入力側に帰還させる抵抗 670とを含んで構成されている。 第 1 9図に示す発振器 5 Eを構成する前段の移相回路 6 1 0 Cは、 ゲ一卜が移 相回路 6 10 Cの入力端に接続された F E T 6 1 2と、 この FET 6 1 2のソ一 ス ' ドレイン間に直列に接続されたキャパシ夕 6 14および抵抗 6 1 6と、 FE T 6 1 2のドレインと正電源との間に接続された抵抗 6 18と、 FE T 6 1 2の ソースとアースとの間に接続された抵抗 620とを含んで構成されている。 なお、 移相回路 6 1 0 C内の抵抗 626は FET 6 1 2に適切なバイァス電圧を印加す るためのものである。 また、 FET 6 1 2および後述する FE T 632は、 少な くとも -方をバイポーラ トランジスタに置き換えるようにしてもよい。 Fig. 19 shows the case where the condenser microphone 614-1 and the capacitor 614-2 included in the FM modulator 1B shown in Fig. 18 are replaced with a fixed capacitance 614. FIG. 3 is a circuit diagram showing a configuration of the vibrator. The oscillator 5E shown in FIG. 3 has two phase shift circuits 6 10C and 630C that perform a phase shift of 360 ° by a total at a predetermined frequency, and the phase of the output signal of the subsequent phase shift circuit 630C. And a resistor 670 that feeds back the output of the non-inverting circuit 650 to the input side of the previous phase shift circuit 6100C. ing. The phase shift circuit 6100C in the preceding stage constituting the oscillator 5E shown in FIG. 19 is composed of a FET 612 connected to the input terminal of the phase shift circuit 610C, A capacitor 6 14 and a resistor 6 16 connected in series between the source 2 and the drain; a resistor 6 18 connected between the drain of the FET 6 12 and the positive power supply; and a FET 6 1 and a resistor 620 connected between the source and ground. The resistor 626 in the phase shift circuit 610 C is for applying an appropriate bias voltage to the FET 612. In addition, at least the FET 612 and the FET 632 described later may be replaced with a bipolar transistor.
ここで、 上述した F E T 6 1 2のソースおよびドレインに接続された 2つの抵 抗 620、 6 1 8の抵抗値はほぼ等しく設定されており、 ゲートに印加される入 力電圧の交流成分に着曰すると、 位相が一致した信号が FE T 6 1 2のソースか ら出力され、 位相が反転するとともにソースから出力される信号と振幅が等しい 信号が F E T 6 1 2のドレインから出力される。 このソースおよびドレインに れる交流電圧の振幅をともに Ei とする。 Here, the resistance values of the two resistors 620 and 618 connected to the source and the drain of the FET 612 described above are set to be substantially equal, and the resistance of the input voltage applied to the gate is reduced to the AC component. In other words, a signal with the same phase is output from the source of the FET 612, and a signal whose phase is inverted and whose amplitude is equal to the signal output from the source is output from the drain of the FET 612. The amplitude of the AC voltage applied to the source and drain is Ei.
この F E T 6 1 2のソース · ドレイン問にはキャパシ夕 6 14と抵抗 6 1 6と により構成される直列回路 (CR回路) が接続されており、 FET 6 1 2のソー スおよびドレインに現れる電圧のそれそれを抵抗 61 6あるいはキャパシ夕 6 1
4を介して合成した信号が移相回路 6 1 0 Cから出力される。 A series circuit (CR circuit) composed of a capacitor 614 and a resistor 616 is connected between the source and drain of the FET 612, and the voltage appearing at the source and drain of the FET 612 It's resistance 61 6 or capacity evening 6 1 The signal synthesized via 4 is output from the phase shift circuit 6100C.
ところで、 キャパシ夕 6 14の両端に現れる電圧 VC1と抵抗 6 1 6の両端に現 れる電圧 VR1とは互いに 90° 位相がずれており、 これらをべク トル的に合成し たものが F E T 6 1 2のソース ' ドレイン間の電圧 2 Ei に等しくなるため、 第 20図に示すように、 電圧 Ei の 2倍を斜辺とし、 キャパシ夕 6 14の両端電圧 VC1と抵抗 6 16の両端電圧 VR1とが :交する 2辺を構成する直角三角形を形成 することになる。 キャパシ夕 6 1 4と抵抗 6 1 6の接続点とグランドレベルとの 電位差を出力電圧 Eo として取り出すものとすると、 この出力電圧 Eo は第 20 図に示した半円においてその中心点を始点とし、 電圧 VC1と亀圧 VR1とが交差す る円周上の一点を終点とするべク トルで表すことができ、 出力信号の振幅は周波 数に関係なく -定であって、 位相シフ ト量は第 20図に示す 5 で表されること がわかる。 By the way, the voltage VC1 appearing at both ends of the capacitor 614 and the voltage VR1 appearing at both ends of the resistor 616 are 90 ° out of phase with each other. Since it is equal to the voltage 2 Ei between the source and the drain of 2, as shown in Fig. 20, twice the voltage Ei is the hypotenuse, and the voltage VC1 across the capacitor 614 and the voltage VR1 across the resistor 616 are : A right triangle that forms the two intersecting sides will be formed. Assuming that the potential difference between the connection point of the capacitor 6 14 and the resistor 6 16 and the ground level is taken out as the output voltage Eo, this output voltage Eo starts from the center point of the semicircle shown in FIG. It can be represented by a vector ending at a point on the circumference where the voltage VC1 and the tortuosity pressure VR1 intersect.The amplitude of the output signal is constant regardless of the frequency, and the amount of phase shift is It can be seen that it is represented by 5 shown in Fig. 20.
また、 第 1 9図に示す発振器 5 Eを構成する後段の移相回路 630 Cは、 ゲ一 卜が移相回路 630 Cの入力端に接続された F E T 632と、 この FET 632 のソース ' ドレイン問に直列に接続された抵抗 636およびキャパシ夕 634と、 FE T 632のドレインと正電源との間に接続された抵抗 638と、 FET 63 2のソースとアースとの間に接絞された抵抗 640とを含んで構成されている。 なお、 移相回路 630 C内の抵抗 646は F E T 632に適切なバイアス電圧を 印加するためのものであり、 移相回路 630 Cと 6 1 0 Cの間に挿入されたキヤ パシ夕 648は .流電流阻止 fflである。 Further, the phase shift circuit 630C of the subsequent stage constituting the oscillator 5E shown in FIG. 19 includes a FET 632 whose gate is connected to the input terminal of the phase shift circuit 630C, and a source ド レ イ ン drain of the FET 632. A resistor 636 and a capacitor 634 connected in series, a resistor 638 connected between the drain of the FET 632 and the positive power supply, and a resistor connected between the source of the FET 632 and ground. 640. The resistor 646 in the phase shift circuit 630 C is for applying an appropriate bias voltage to the FET 632, and the capacitor 648 inserted between the phase shift circuits 630 C and 6100 C is. The current is ffl.
この移相回路 630 Cは、 基本的な構成は前段の移相回路 6 10 Cと じであ り、 抵抗 636とキャパシ夕 634からなる CR回路の接続を前段の移相回路 6 10 C内のキャパシ夕 6 14と抵枋 6 1 6からなる CR回路の接続と反対にした 点が異なっている。 This phase shift circuit 630C has the same basic configuration as that of the previous phase shift circuit 610C, and connects the CR circuit composed of the resistor 636 and the capacitor 634 in the preceding phase shift circuit 610C. The difference is that it is opposite to the connection of the CR circuit consisting of the capacity 6 14 and the fan 6 6.
したがって、 抵抗 636の 端電圧を VR2、 キャパシ夕 634の両端電圧を V C2としてこれらの位相関係を考えると、 第 2 1図に示すように、 ¾圧 Ei の 2倍 を斜辺とし、 抵抗 636の |{¾端電圧 VR2とキャパシ夕 634の両端電圧 VC2とが 直交する 2辺を構成する直角三角形を形成することになる。 抵抗 636とキャパ シ夕 634の接続点とグランドレベルとの^位差を出力電圧 Eo として取り出す
ものとすると、 この出力電圧 E o は第 2 1図に示した半円においてその中心点を 始点とし、 電圧 VR2と電圧 VC2とが交差する円周上の一点を終点とするべク トル で表すことができ、 出力信号の振幅は周波数に関係なく一定であって、 位相シフ 卜量は第 2 1図に示す 06 で表されることがわかる。 Therefore, considering the phase relationship between the terminal voltage of the resistor 636 as VR2 and the voltage across the capacitor 634 as V C2, as shown in Fig. 21, as shown in Fig. 21, the oblique side is twice the overvoltage Ei, | {The terminal voltage VR2 and the voltage VC2 across the capacitor 634 form a right-angled triangle that forms two orthogonal sides. Extract the difference between the ground point and the connection point of the resistor 636 and the capacitor 634 as the output voltage Eo In this case, the output voltage E o is represented by a vector whose starting point is the center point of the semicircle shown in Fig. 21 and whose end point is a point on the circumference where voltage VR2 and voltage VC2 intersect. It can be seen that the amplitude of the output signal is constant irrespective of the frequency, and the amount of phase shift is represented by 06 shown in FIG.
このようにして、 2つの移相回路 6 1 0 C、 6 3 0 Cのそれそれにおいて位相 が所定量シフ トされ、 所定の周波数において 2つの移相回路 6 1 0 C、 6 3 0 C の全体により位相シフ ト量の合 3トが 3 6 0 ° となる信号が出力される。 In this way, the phase is shifted by a predetermined amount at each of the two phase shift circuits 61 0 C and 63 0 C, and the two phase shift circuits 61 0 C and 63 0 C are shifted at a predetermined frequency. A signal in which the sum of the phase shift amounts to 360 ° is output by the whole.
また、 笫 1 9図に示した非反転回路 6 5 0は、 ドレインと ΪΗ電源との問に抵抗 6 5 4が、 ソースとアースとの間に抵抗 6 5 6がそれぞれ接続された F E T 6 5 2と、 ベースが F E T 6 5 2のドレインに接続されているとともにコレクタが抵 抗 6 6 0を介してソースに接続されたトランジスタ 6 5 8と、 F E T 6 5 2に適 切なバイアス ¾圧を印加するための抵抗 6 6 2とを含んで構成されている。 なお、 第 1 9図に示した非反転回路 6 5 0の前段に設けられたキャパシ夕 6 6 4は、 後 段の移相回路 6 3 0 Cの出力から直流成分を取り除く ¾流電流阻止用であり、 交 流成分のみが非反転回路 6 5 0に人力される。 The non-inverting circuit 650 shown in Fig. 19 has a resistor 654 connected between the drain and the power supply, and a FET 65 connected between the source and ground. 2, transistor 658 with base connected to the drain of FET 652 and collector connected to the source via resistor 660, and appropriate bias voltage applied to FET 652 And a resistor 6 62. The capacitor 664 provided in the preceding stage of the non-inverting circuit 650 shown in FIG. 19 is used to remove the DC component from the output of the subsequent phase shifting circuit 630 C. And only the AC component is manually input to the non-inverting circuit 650.
F E T 6 5 2は、 ゲートに交流 β¾が入力されると、 逆相の^号をドレインか ら出力する。 また、 トランジスタ 6 5 8は、 ベースにこの逆相の^号が入力され ると、 さらに位相を反転した信号、 すなわち F E Τ 6 5 2のゲートに入力された 信号の位相を基準に考えると同相の^号をコレクタから出力し、 この同相の信 が非反転回路 6 5 0から出力される。 この非反転回路 6 5 0の出力は、 出力端子 9 2から発振器 5 Εの出力として取り出されるとともに、 抵抗 6 7 0を介して前 段の移相回路 6 1 0 Cの入力側に帰還されている。 When the AC β 2 is input to the gate, the F ET 652 outputs an opposite-phase ^ sign from the drain. Also, when the opposite phase ^ is input to the base of the transistor 658, the phase of the signal further inverted, that is, the phase of the signal input to the gate of the FEΤ652 becomes in-phase. Is output from the collector, and this in-phase signal is output from the non-inverting circuit 650. The output of the non-inverting circuit 650 is taken out from the output terminal 92 as the output of the oscillator 510, and is fed back to the input side of the preceding phase shift circuit 610C via the resistor 670. I have.
上述した非反転回路 6 5 0の増幅度は、 上述した抵抗 6 5 4、 6 5 6、 6 6 0 の各抵抗値によって決まり、 これら各抵抗の抵抗値を調整することにより、 第 1 9図に示した 2つの移相回路 6 1 0 C、 6 3 0 Cおよび抵抗 6 7 0を含んで形成 される帰還ループのループゲインを 1以 hに設定することができ、 一巡したとき に位相シフ ト量の合計がが 3 6 0 ° となるような周波数で正弦波発振が行われる c ところで、 第 1 8図に示した F M変調装匿 1 Bは、 J:述した発捩器 5 Eに含ま れる前段の移相回路 6 1 0 Cをコンデンサマイクを含む移相回路 7 1 0 Cに置き
換えた構成を^しており、 次にこのような構成を有する FM変調装置 1 Bについ て説明する。 The amplification of the above-described non-inverting circuit 650 is determined by the resistance values of the above-described resistors 654, 656, and 660. By adjusting the resistance values of these resistors, the amplification degree of FIG. The loop gain of the feedback loop formed by including the two phase shift circuits 61 0 C and 63 0 C and the resistor 67 0 shown in Fig. 10 can be set to 1 or more h. c total bet amount sinusoidal oscillation is performed at a frequency such that the 3 6 0 ° Incidentally, FM modulation So匿1 B shown in the first 8 figures, J: the predicate was Hatsusuji device 5 E The included phase shift circuit 610 C of the previous stage is placed in the phase shift circuit 710 C including the condenser microphone. Next, the FM modulator 1B having such a configuration will be described.
第 22図は、 コンデンサマイクを含む移相回路の構成を示す回路図であり、 同 図 (A) には FM変調装置 1 Bに含まれる前段の移相回路 7 10 Cの構成が示さ れている。 この移相回路 7 10 Cは、 第 1 9図に示した発振器 5 Eに含まれる前 段の移相回路 6 1 0 Cにおいて、 キャパシ夕 6 14と抵抗 6 16からなる CR回 路をコンデンサマイク 6 14— 1およびキャパシ夕 6 14- 2と抵抗 6 16から なる CR回路に置き換えた構成を有している。 FIG. 22 is a circuit diagram showing a configuration of a phase shift circuit including a condenser microphone. FIG. 22 (A) shows a configuration of a preceding phase shift circuit 710C included in the FM modulator 1B. I have. This phase shift circuit 710C is a capacitor microphone which is composed of a capacitor circuit 614 and a resistor 616 in the former stage phase shift circuit 610C included in the oscillator 5E shown in FIG. It has a configuration in which it is replaced with a CR circuit consisting of 614-1 and capacity 614-2 and resistor 616.
このように、 コンデンサマイク 6 14— 1を含んで CR回路を構成した場合に は、 この CR回路の時定数 T ( = CR) がコンデンサマイク 6 14— 1が有する 静 ¾容 に応じて微小変化する。 この静電容景の変化は、 コンデンサマイク 6 1 4一 1が抬ぅ音圧変化を反映しており、 音圧に応じて一方の移相问路 7 10 Cに 含まれる CR回路の時定数が変化することから発振出力の周波数も変化する。 す なわち、 コンデンサマイク 6 14— 1を CR回路の一部に使用することにより、 FM変調された信 ¾を容易に^ることができる。 Thus, when a CR circuit including the condenser microphone 614-1 is formed, the time constant T (= CR) of the CR circuit varies slightly according to the static capacity of the condenser microphone 614-1. I do. This change in the electrostatic landscape reflects the change in the rising sound pressure of the condenser microphone 6114, and the time constant of the CR circuit included in one of the phase shift circuits 7 10 C according to the sound pressure changes. Because of the change, the frequency of the oscillation output also changes. In other words, by using the condenser microphone 614-1 as a part of the CR circuit, FM-modulated signals can be easily obtained.
なお、 第 1 8図に示した FM変調装置 1 Bでは、 前段の移相回路にコンデンサ マイクを ませたが、 後段の移相回路にコンデンサマイクを含ませるようにして もよい。 すなわち、 第 19図に^した発振器 5 Eを構成する後段の移相回路 63 0 Cを、 ί¾22図 (Β) に示す移相回路 730 C (移相回路 630 C内のキャパ シタ 634の代わりにコンデンサマイク 634— 1とキャパシ夕 634— 2を川 いたもの) に置き換えてもよい。 In the FM modulator 1B shown in FIG. 18, a condenser microphone is not provided in the preceding phase shift circuit, but a condenser microphone may be included in the subsequent phase shift circuit. That is, the phase shift circuit 630 C of the subsequent stage constituting the oscillator 5 E shown in FIG. 19 is replaced with the phase shift circuit 730 C shown in FIG. 22 (Β) instead of the capacitor 634 in the phase shift circuit 630 C. The condenser microphone 634-1 and the capacity 634-2 may be replaced by rivers.
また、 上述した FM変調装置 1 Βは、 2つの移相回路をともに CR回路を含ん で構成したが、 コンデンサマイクを含まない一方の移相回路を LR回路を含む移 相回路に置き換えることもできる。 In the above-described FM modulator 1, the two phase shift circuits are both configured to include the CR circuit, but one of the phase shift circuits that does not include the condenser microphone can be replaced with a phase shift circuit that includes an LR circuit. .
例えば、 第 23図は、 第 1 9図に示した発&器 5 Εに含まれる前段の移相回路 6 10 Cと置き換え可能な移相回路 6 10 Lの構成を示す回路図である。 第 23 図に示す移相回路 6 1 0 Lは、 第 1 9図に示した移相回路 6 1 0 C内のキャパシ タ 6 14と抵抗 6 1 6からなる CR回路を、 抵抗 6 1 6とインダク夕 6 17から なる LR回路に置き換えた構成を有している。 抵抗 6 1 6の両端電圧を VR1、 ィ
ンダクタ 6 1 7の両端 ¾圧を VL1とすると、 第 2 4図に示すように、 電圧 E i の 2倍を斜辺とし、 抵抗 6 1 6の r 端 ^圧 VR1とインダク夕 6 1 7の両端電圧 VL1 とが直交する 2辺を構成する商角三角形を形成することになる。 抵抗 6 1 6とィ ンダク夕 6 1 7の接続点とグランドレベルとの電位差を出力電圧 E o として取り 出すものとすると、 この出力電圧 E o は第 2 4図に示した半円においてその中心 点を始点とし、 電圧 VR1と電圧 VL1とが交差する円周 I:の -点を終点とするべク トルで表すことができ、 出力信号の振幅は周波数に関係なく一定であって、 位相 シフ ト量は第 2 4図に示す 07 で表されることがわかる。 For example, FIG. 23 is a circuit diagram showing a configuration of a phase shift circuit 610 L that can be replaced with the preceding phase shift circuit 610 C included in the generator & generator 5 shown in FIG. The phase shift circuit 610 L shown in FIG. 23 is a CR circuit consisting of the capacitor 614 and the resistor 616 in the phase shift circuit 610 C shown in FIG. It has a configuration in which it is replaced with an LR circuit consisting of an inductor 617. Connect the voltage across resistor 6 16 to VR1, Assuming that the pressure at both ends of the inductor 6 17 is VL1, as shown in FIG. 24, the oblique side is twice the voltage E i, the r end of the resistor 6 16 ^ pressure VR1 and both ends of the inductor 6 17 The voltage VL1 forms a quotient triangle forming two sides that are orthogonal to each other. Assuming that the potential difference between the connection point between the resistor 6 16 and the inductor 6 17 and the ground level is taken out as the output voltage E o, this output voltage E o is the center of the half circle shown in Fig. 24. It can be expressed as a vector with the point as the starting point and the end point at the-point of the circumference I: where the voltage VR1 and the voltage VL1 intersect. The amplitude of the output signal is constant regardless of the frequency, and the phase shift It can be seen that the data amount is represented by 07 shown in FIG.
ところで、 第 1 9図に示した移相回路 6 1 0 C內の C R回路の時定数と第 2 3 図に示した移相回路 6 1 0 L内の L R回路の時定数をともに Tとすると、 これら の移相回路 6 1 0 C、 6 1 0 Lの伝達関数はともに a ( 1 - T s ) / ( 1 + T s ) となる。 ここで、 s = j wであり、 aは各移相回路の利得である。 By the way, assuming that the time constant of the CR circuit of the phase shift circuit 6 10 C 內 shown in FIG. 19 and the time constant of the LR circuit in the phase shift circuit 6 10 L shown in FIG. The transfer functions of these phase shift circuits 6110C and 610L are both a (1−Ts) / (1 + Ts). Here, s = jw, and a is the gain of each phase shift circuit.
このように、 移相回路 6 1 0 Lは移相回路 6 1 0 Cと等価であり、 移相回路 6 1 0 Cを移相回路 6 1 0 Lに置き換えることが": I能となる。 したがって、 第 1 9 図に示した発振器 5 Eにおいて、 前段の移相回路 6 1 0 Cを第 2 3図に示した移 相回路 6 1 0 Lに置き換えるとともに、 後段の移相冋路 6 3 0 Cをコンデンサマ ィクを含む第 2 2図 (B ) に示した移相回路 7 3 0 Cに置き換えることにより、 2つの移相回路のそれそれが L R |【,j路ぁるいは C R IiiJ路を含んだ F M変調装置を 構成することができる。 As described above, the phase shift circuit 610L is equivalent to the phase shift circuit 610C, and replacing the phase shift circuit 610C with the phase shift circuit 610L is an ": I function." Therefore, in the oscillator 5E shown in FIG. 19, the phase shift circuit 6100 of the preceding stage is replaced with the phase shift circuit 6100L shown in FIG. By replacing 0 C with the phase shift circuit 730 C shown in Fig. 22 (B) that includes a capacitor mark, each of the two phase shift circuits becomes LR | An FM modulator including the IiiJ path can be configured.
また、 第 2 5図は第 1 9図に示した発振器 5 Eに含まれる後段の移相回路 6 3 0 Cと置き換え可能な移相回路 6 3 0 Lの構成を示す回路図である。 第 2 5図に 示す移相回路 6 3 0 Lは、 第 1 9図に示した移相回路 6 3 0 C内の抵抗 6 3 6と キャパシタ 6 3 4からなる C R问路を、 インダク夕 6 3 7と抵抗 6 3 6からなる L R回路に置き換えた構成を有している。 インダク夕 6 3 7の両端電圧を VL2、 抵抗 6 3 6の両端電圧を VR2とすると、 第 2 6図に示すように、 電圧 E i の 2倍 を斜辺とし、 インダク夕 6 3 7の両端電圧 VL2と抵抗 6 3 6の両端電圧 VR2とが 直交する 2辺を構成する直角 角形を形成することになる。 ィンダクタ 6 3 7と 抵抗 6 3 6の接続点とグランドレベルとの電位差を出力電圧 E o として取り出す ものとすると、 この出力電/: H E o は第 2 6図に示した半円においてその中心点を
始点とし、 電圧 V L2と電圧 V R2とが交差する円周 hの一点を終点とするべク トル で表すことができ、 出力信号の振幅は周波数に関係なく -定であって、 位相シフ ト Mは第 2 6図に示す 0 8 で表されることがわかる。 FIG. 25 is a circuit diagram showing a configuration of a phase shift circuit 630 L which can be replaced with a subsequent phase shift circuit 630 C included in the oscillator 5 E shown in FIG. The phase shift circuit 63 0 L shown in Fig. 25 is connected to the CR circuit consisting of the resistor 6 36 and the capacitor 63 4 in the phase shift circuit 63 0 C shown in Fig. It has a configuration in which it is replaced with an LR circuit consisting of 3 7 and resistor 6 3 6. Assuming that the voltage between both ends of the inductor 6 3 7 is VL2 and the voltage between both ends of the resistor 6 3 6 is VR2, as shown in Fig. 26, twice the voltage E i is the hypotenuse, and the voltage across the inductor 6 3 7 is VL2 and the voltage VR2 across the resistor 636 form a right-angled rectangle that forms two orthogonal sides. Assuming that the potential difference between the connection point between the inductor 637 and the resistor 636 and the ground level is taken out as the output voltage Eo, this output voltage /: HEo is the center point of the semicircle shown in Fig. 26. To It can be represented by a vector with the start point as the end point of the circumference h where the voltage V L2 and the voltage V R2 intersect.The amplitude of the output signal is constant regardless of the frequency, and the phase shift It can be seen that M is represented by 08 shown in FIG.
ところで、 第 1 9図に示した移相回路 6 3 0 C内の C R 路の時定数と第 2 5 1に示した移相回路 6 3 0 L内の L R回路の時定数をともに Tとすると、 これら の移相冋路 6 3 0 C、 6 3 0 Lの伝達関数はともに一 a ( 1— T s ) / ( 1 + T s ) となる。 By the way, assuming that the time constant of the CR circuit in the phase shift circuit 63 0 C shown in FIG. 19 and the time constant of the LR circuit in the phase shift circuit 63 0 L shown in FIG. The transfer functions of these phase shift circuits 63 0 C and 63 0 L are both 1 a (1-T s) / (1 + T s).
このように、 移相回路 6 3 0 Lは移相回路 6 3 0 Cと等価であり、 移相回路 6 3 0 Cを移相回路 6 3 0 Lに置き換えることが可能となる。 したがって、 第 1 9 図に示した発振器 5 Eにおいて、 後段の移相冋路 6 3 0 Cを第 2 5図に示した移 相回路 6 3 0 Lに置き換えるとともに、 前段の移相回路 6 1 0 Cをコンデンサマ イクを む第 2 2図 (A ) に示した移相回路 7 1 0 Cに置き換えることにより、 2つの移ネ Π Ι"1路のそれそれが L R回路あるいは C R回路を含んだ F M変調装置を 構成することができる。 As described above, the phase shift circuit 630L is equivalent to the phase shift circuit 630C, and the phase shift circuit 630C can be replaced with the phase shift circuit 630L. Therefore, in the oscillator 5E shown in FIG. 19, the subsequent phase shift circuit 63 0 C is replaced with the phase shift circuit 63 0 L shown in FIG. By replacing 0 C with the phase shift circuit 710 C shown in Fig. 22 (A), which includes a capacitor microphone, each of the two shift circuits includes an LR circuit or a CR circuit. An FM modulator can be configured.
〔第 8の灾施形態〕 [Eighth Embodiment]
上述した第 7の実施形態で説明した各種の: F M変調装^においては、 2つの移 相 0路による位相シフ ト量の合計が 3 6 0 ° となる 波数の発振動作を行ってい るが、 閉ル一プ內に位相反転问路を接続することにより、 2つの移相回路による 位相シフ ト景の合計が 1 8 0 ° となる周波数で発振動作を行わせるようにしても よい。 In the various types of FM modulator described in the seventh embodiment described above, the oscillation operation is performed at a wave number where the total phase shift amount of the two phase-shifted zero paths is 360 °. By connecting a phase inversion circuit to the closed loop, the oscillation operation may be performed at a frequency at which the sum of the phase shift views by the two phase shift circuits is 180 °.
第 2 7図は、 2つの移相回路と位相反転冋路とを用いて構成した発振器の回路 冈である。 同図に示す発振器 5 Fは、 第 1 9図に した発振器 5 Ε内の前段の移 相回路 6 1 0 Cを 2段縦続接続するとともに、 その後段に F E T 6 8 2と抵抗 6 8 4および 6 8 6からなる位相反転回路 6 8 0を接続し、 この位相反転回路 6 8 0の出力を抵抗 6 7 0を介して前段の移相 [ J路 6 1 0 Cの人力側に帰還させてい る。 FIG. 27 is a circuit diagram of an oscillator configured using two phase shift circuits and a phase inversion circuit. The oscillator 5F shown in the figure is composed of a cascade connection of two stages of the phase shift circuit 610C at the preceding stage in the oscillator 5 1 shown in Fig. 19, and the FET 682 and the resistor 6884 and A phase inverting circuit 680 consisting of 686 is connected, and the output of the phase inverting circuit 680 is fed back to the human side of the previous stage via the resistor 670 through the phase shifter [J path 610 C]. You.
位相反転冋路 6 8 0によって信号が反転するため、 2つの移相回路 6 1 0 Cに よる位相シフ ト量の合計が 1 8 0 ° となるときに、 閉ループを一巡したときの位 相シフ ト量が 3 6 0 ° となり、 このときの帰還ループのループゲイン * 1以ヒに
設定することにより所定の允振動作が行われる。 Since the signal is inverted by the phase inversion circuit 680, when the total phase shift amount of the two phase shift circuits 610C is 180 °, the phase shift when the circuit goes through a closed loop is completed. In this case, the feedback amount becomes 360 °, and the loop gain of the feedback loop at this time * 1 By setting, a predetermined swing operation is performed.
したがって、 発振器 5 Fに含まれる 2つの移相回路 6 1 0 Cのいずれか一方を 第 2 2図 (A ) に示した移相回路 7 1 0 Cに置き換えることにより、 コンデンサ マイクによって集音した音^を F M変調信号に用いた F M変調装置を構成するこ とができる。 あるいは、 発振器 5 Fに含まれる 2つの移相回路 6 1 0 Cのいずれ か一方を第 2 2図 (A ) に示した移相回路 7 1 0 Cに置き換えるとともに、 他方 を第 2 3図に示した移相冋路 6 1 0 Lに置き換えて F M変調装置を構成してもよ い。 Therefore, by replacing one of the two phase shift circuits 6 10 C included in the oscillator 5 F with the phase shift circuit 7 10 C shown in FIG. 22 (A), the sound was collected by the condenser microphone. It is possible to configure an FM modulator using sound ^ for an FM modulation signal. Alternatively, one of the two phase shift circuits 6 10 C included in the oscillator 5 F is replaced with the phase shift circuit 7 10 C shown in FIG. 22 (A), and the other is replaced with the phase shift circuit 7 10 C shown in FIG. The FM modulator may be configured in place of the phase shift path 6101L shown.
〔第 9の実施形態〕 (Ninth embodiment)
2 8図は、 2つの移相回路と位相反転回路とを縦続接続して構成した他の発 振器の冋路図である。 同図に示す発振器 5 Gは、 第 1 9図に示した発振器 5 E内 の後段の移相回路 6 3 0 Cを 2段縦続接続するとともに、 その後段に位相反転回 路 6 8 0を接続し、 この位相反転回路 6 8 0の出力を抵抗 6 7 0を介して前段の 移相回路 6 3 0 Cの入力側に帰還させている。 FIG. 28 is a circuit diagram of another oscillator configured by cascade-connecting two phase shift circuits and a phase inversion circuit. The oscillator 5G shown in the figure has a two-stage cascade connection of the subsequent phase shift circuit 63 0 C in the oscillator 5 E shown in FIG. 19, and a phase inversion circuit 680 connected to the subsequent stage. Then, the output of the phase inversion circuit 680 is fed back to the input side of the preceding phase shift circuit 630 C via the resistor 670.
位相乂転回路 6 8 0によって信号が反転するため、 2つの移相回路 6 3◦ Cに よる位相シフ ト量の合計が 1 8 0 ° となるときに、 閉ループを一巡したときの位 相シフ ト量が 3 6 0。 となり、 このときの帰還ループのループゲインを 1以卜.に 設定することにより所定の発振動作が行われる。 Since the signal is inverted by the phase shift circuit 680, when the total phase shift amount of the two phase shift circuits 63 ° C reaches 180 °, the phase shift when the circuit goes through a closed loop is completed. 3600. A predetermined oscillation operation is performed by setting the loop gain of the feedback loop at this time to 1 or less.
したがって、 発振器 5 Gに含まれる 2つの移相回路 6 3 0 Cのいずれか一方を 第 2 2図 (B ) に した移相回路 7 3 0 Cに置き換えることにより、 コンデンサ マイクによって集音した音声を F M変調信号に用いた F M変調装置を構成するこ とができる。 あるいは、 発振器 5 Gに含まれる 2つの移相回路 6 3 0 Cのいずれ か一方を第 2 2図 ( B ) に した移相回路 7 3 0 Cに ¾き換えるとともに、 他方 を第 2 5図に示した移相回路 6 3 0 Lに置き換えて F M変調装置を構成してもよ い。 Therefore, by replacing one of the two phase-shift circuits 630C included in the oscillator 5G with the phase-shift circuit 730C shown in FIG. 22 (B), the sound collected by the condenser microphone can be obtained. This makes it possible to configure an FM modulator using the FM modulation signal. Alternatively, one of the two phase shift circuits 630C included in the oscillator 5G is replaced with a phase shift circuit 730C shown in FIG. 22 (B), and the other is replaced with the phase shift circuit shown in FIG. The FM modulator may be configured by replacing the phase shift circuit 630 L shown in FIG.
i 1 0の実施形態〕 Embodiment of i10)
第 2 9図は、 F M変調装置の第 1 0の実施形態の詳細構成を示す回路図である c 同図に^す F M変調装置 1 Cは、 入力される交流 ig号の位相を変えずに出力する 非反転回路 8 5 0と、 所定の周波数において合計で 3 6 0 ° の位相シフ トを行う
2つの移相冋路 9 1 0 C、 8 3 0 Cと、 帰還抵抗 8 7 0とを含んで構成されてい る。 FIG. 29 is a circuit diagram showing the detailed configuration of the tenth embodiment of the FM modulator c . The FM modulator 1C shown in FIG. 29 does not change the phase of the input AC signal. Performs a total of 360 ° phase shift at the specified frequency with the non-inverting circuit 850 output It is configured to include two phase shift circuits 910C and 830C, and a feedback resistor 870.
非反転回路 8 5 0は、 バッファ回路として機能するものであり、 例えばェミツ 夕ホロワ回路やソースホロワ回路等により構成されている。 なお、 直接接続した 場合の損失等を最小限に抑えるように帰還抵抗 8 7 0等の各素子の素子定数を選 定した場合には、 この非反転回路 8 5 0を省略して F M変調装置 1 Cを構成して もよい。 The non-inverting circuit 850 functions as a buffer circuit, and includes, for example, an emitter follower circuit, a source follower circuit, and the like. When the element constant of each element such as the feedback resistor 870 is selected so as to minimize the loss and the like when directly connected, the non-inverting circuit 850 is omitted and the FM modulator is omitted. 1 C may be configured.
第 2 9 に示す F M変調装置 1 Cは、 コンデンサマイクを含んで構成されてお り (詳細は後述する) 、 このコンデンサマイクで集音した音声を F M変調して出 力する。 The FM modulator 1C shown in the 29th is configured to include a condenser microphone (details will be described later), and FM-modulates the sound collected by the condenser microphone and outputs it.
例えば、 ¾ 2 9図に小すように、 F M変調装置 1 Cの後段に増幅器 2およびァ ンテナ 3を接続し、 F M変調装置 1の出力を増幅器 2によって増幅してアンテナ 3から空中に送出すれば F Mワイヤレスマイクとなる。 また、 アンテナ 3から空 中に送出する場合の他、 第 2図に示すように送信ドライバ 4を介して伝送路 4 0 0に送出するようにしてもよい。 For example, as shown in Fig. 29, amplifier 2 and antenna 3 are connected after FM modulator 1C, and the output of FM modulator 1 is amplified by amplifier 2 and sent out from antenna 3 to the air. It would be an FM wireless microphone. In addition to transmitting the signal from the antenna 3 to the air, the signal may be transmitted to the transmission line 400 via the transmitting driver 4 as shown in FIG.
2 9図に示した F M変調装置 1 Cの詳細について説明する前に、 その基本と なる発 器の動作について説明する。 Before describing the details of the FM modulator 1C shown in FIG. 29, the basic operation of the generator will be described.
第 3 0図は、 第 2 9図に示した F M変調装 i¾: 1 Cに含まれるコンデンサマイク 8 1 4 - 1およびキャパシ夕 8 1 4— 2を静電容 Sが固定のキャパシ夕 8 1 4に ^き換えて簡略化した発振器 5 Hの構成を示す回路図である。 FIG. 30 shows the condenser microphone 8 14-1 and the capacity 8 1 4-2 included in the FM modulator i¾: 1 C shown in FIG. FIG. 3 is a circuit diagram showing a configuration of an oscillator 5H simplified in place of FIG.
同図に示す前段の移相回路 8 1 0 Cは、 2入力の差分電圧を所定の増幅度で増 幅して出力する差動増幅器 8 1 2と、 人力された交流信号の位相を所定 IIシフ 卜 させて差動増幅器 8 1 2の非反転入力端子に入力するキャパシ夕 8 1 4および抵 抗 8 1 6と、 入力された交流信号の位相を変えずにその電圧レベルを約 1 Z 2に 分 ίΐして差動増幅器 8 1 2の反転入力端子に入力する抵抗 8 1 8および 8 2 0と を含んで構成されている。 The phase shift circuit 810C at the front stage shown in the figure includes a differential amplifier 812 that amplifies the differential voltage of the two inputs with a predetermined amplification and outputs the amplified signal, and a phase II that converts the phase of the input AC signal to a predetermined II. The capacitor 814 and the resistor 816 which are shifted to the non-inverting input terminal of the differential amplifier 812 and the voltage level of about 1 Z2 without changing the phase of the input AC signal. And a resistor 818 and 820 which are input to the inverting input terminal of the differential amplifier 812.
第 3 1図は、 第 3 0図に^す移相问路 8 1 0 Cの入出力電圧とキャパシ夕等に 現れる電圧との関係を示すべク トル図である。 FIG. 31 is a vector diagram showing the relationship between the input / output voltage of the phase shift circuit 8100 C shown in FIG. 30 and the voltage appearing in the capacity and the like.
同図に示すように、 抵抗 8 1 6の両端に現れる電圧 VR1とキャパシ夕 8 1 4の
両端に現れる電圧 VC1は互いに位相が 90 ° ずれており、 これらをベク トル的に 加算したものが移相回路 8 1 0 Cの入力電圧 Ei に相当する。 したがって、 入力 電圧 Ei の振幅が一定で周波数のみが変化した場合には、 第 3 1図に示す半円の 円周に沿って抵抗 8 1 6の両端電圧 VR1とキャパシ夕 8 14の両端電圧 VC1とが 変化する。 As shown in the figure, the voltage VR1 appearing across the resistor 8 16 and the voltage The voltage VC1 appearing at both ends is 90 ° out of phase with each other, and the vector sum of these is equivalent to the input voltage Ei of the phase shift circuit 8100C. Accordingly, when the amplitude of the input voltage Ei is constant and only the frequency changes, the voltage VR1 across the resistor 8 16 and the voltage VC1 across the capacitor 8 14 along the circumference of the semicircle shown in FIG. Changes.
また、 差動増幅器 8 1 2の非反転入力端子に印加される電圧 (キャパシ夕 8 1 4の両端電圧 VC1) から反転入力端子に印加される電圧 (抵抗 820の両端電圧 Ei /2) をべク トル的に減算したものが差分電圧 Eo ' となる。 この差分電圧 Eo ' は、 第 3 1図に示した半円において、 その中心点を始点とし、 電圧 VC1と 電圧 VR1とが交差する円周上の一点を終点とするべク トルで表すことができ、 そ の大きさは半円の半径 Ei /2に等しくなる。 In addition, the voltage applied to the non-inverting input terminal of the differential amplifier 812 (the voltage VC1 across the capacitor 814) to the voltage applied to the inverting input terminal (the voltage Ei / 2 across the resistor 820) is calculated. The difference obtained by the torque is the difference voltage Eo '. This differential voltage Eo 'can be represented by a vector whose center point is the starting point and whose end point is a point on the circumference where voltage VC1 and voltage VR1 intersect in the semicircle shown in Fig. 31. And its size is equal to the radius of the semicircle Ei / 2.
差動増幅器 8 1 2の出力電圧 Eo はこの差分 ¾圧 Eo ' を所定の増幅度で増幅 したものとなる。 したがって、 上述した移相回路 8 1 0 Cは、 出力電圧 Eo が入 力電圧 Ei の周波数によらず一定であって、 全域通過回路として動作する。 また、 第 3 1図から明らかなように、 電圧 VC1と電圧 VR1とは円周上で直角に 交わるため、 入カ¾圧 Ei と電圧 VC1との位相差は、 周波数 ωが 0から∞まで変 化するに従って、 人力電圧 Ei を基準として時計回り方向 (位相遅れ方向) に◦ ° から 90。 まで変化する。 そして、 移相回路 8 10 C全体の位相シフ 卜量 09 は、 周波数に応じて 0° から 1 80° まで変化する。 The output voltage Eo of the differential amplifier 812 is obtained by amplifying the differential voltage Eo ′ with a predetermined amplification factor. Therefore, the above-described phase shift circuit 8100C operates as an all-pass circuit, in which the output voltage Eo is constant regardless of the frequency of the input voltage Ei. Further, as is clear from FIG. 31, since the voltage VC1 and the voltage VR1 intersect at right angles on the circumference, the phase difference between the input pressure Ei and the voltage VC1 varies from a frequency ω of 0 to ∞. From 90 ° in the clockwise direction (phase lag direction) based on the human-power voltage Ei. To change. Then, the phase shift amount 09 of the entire phase shift circuit 8 10 C changes from 0 ° to 180 ° according to the frequency.
同様に、 第 30図に示す後段の移相回路 830 Cは、 2入力の差分電圧を所定 の増幅度で増幅して出力する差動増幅器 832と、 入力された交流信号の位相を 所定量シフ 卜させて差動増幅器 832の非反転入力端子に入力するキャパシ夕 8 34および抵抗 836と、 入力された交流信号の位相を変えずにその電圧レベル を約 1/2に分圧して差動増幅器 8 1 2の反転入力端子に入力する抵抗 838お よび 840とを含んで構成されている。 Similarly, the subsequent phase shift circuit 830C shown in FIG. 30 includes a differential amplifier 832 that amplifies the differential voltage of the two inputs with a predetermined amplification and outputs the amplified voltage, and shifts the phase of the input AC signal by a predetermined amount. And a capacitor 834 and a resistor 836 that are input to the non-inverting input terminal of the differential amplifier 832, and the voltage level is divided into about 1/2 without changing the phase of the input AC signal, and the differential amplifier is divided. It is configured to include resistors 838 and 840 input to the inverting input terminal of 812.
第 32図は、 第 30図に示す移相回路 830 Cの入出力電圧とキャパシ夕等に 現れる電圧との関係を示すべク トル図である。 FIG. 32 is a vector diagram showing the relationship between the input / output voltage of the phase shift circuit 830C shown in FIG. 30 and the voltage appearing in the capacity and the like.
同図に示すように、 キャパシ夕 834の両端に現れる電圧 VC2と抵抗 836の 両端に現れる電圧 VR2は、 Ώいに位相が 90° ずれており、 これらをベク トル的
に加算したものが入力電圧 E i となる。 したがって、 入力信号の振幅が一定で周 波数のみが変化した場合には、 第 3 2 に示す半円の円周に沿ってキャパシタ 8 3 4の両端電圧 VC2と抵抗 8 3 6の両端電圧 VR2とが変化する。 As shown in the figure, the voltage VC2 appearing at both ends of the capacitor 834 and the voltage VR2 appearing at both ends of the resistor 836 are out of phase with each other by 90 °. Is the input voltage E i. Therefore, when the amplitude of the input signal is constant and only the frequency changes, the voltage VC2 across the capacitor 834 and the voltage VR2 across the resistor 836 along the circumference of the semicircle shown in FIG. Changes.
また、 差動増幅器 8 3 2の非反転入力端子に印加される電圧 (抵抗 8 3 6の両 端電圧 VR2) から反転入力端子に印加される電圧 (抵抗 8 4 0の両端電圧 E i / 2 ) をベク トル的に減算したものが差分電圧 E o ' となる。 この差分電圧 E o ' は、 第 3 2図に示した半円において、 その中心点を始点とし、 電圧 VR2と電圧 V C2とが交差する円周上の一点を終点とするべク トルで表すことができ、 その大き さは半円の半径 E i / 2に等しくなる。 Also, the voltage applied to the non-inverting input terminal of the differential amplifier 832 (the voltage VR2 across the resistor 836) and the voltage applied to the inverting input terminal (the voltage E i / 2 ) Is the difference voltage E o '. This difference voltage E o 'is represented by a vector whose center point is the start point and whose end point is a point on the circumference where voltage VR2 and voltage VC2 intersect in the semicircle shown in Fig. 32. Whose size is equal to the radius of the semicircle E i / 2.
差動増幅器 8 3 2の出力電圧 E o はこの差分電圧 E o ' を所定の増幅度で増幅 したものとなる。 したがって、 上述した移相回路 8 3 0 Cは、 出力電圧 E o が入 力信号の周波数によらず一定であって、 全域通過回路として動作する。 The output voltage E o of the differential amplifier 832 is obtained by amplifying the difference voltage E o 'with a predetermined amplification factor. Therefore, the above-described phase shift circuit 830C operates as an all-pass circuit, since the output voltage E o is constant regardless of the frequency of the input signal.
また、 第 3 2図から明らかなように、 電 H: VR2と電圧 VC2とは円周上で直角に 交わるため、 入力電圧 E i と電圧 VR2との位相差は、 周波数 ωが 0から∞まで変 化するに従って 1 8 0。 から 2 7 0 ° まで変化する。 そして、 移相回路 8 3 0 C 全体の位相シフ 卜量 0 10は周波数に応じて 1 8 0 ° から 3 6 0 ° まで変化する。 このようにして、 2つの移相回路 8 1 0 C、 8 3 0 Cのそれそれにおいて位相 が所定量シフ トされ、 所定の周波数において 2つの移相回路 8 1 0 C、 8 3 0 C の全体により位相シフ ト- Mの合計が 3 6 0 ° となる信号が出力される。 Further, as is clear from FIG. 32, since the voltage H: VR2 and the voltage VC2 intersect at right angles on the circumference, the phase difference between the input voltage Ei and the voltage VR2 is 180 as it changes. To 270 °. Then, the phase shift amount 010 of the entire phase shift circuit 830C changes from 180 ° to 360 ° according to the frequency. In this way, the phase is shifted by a predetermined amount at each of the two phase shift circuits 8100C and 8300C, and the two phase shift circuits 8100C and 8300C are shifted at a predetermined frequency. A signal is output in which the sum of the phase shifts -M is 360 °.
ところで、 第 2 9図に示した F M変調装置 1 Cは、 第 3 0図に示す発振器 5 H に含まれる前段の移相回路 8 1 0 Cを、 コンデンサマイクを含む移相回路 9 1 0 Cに置き換えた構成を有しており、 次にこのような構成を有する F M変調装置 6 について説明する。 By the way, the FM modulator 1 C shown in FIG. 29 includes a phase shift circuit 8 10 C in the preceding stage included in the oscillator 5 H shown in FIG. Next, the FM modulator 6 having such a configuration will be described.
第 3 3図は、 コンデンサマイクを含む移相回路の構成を示す冋路図であり、 同 図 (A ) には F M変調装置 1 Cに含まれる前段の移相回路 9 1 0 Cの構成が示さ れている。 この移相回路 9 1 0 Cは、 第 3 0図に示した発振器 5 Hに含まれる前 段の移相回路 8 1 0 Cにおいて、 キャパシ夕 8 1 4と抵抗 8 1 6からなる C R回 路をコンデンサマイク 8 1 4— 1およびキャパシタ 8 1 4— 2と抵抗 8 1 6から なる C R回路に置き換えた構成を有している。
このように、 コンデンサマイク 8 14— 1を含んで CR回路を構成した場合に は、 この CR冋路の時定数 T ( = CR) は、 コンデンサマイク 8 14— 1が有す る静電容 ¾に応じて微小変化する。 この静電容量の変化は、 コンデンサマイク 8 14— 1が拾う音圧変化を反映しており、 音圧に応じて 方の移相回路 9 1 0 C に含まれる CR回路の時定数が変化することから発振出力の周波数も変化する。 すなわち、 コンデンサマイク 8 14— 1を CR回路の --部に使用することにより、 簡単に FM変調された β号を得ることができる。 FIG. 33 is a circuit diagram showing a configuration of a phase shift circuit including a condenser microphone. FIG. 33A shows a configuration of a phase shift circuit 910C of a preceding stage included in the FM modulator 1C. It is shown. This phase shift circuit 910C is a CR circuit comprising a capacitor 814 and a resistor 816 in the previous phase shift circuit 8100C included in the oscillator 5H shown in FIG. Is replaced by a CR circuit consisting of a condenser microphone 8 14-1 and a capacitor 8 14-2 and a resistor 8 16. In this way, when a CR circuit is configured to include the condenser microphone 814-1, the time constant T (= CR) of this CR circuit is equal to the capacitance of the condenser microphone 814-1. It changes minutely in response. This change in capacitance reflects the change in sound pressure picked up by the condenser microphone 814-1, and the time constant of the CR circuit included in the phase shift circuit 910C changes according to the sound pressure. Therefore, the frequency of the oscillation output also changes. That is, by using the condenser microphone 814-1 in the-part of the CR circuit, it is possible to easily obtain the FM-modulated β signal.
なお、 第 29図に示した FM変調装置 1 Cでは、 前段の移相回路にコンデンサ マイクを^ませたが、 後段の移相冋路にコンデンサマイクを含ませるようにして もよい。 すなわち、 第 30図に示した発振器 5 Ηを構成する後段の移相回路 83 0 Cを、 第 3311 (Β) に示す移相回路 930 C (移相问路 830 C内のキャパ シ夕 834の代わりにコンデンサマイク 834— 1とキャパシ夕 834— 2を用 いたもの) に ^き換えるようにしてもよい。 In the FM modulator 1C shown in FIG. 29, a condenser microphone is provided in the preceding phase shift circuit, but a condenser microphone may be included in the subsequent phase shift circuit. That is, the subsequent phase shift circuit 830 C constituting the oscillator 5 shown in FIG. 30 is connected to the phase shift circuit 930 C shown in FIG. 3311 (Β) (the capacity of the capacitor 834 in the phase shift circuit 830 C). Alternatively, a condenser microphone 834-1 and a capacitor 834-2 may be used.
また、 上述した FM変調装置 1 Cは、 2つの移相回路をともに CR回路を含ん で構成したが、 コンデンサマイクを含まない移相回路を LR [口 I路を含む移相回路 に置き換えることもできる。 Although the above-mentioned FM modulator 1C has the two phase shift circuits both including the CR circuit, it is also possible to replace the phase shift circuit not including the condenser microphone with the LR [phase shift circuit including the port I path]. it can.
例えば、 第 34図は、 第 30図に示した発振器 5 Ηに含まれる前段の移相回路 8 1 0 Cと ^き換え可能な移相回路 8 1 0 Lの構成を示す回路図である。 第 34 図に示す移相回路 8 1 0 Lは、 第 30図に示した移相回路 8 10 Cに対して、 キ ャパシタ 8 14と抵抗 8 1 6からなる CR回路を、 抵抗 8 1 6とインダク夕 8 1 7からなる L R回路に置き換えた構成を している。 For example, FIG. 34 is a circuit diagram showing a configuration of a phase shift circuit 8110L replaceable with the preceding phase shift circuit 8100C included in the oscillator 5 shown in FIG. The phase shift circuit 810 L shown in FIG. 34 is different from the phase shift circuit 810 C shown in FIG. 30 in that a CR circuit consisting of a capacitor 814 and a resistor 8 16 The configuration is such that it is replaced with an LR circuit consisting of an inductor 8 17.
ところで、 第 30図に示した移相回路 8 10 C内の CR回路の時定数と第 34 図に示した移相回路 8 10 L内の LR回路の時定数をともに Τとすると、 これら の移相回路 8 10 C、 8 1 0 Lの伝達関数はともに a ( 1— T s) / ( 1 +T s) となる。 ここで、 s = j wであり、 aは各移相回路の利得である。 By the way, assuming that the time constant of the CR circuit in the phase shift circuit 810C shown in FIG. 30 and the time constant of the LR circuit in the phase shift circuit 810L shown in FIG. The transfer functions of the phase circuits 810C and 810L are both a (1—Ts) / (1 + Ts). Here, s = jw, and a is the gain of each phase shift circuit.
このように、 移相问路 8 1 0 Lは移相 路 8 1 0 Cと等価であり、 移相回路 8 10 Cを移相回路 8 1 0 Lに置き換えることが可能となる。 したがって、 第 30 図に示した発振器 5 Hにおいて、 前段の移相回路 8 1 0 Cを第 34図に示した移 相回路 8 1 0 Lに置き換えるとともに、 後段の移相回路 830 Cをコンデンサマ
イクを含む第 3 3図 (B ) に示した移相回路 9 3 0 Cに置き換えることにより、 2つの移相回路のそれそれが L R回路あるいは C R回路を含んだ F M変調装置を 構成することができる。 As described above, the phase shift circuit 8101L is equivalent to the phase shift circuit 8110C, and the phase shift circuit 810C can be replaced with the phase shift circuit 8101L. Therefore, in the oscillator 5H shown in FIG. 30, the phase shift circuit 810C in the preceding stage is replaced with the phase shift circuit 810C shown in FIG. By replacing the phase shift circuit 930C shown in Fig. 33 (B), which includes the phase shifter, each of the two phase shift circuits can constitute an FM modulator including an LR circuit or a CR circuit. it can.
また、 第 3 5図は第 3 0図に示した発振器 5 Hに含まれる後段の移相回路 8 3 0 Cと置き換え 能な移相回路 8 3 0 Lの構成を小す回路図である。 第 3 5図に 示す移相回路 8 3 0 Lは、 第 3 0図に示した移相回路 8 3 0 Cに対して、 抵抗 8 3 6とキャパシ夕 8 3 4からなる C R回路をィンダク夕 8 3 7と抵抗 8 3 6から なる L R回路に置き換えた構成を有している。 FIG. 35 is a circuit diagram for reducing the configuration of the phase shift circuit 830 L which can be replaced with the subsequent phase shift circuit 830 C included in the oscillator 5 H shown in FIG. The phase shift circuit 830 L shown in FIG. 35 is different from the phase shift circuit 830 C shown in FIG. 30 in that a CR circuit consisting of a resistor 836 and a capacitor 834 is connected to the inductor circuit. It has a configuration in which it is replaced by an LR circuit consisting of 837 and a resistor 836.
ところで、 第 3 0図に示した移相回路 8 3 0 C内の C R回路の時定数と第 3 5 図に示した移相冋路 8 3 0 L内の L R回路の時定数をともに Tとすると、 これら の移相回路 8 3 0 C、 8 3 0 Lの伝達関数はともに— a ( 1 - T s ) / ( 1 + T s ) となる。 By the way, the time constant of the CR circuit in the phase shift circuit 830 C shown in FIG. 30 and the time constant of the LR circuit in the phase shift circuit 830 L shown in FIG. Then, the transfer functions of these phase shift circuits 830C and 830L are both -a (1-Ts) / (1 + Ts).
このように、 移相回路 8 3 0 Lは移相回路 8 3 0 Cと等価であり、 移相回路 8 3 0 Cを移相回路 8 3 0 Lに置き換えることが^能となる。 したがって、 第 3 0 図に示した発振器 5 Hにおいて、 後段の移相回路 8 3 0 Cを第 3 5図に示した移 相回路 8 3 0 Lに置き換えるとともに、 前段の移相问路 8 1 0 Cをコンデンサマ イクを含む第 3 3図 ( A ) に^した移相回路 9 1 0 Cに置き換えることにより、 L R回路を含む移相回路と C R回路を含む移相回路とを縦絞接続した F M変調装 置を構成することができる。 Thus, the phase shift circuit 830 L is equivalent to the phase shift circuit 830 C, and it is possible to replace the phase shift circuit 830 C with the phase shift circuit 830 L. Therefore, in the oscillator 5H shown in FIG. 30, the subsequent phase shift circuit 830C is replaced with the phase shift circuit 830L shown in FIG. By replacing 0 C with the phase shift circuit 910 C shown in Fig. 33 (A) including a capacitor microphone, the phase shift circuit including the LR circuit and the phase shift circuit including the CR circuit are connected vertically. This makes it possible to construct a modified FM modulation device.
〔第 1 1の実施形態〕 [Eleventh embodiment]
上述した第 2 9図に示した F M変調装置 1 Cにおいては、 2つの移相回路によ る位相シフ ト鲎の合計が 3 6 0 ° となる周波数の発振動作を行っているが、 閉ル —プ内に位相反転回路を接続することにより、 2つの移相冋路による位相シフ ト 量の合計が 1 8 0 ° となる周波数で発振動作を行わせるようにしてもよい。 In the FM modulator 1C shown in FIG. 29 described above, the oscillation operation is performed at a frequency at which the sum of the phase shifts by the two phase shift circuits becomes 360 °. By connecting a phase inversion circuit in the loop, the oscillation operation may be performed at a frequency at which the sum of the phase shift amounts of the two phase shift circuits is 180 °.
第 3 6図は、 2つの移相回路と位相反転回路とを用いて構成した発振器の構成 を示す回路図である。 同図に示す発振器 5 Jは、 入力される交流信号の位相を反 転して出力する位相反転回路 8 8 0と、 所定の周波数において合計で 1 8 0。 の 位相シフ トを行う 2つの移相回路 8 1 0 Cと、 帰還抵抗 8 7 0とを含んで構成さ れている。
2つの移相回路 8 1 0 Cの入出力信号の位相関係は第 3 1図を用いて説明した 通りであり、 所定の周波数において、 2つの移相回路 8 1 0 Cの全体による位相 シフ 卜 fiの合計が 1 8 0 ° となる。 FIG. 36 is a circuit diagram showing a configuration of an oscillator configured using two phase shift circuits and a phase inversion circuit. The oscillator 5J shown in the figure has a phase inverting circuit 880 for inverting the phase of an input AC signal and outputting the inverted signal, and a total of 180 at a predetermined frequency. It is configured to include two phase shift circuits 8100 C for performing the phase shift of the above and a feedback resistor 870. The phase relationship between the input and output signals of the two phase shift circuits 8100C is as described with reference to FIG. 31. At a predetermined frequency, the phase shift by the entire two phase shift circuits 8100C is performed. The sum of fi is 180 °.
また、 2つの移相回路 8 1 0 Cの前段に接続された位相反転回路 8 8 0は、 入 力される交流信号の位相を反転するものであり、 例えば、 ェミッタ接地回路ゃソ —ス接地回路あるいはオペアンプと抵抗を組み合わせた回路によって実現される c このように、 位相反転回路 8 8 0によって信号の位相が反転するため、 2つの 移相回路 8 1 0 Cによる位相シフ ト量の合 十が 1 8 0 ° となるときに、 閉ループ を一巡したときの位相シフ ト量が 3 6 0 ° となり、 このときの帰還ループのルー プゲインを 1以上に設定することにより所定の発振動作が行われる。 Further, a phase inversion circuit 880 connected in front of the two phase shift circuits 810C inverts the phase of the input AC signal. For example, an emitter ground circuit and a source ground c Thus realized by a circuit that combines a circuit or an operational amplifier resistor, the phase inversion circuit 8 8 0 for the phase of the signal is inverted by the phase shift amount of the interleaf ten by two phase shifting circuits 8 1 0 C Is 180 °, the phase shift amount when the circuit goes through the closed loop becomes 360 °, and the specified oscillation operation is performed by setting the loop gain of the feedback loop to 1 or more at this time. .
したがって、 発振器 5 Jに含まれる 2つの移相回路 8 1 0 Cのいずれか一方を 第 3 3 fel ( A ) に示した移相回路 9 1 0 Cに i き換えることにより、 コンデンサ マイクによって集音した音声を F M変調信 に用いた F M変調装置を構成するこ とができる。 あるいは、 発振器 5 Jに含まれる 2つの移相回路 8 1 0 Cのいずれ か一方を第 3 3図 (A ) に示した移相回路 9 1 0 Cに [Hき換えるとともに、 他方 を第 3 4図に示した移相回路 8 1 0 Lに置き換えて F M変調装置を構成してもよ い。 Therefore, by replacing one of the two phase shift circuits 810C included in the oscillator 5J with the phase shift circuit 910C shown in the third fel (A), the signal is collected by the condenser microphone. It is possible to configure an FM modulator that uses the sound that has been emitted for the FM modulation signal. Alternatively, one of the two phase-shift circuits 8100C included in the oscillator 5J is replaced with the phase-shift circuit 9110C shown in FIG. 33 (A) [H] and the other is shifted to the third The FM modulator may be configured by replacing the phase shift circuit 8101L shown in FIG.
〔第 1 2の実施形態〕 [First and second embodiments]
3 7図は、 2つの移相 [nl路と位相反転冋路とを用いて構成した他の発振器の 回路図である。 同 ¾に示す発振器 5 Kは、 第 3 0図に示した発振器 5 H内の後段 の移相回路 8 3 0 Cを 2つ用いて縦続接続するとともに、 前段の移相回路 8 3 0 Cの入力側に位相反転回路 8 8 0を接続し、 後段の移相回路 8 3 0 Cの出力を帰 還抵抗 8 7 0を介して位相反転问路 8 8 0の入力側に帰還させている。 FIG. 37 is a circuit diagram of another oscillator configured using two phase shift circuits [nl path and phase inversion path]. The oscillator 5K shown in the same figure is connected in cascade using two phase shift circuits 830C at the subsequent stage in the oscillator 5H shown in FIG. A phase inversion circuit 880 is connected to the input side, and the output of the subsequent phase shift circuit 830C is fed back to the input side of the phase inversion circuit 880 via the feedback resistor 870.
位相反転回路 8 8 0によって信号が反転するため、 2つの移相冋路 8 3 0 Cに よる位相シフ ト量の合計が 1 8 0 ° となるときに、 閉ループを一巡したときの位 相シフ ト量が 3 6 0。 となり、 このときの帰還ループのループゲインを 1以 hに 設定することにより所定の発 動作が行われる。 Since the signal is inverted by the phase inverting circuit 880, when the total phase shift amount of the two phase shift circuits 830C is 180 °, the phase shift when the circuit goes through a closed loop is completed. 3600. By setting the loop gain of the feedback loop to 1 or more h at this time, a predetermined generating operation is performed.
したがって、 発振器 5 Kに含まれる 2つの移相回路 8 3 0 Cのいずれか ·方を 第 3 3図 (B ) に示した移相回路 9 3 0 Cに置き換えることにより、 コンデンサ
マイクによって集音した音声を F M変調信号に用いた F M変調装置を構成するこ とができる。 あるいは、 発振器 5 Kに含まれる 2つの移相回路 8 3 0 Cのいずれ か- 方を第 3 3図 (B ) に示した移相回路 9 3 0 Cに置き換えるとともに、 他方 を第 3 5図に示した移相回路 8 3 0 Lに置き換えて F M変調装置を構成してもよ い Therefore, by replacing one of the two phase shift circuits 830C included in the oscillator 5K with the phase shift circuit 930C shown in FIG. It is possible to configure an FM modulator using the sound collected by the microphone as the FM modulation signal. Alternatively, one of the two phase shifters 830C included in the oscillator 5K is replaced with the phase shifter 930C shown in FIG. 33 (B), and the other is replaced with the one shown in FIG. The FM modulator may be configured by replacing the phase shift circuit 8
ところで、 上述した発振器 5 C、 5 D、 5 E、 5 F、 5 G、 5 H、 5 J、 5 K 等は、 非反転回路と 2つの移相回路あるいは位相反転回路と 2つの移相回路を含 んで構成されており、 接続された 3つの回路の全体によつて所定の周波数におい て合計の位相シフ ト是を 3 6 0 ° にすることにより所定の同調動作を行うように なっている。 したがって、 位相シフ ト量だけに着目すると、 2つの移相回路のど ちらを前段に用いるか、 あるいは上述した 3つの问路をどのような順番で接続す るかはある程度の ΰ由度があり、 必要に応じて接続順番を決めることができる。 By the way, the oscillators 5 C, 5 D, 5 E, 5 F, 5 G, 5 H, 5 J, 5 K, etc., described above, are a non-inverting circuit and two phase shifting circuits or a phase inverting circuit and two phase shifting circuits. And a predetermined tuning operation by setting the total phase shift to 360 ° at a predetermined frequency by all three connected circuits. . Therefore, focusing only on the amount of phase shift, there is a certain degree of freedom as to which of the two phase shift circuits is used in the preceding stage or in which order the three circuits described above are connected. The connection order can be determined as needed.
〔その他の実施形態〕 [Other embodiments]
なお、 本発明は上記実施形態に限定されるものではなく、 本発明の要旨の範囲 内で種々の変形実施が可能である。 Note that the present invention is not limited to the above embodiment, and various modifications can be made within the scope of the present invention.
例えば、 上述した各実施形態においては、 コンデンサマイクとキャパシ夕とを 直列接絞した場合について説明したが、 原理的にはこれら直列接続されたコンデ ンサマイクとキャパシ夕の全体を 1つのキャパシ夕として機能させればよいこと から、 コンデンサマイクに 列に接続したキャパシ夕を省略したり、 コンデンサ マイクとキャパシ夕とを並列接続、 あるいはコンデンサマイクと複数のキャパシ 夕を並列および直列に組み合わせるようにしてもよい。 コンデンサマイクを単体 で用いる ¾合と、 コンデンサマイクとキャパシ夕とを直列または並列あるいはこ れらを組み合わせて接続した場合とを比べると、 後者の場合には接続するキャパ シ夕の静電容量の値を変えることにより全体の静電容 fiの変化の度合い、 すなわ ち F M変調度を調整できる利点がある。 For example, in each of the embodiments described above, the case where the condenser microphone and the capacitor are connected in series has been described. Therefore, the capacity connected to the condenser microphone in a row may be omitted, the condenser microphone and the capacity may be connected in parallel, or the condenser microphone and a plurality of capacity may be combined in parallel and in series. . Comparing the case where a condenser microphone is used alone and the case where a condenser microphone and a capacitor are connected in series or in parallel, or a combination of these, the capacitance of the connected capacitor is the latter. By changing the value, there is an advantage that the degree of change of the overall capacitance fi, that is, the FM modulation degree can be adjusted.
また、 上述した各突施形態においては、 各移相冋路内の各素子の桌子定数を固 定して、 キャリア周波数が固定の F M変調装 iSを実現したが、 各素子定数を可変 して周波数を任意に変更できるようにしてもよい。 例えば第 1図に示した F M変 調装置 1を例にとって説明すると、 移相回路 1 1 0 Cあるいは 3 0 C内の抵抗 1
6、 3 6の少なくとも一方を可変抵抗に置き換えてこの抵抗値を可変することに より、 あるいは移相回路 1 1 0 Cあるいは 3 0 C内のキャパシ夕 1 4— 2、 3 4 の少なくとも一方を可変容量素了-に置き換えてこの静電容量を可変することによ り、 各移相回路により位相シフ ト量を変化させて、 F M変調装置 1から出力する 信号の周波数を変史することができる。 さらに具体的には、 上述した可変抵抗を ゲ一ト電圧が変更可能な F E Tのチヤネル抵抗を利用して形成することができ、 可変容量素子をァノ一ド · カソード間に印加する逆バイアス電圧が変更可能な可 変容量ダイォードによって、 あるいはゲート電圧によってゲート容量が変更可能 な F E Tによって形成することができる。 Also, in each of the above-described embodiments, the FM constants of the FM modulator iS with a fixed carrier frequency are realized by fixing the element constant of each element in each phase shift path. The frequency may be arbitrarily changed. For example, taking the FM modulator 1 shown in FIG. 1 as an example, the phase shift circuit 110 By changing at least one of 6, 6 and 6 with a variable resistor to vary this resistance value, or by changing at least one of the capacity 14-2 or 34 in the phase shift circuit 110 C or 30 C. By changing this capacitance by replacing it with a variable capacitance element, the frequency of the signal output from the FM modulator 1 can be changed by changing the amount of phase shift by each phase shift circuit. it can. More specifically, the above-described variable resistor can be formed by using a channel resistance of a FET whose gate voltage can be changed, and a reverse bias voltage for applying a variable capacitance element between an anode and a cathode can be used. Can be formed by a variable-capacitance diode whose gate capacitance can be changed, or by a FET whose gate capacitance can be changed by a gate voltage.
特に、 F E Tのチャネル抵抗を利用して" J変抵抗を形成する場合には、 pチヤ ネルの F E Tと nチャネルの F E Tとを並列接続して 1つの可変抵抗を構成する ことにより、 F E Tの非線形領域の改善を行うことができ、 高調波成分を低減し た歪みの少ない^号を出力することができる。 同様に、 移相回路内に L R回路が 含まれる場合には、 L R回路内のィンダクタあるいは抵抗を可変ィンダク夕素子 や可変抵抗を用いて構成してもよい。 In particular, when the "J variable resistance" is formed by using the channel resistance of the FET, the p-channel FET and the n-channel FET are connected in parallel to form a single variable resistor, which reduces the nonlinearity of the FET. It is possible to improve the area and output a signal with less distortion with reduced harmonic components Similarly, when the LR circuit is included in the phase shift circuit, the inductor in the LR circuit Alternatively, the resistor may be configured using a variable inductor or a variable resistor.
なお、 I:述した各種の発振器において、 後段の移相回路の出力側に分圧回路を 接続し、 この分/十:回路の人力 ΐ¾ を発振出力として取り出すとともに、 分圧出力 を帰還抵抗 7 0等を介して前段の移相回路の入力側に' させてもよい。 このよ うな分圧回路を設けることにより、 発振 β号を所定の増幅度で増幅することがで きる。 したがって、 第 1図に示した増幅器 2を 略したり、 増幅器 2の培'幅度を 小さくできる。 In addition, I: In each of the various oscillators described above, a voltage divider is connected to the output side of the phase shifter at the subsequent stage, and the manpower of the circuit is extracted as the oscillation output, and the divided output is fed back to the feedback resistor. It may be connected to the input side of the preceding phase shift circuit via 0 or the like. By providing such a voltage dividing circuit, the oscillation β can be amplified with a predetermined amplification degree. Therefore, the amplifier 2 shown in FIG. 1 can be omitted, or the amplification degree of the amplifier 2 can be reduced.
また、 上述した第 1〜第 6の実施形態においては、 オペアンプを用いた移相回 路 1 1 0 C、 3 0 C等を用いて F M変調装置を構成することにより高い安定度を 実現することができるが、 移相回路 1 1 0 C、 3 0 C等のような使い方をする場 合にはオフセッ ト電圧や電圧利得はそれほど高性能なものが要求されないため所 定のゲインを有する差動増幅器を 移相回路内のオペアンプの代わりに使用する ようにしてもよい。 In the first to sixth embodiments described above, high stability is realized by configuring an FM modulator using phase shift circuits 110 C, 30 C, etc. using operational amplifiers. However, when the phase shift circuit is used in 110 C, 30 C, etc., the offset voltage and the voltage gain are not required to be so high, so the differential with the specified gain is required. An amplifier may be used instead of the operational amplifier in the phase shift circuit.
第 3 8図は、 オペアンプの構成の中で移相回路の動作に必要な部分を抽出した 问路図であり、 全体が所定のゲインを有する差動増幅器として動作する。 同図に
示す差動増幅器は、 F E Tにより構成された差動入力段 1 0 0と、 この差動入力 段 1 0 0に定電流を与える定電流回路 1 0 2と、 定電流回路 1 0 2に所定のバイ ァス電圧を与えるバイァス回路 1 0 4と、 差動入力段 1 0 0に接続された出力ァ ンプ 1 0 6とによって構成されている。 同図に示すように、 実際のオペアンプに 含まれている電圧利得を稼ぐための多段増幅回路を省略して、 差動増幅器の構成 を簡略化し、 広帯域化を図ることができる。 このように、 回路の簡略化を行うこ とにより、 動作周波数の上限を; 0ίくすることができるため、 その分この差動増幅 器を用いて構成した F M変調装置の出力周波数の上限を高くすることができる。 産栗上の利用可能性 FIG. 38 is a circuit diagram in which a part necessary for the operation of the phase shift circuit in the configuration of the operational amplifier is extracted, and the whole operates as a differential amplifier having a predetermined gain. In the figure The differential amplifier shown includes a differential input stage 100 composed of FETs, a constant current circuit 102 for supplying a constant current to the differential input stage 100, and a predetermined current supplied to the constant current circuit 102. It comprises a bias circuit 104 for applying a bias voltage and an output amplifier 106 connected to the differential input stage 100. As shown in the figure, the multistage amplifier circuit for gaining the voltage gain included in the actual operational amplifier is omitted, so that the configuration of the differential amplifier can be simplified and a wider band can be achieved. In this way, by simplifying the circuit, the upper limit of the operating frequency can be reduced to 0 °, and accordingly, the upper limit of the output frequency of the FM modulator configured using this differential amplifier is increased. can do. Availability on litter
縦 接続された 2つの移相回路の一方にコンデンサマイクを設け、 コンデンサ マイクで集音した音声を直接 F M変調して出力するため、 コンデンサマイクの静 電容量の変化をいつたん電圧に変換するための付加回路等が不要となり、 F M変 調装置全体の回路構成を簡略化することができる。
A condenser microphone is installed in one of the two phase-shift circuits connected vertically, and the sound collected by the condenser microphone is directly FM-modulated and output, so that changes in the capacitance of the condenser microphone are converted to a voltage. This eliminates the need for additional circuits and the like, and can simplify the circuit configuration of the entire FM modulator.
Claims
1 . 差動増幅器と C R回路とを含む全域通過型の 2つの移相回路を備え、 これら 2つの移相回路を縦続接続して後段の前記移相回路の出力を前段の前記移相 [Ml路 の人力側に帰還させるとともに、 前記 2つの移相回路のいずれか一方に含まれる 前記 C R回路内のキャパシタとしてコンデンサマイクを用いることにより、 前記 2つの移相回路のいずれかから F M変調された 号を出力することを特徴とする F M変調装置。 1. Two phase-shift circuits of the all-pass type including a differential amplifier and a CR circuit are provided, and these two phase-shift circuits are cascaded to output the output of the latter-stage phase-shift circuit to the preceding-stage phase-shifter [Ml By using a condenser microphone as a capacitor in the CR circuit included in one of the two phase-shift circuits, the signal was FM-modulated from one of the two phase-shift circuits. A FM modulator characterized by outputting a signal.
2 . 前 £ 2つの移相回路の少なくとも一方は、 前記差動増幅器の反転入力端子に ー/J端が接続され他方端が前記 C R回路に接続された第 1の抵抗と、 前^差動增 幅器の出力端子と反転入力端子との問に接続された第 2の抵抗とを冇しており、 前記第 1の抵抗を介して前記差動増幅器の反転入力端子に交流信号を人力し、 前 記 C R回路内の前記キャパシ夕と抵抗との接続部を前記差動増幅器の非反転入力 端子に接続したことを特徴とする請求の範囲第 1項記載の F M変調装置。 2. At least one of the two phase-shift circuits includes a first resistor having a − / J terminal connected to the inverting input terminal of the differential amplifier and the other end connected to the CR circuit, A second resistor connected between the output terminal and the inverting input terminal of the amplifier; and an AC signal is manually input to the inverting input terminal of the differential amplifier via the first resistor. 2. The FM modulator according to claim 1, wherein a connection between the capacitor and the resistor in the CR circuit is connected to a non-inverting input terminal of the differential amplifier.
3 . 前記 2つの移相回路の少なくとも一方は、 前記差動増幅器の反転入力端子に - 方端が接続され他方端が前記 C R回路に接続された第 1の抵抗と、 前記差動増 幅器の出力端了に接続された分 回路と、 前記分圧回路の出力端子と前記差動増 幅器の反転入力端子との間に接続された第 2の抵抗とを有しており、 前記第 1の 抵抗を介して前 己差動増幅器の反転入力端子に交流信号を入力し、 前記 C R iuJ路 内の前記キャパシタと抵抗との接続部を前記差動 ¾幅器の非反転人力端子に接続 したことを特徴とする請求の範囲笫 1項 a d載の F M変調装置。 3. At least one of the two phase shift circuits includes a first resistor having a negative terminal connected to the inverting input terminal of the differential amplifier and the other terminal connected to the CR circuit, and the differential amplifier. And a second resistor connected between an output terminal of the voltage divider circuit and an inverting input terminal of the differential amplifier. An AC signal is input to the inverting input terminal of the self-differential amplifier via the resistor of Step 1, and the connection between the capacitor and the resistor in the CR iuJ path is connected to the non-inverting human terminal of the differential amplifier. The FM modulator according to claim 1, characterized in that:
4 . 前記 2つの移相回路の少なくとも一方は、 前記差動増幅器の反転人力端子に 一方端が接続され他方端が ^ C R回路に接続された第 1の抵抗と、 前記差動增 幅器の出力端子と反転入力端子との間に接続された第 2の抵抗と、 一方端が前記 差動増幅器の反転入力端子に接続され他方端が接地された第 3の抵抗とを有して おり、 前記第 1の抵抗を介して前記差動増幅器の反転入力端子に交流信号を入力 し、 前記 C R回路内の キャパシ夕と抵抗との接続部を前記差動増幅器の非反 転入力端子に接続したことを特徴とする請求の範 ffl第 1項記載の F M変調装置。 4. At least one of the two phase shift circuits includes a first resistor having one end connected to an inverted human input terminal of the differential amplifier and the other end connected to a ^ CR circuit, and a differential resistor of the differential amplifier. A second resistor connected between the output terminal and the inverting input terminal; and a third resistor having one end connected to the inverting input terminal of the differential amplifier and the other end grounded, An AC signal is input to the inverting input terminal of the differential amplifier via the first resistor, and a connection between the capacitor and the resistor in the CR circuit is connected to a non-inverting input terminal of the differential amplifier. The FM modulator according to claim 1, wherein the FM modulator is characterized in that:
5 . 前記 2つの移相回路の少なくとも一方は、 抵抗値がほぼ等しい第 1および第 2の抵抗により構成される分圧回路を冇しており、 前記分圧回路の出力端子の ¾
位と前記 C R回路内の前記キャパシ夕および抵抗の接続点の電位との電位差を前 差動増幅器により所定の増幅度で増幅して出力することを特徴とする :求の範 囲第 1項記載の F M変調装置。 5. At least one of the two phase shift circuits indicates a voltage dividing circuit including first and second resistors having substantially equal resistance values, and a voltage dividing terminal of the output terminal of the voltage dividing circuit. Position to said Capacity evening and resistance characteristics to output is amplified by a predetermined amplification degree by the potential difference between the potential of the connection point before the differential amplifier in the CR circuit: determined in range囲第1 wherein FM modulator.
6 . 前記 2つの移相回路を縱続接続して形成される帰還ループの一部に信号の位 相を変えずに出力する非反転回路を接続し、 6. A non-inverting circuit that outputs the signal without changing its phase is connected to a part of a feedback loop formed by cascading the two phase shifting circuits,
前記 C R回路内の前記キャパシ夕と抵抗との接続順序を前記 2つの移相回路で 反対にしたことを特徴とする請求の範囲第 1項記載の F M変調装置。 2. The FM modulator according to claim 1, wherein the order of connection between the capacitor and the resistor in the CR circuit is reversed in the two phase shift circuits.
7 . 前記 2つの移相回路を縦続接続して形成される帰還ループの一部に信号の位 相を反転して出力する位相反転回路を接続し、 7. Connect a phase inversion circuit that inverts the phase of the signal and outputs it to a part of a feedback loop formed by cascading the two phase shift circuits,
前記 C R回路内の前記キャパシ夕と抵抗との接続順序を前記 2つの移相回路で 同じにしたことを特徴とする請求の範囲第 1項記載の F M変調装置。 2. The FM modulator according to claim 1, wherein the order of connection between the capacitor and the resistor in the CR circuit is the same in the two phase shift circuits.
8 . 前記 C R回路の少なくとも一方は、 可変抵抗あるいは可変容量素子を含んで おり、 8. At least one of the CR circuits includes a variable resistor or a variable capacitance element,
前記可変抵抗の抵抗値あるいは前記可変容量素子の静電容量を変えることによ り、 前記 F M変調された信号のキヤリァ周波数を可変することを特徴とする請求 の範囲第 1項記載の F M変調装置。 The FM modulator according to claim 1, wherein a carrier frequency of the FM-modulated signal is varied by changing a resistance value of the variable resistor or a capacitance of the variable capacitance element. .
9 . 差動増幅器と C R回路とを含む全域通過型の第 1の移相回路と、 差動増幅器 と L R回路とを含む全域通過型の第 2の移相回路とを備え、 これら第 1および第 2の移相 路を所定の順序で縦続接絞して後段の前記移相回路の出力を前段の前 記移相回路の入力側に帰還させるとともに、 前記第 1の移相回路に含まれる前記 C R回路内のキャパシ夕としてコンデンサマイクを用いることにより、 前記第 1 および第 2の移相回路のいずれかから F M変調された信号を出力することを特徴 とする F M変調装置。 9. An all-pass type first phase shift circuit including a differential amplifier and a CR circuit, and an all-pass type second phase shift circuit including a differential amplifier and an LR circuit. The second phase shift circuit is cascaded in a predetermined order, and the output of the subsequent phase shift circuit is fed back to the input side of the preceding phase shift circuit, and is included in the first phase shift circuit. An FM modulation device, comprising: outputting a FM-modulated signal from one of the first and second phase shift circuits by using a condenser microphone as a capacity in the CR circuit.
1 0 . 前記第 1および第 2の移相回路の少なくとも一方は、 前記差動増幅器の反 転入力端子に一方端が接続され他方端が前記 C R回路あるいは前記 L R回路に接 続される第 1の抵抗と、 前記差動増幅器の出力端子と反 入力端子との間に接続 される第 2の抵抗とを有しており、 前記第 1の抵抗を介して前記差動増幅器の反 転入力端子に交流信号を入力し、 前記 C R回路内の前記キヤパシ夕と抵抗との接 続部あるいは前記 L R回路内のィンダク夕と抵抗との接続部を前記差動増幅器の
非反転人力端— f-に接続したことを特徴とする請求の範囲第 9項記載の F M変調装 10. At least one of the first and second phase shift circuits has a first end connected to the inverting input terminal of the differential amplifier and a second end connected to the CR circuit or the LR circuit. And a second resistor connected between the output terminal and the non-input terminal of the differential amplifier, and the inverting input terminal of the differential amplifier via the first resistance. An AC signal is input to the differential amplifier, and the connection between the capacitor and the resistor in the CR circuit or the connection between the inductor and the resistor in the LR circuit is connected to the differential amplifier. 10. The FM modulator according to claim 9, wherein the FM modulator is connected to a non-inverting human input terminal f-.
1 1 . 前記第 1および第 2の移相回路の少なくとも一方は、 前記差動増幅器の反 転入力端子に-方端が接続され他方端が前記 C R回路あるいは前記 L R回路に接 続された第 1の抵抗と、 前記差動増幅器の出力端子に接続された分圧回路と、 前 d分圧回路の出力端子と前記差動増幅器の反転人力端子との間に接続された第 2 の抵抗とを有しており、 前記第 1の抵抗を介して前記差動増幅器の反転入力端子 に交流信号を入力し、 前記 C R ln]路内の前記キャパシ夕と抵抗との接続部あるい は前記 L R回路内のィンダク夕と抵抗との接続部を前記差動増幅器の非反転入力 端子に接続したことを特徴とする請求の範囲第 9項記載の F M変調装置。 At least one of the first and second phase shift circuits has a negative end connected to the inverting input terminal of the differential amplifier and a second end connected to the CR circuit or the LR circuit. A voltage divider connected to the output terminal of the differential amplifier; a second resistor connected between the output terminal of the preceding voltage divider and the inverting human terminal of the differential amplifier. An AC signal is input to the inverting input terminal of the differential amplifier via the first resistor, and a connection between the capacitor and the resistor in the CR ln] path or the LR is input. 10. The FM modulator according to claim 9, wherein a connection between the inductor and the resistor in the circuit is connected to a non-inverting input terminal of the differential amplifier.
1 2 . 前記第 1および第 2の移相回路の少なくとも一方は、 前記差動増幅器の反 転人力端子に一方端が接続され他方端が前記 C R回路あるいは前記 L R回路に接 続された第 1の抵抗と、 前記差動増幅器の出力端子と反転人力端子との間に接続 された第 2の抵抗と、 一方端が前記差動増幅器の反転入力端子に接続され他方端 が接地された第 3の抵抗とを有しており、 前記第 1の抵抗を介して ^記差動増幅 器の反転入力端子に交流信 を入力し、 前記 C R冋路内の前記キャパシ夕と抵抗 との接続部あるいは前記 L R回路内のィンダクタと抵抗との接続部を前記差動增 幅器の非反転入力端子に接続したことを特徴とする請求の範囲第 9 ί貝記載の F M 変調装^。 12. At least one of the first and second phase shift circuits has a first end connected to an inverting input terminal of the differential amplifier and a second end connected to the CR circuit or the LR circuit. A second resistor connected between the output terminal of the differential amplifier and the inverted human input terminal; and a third resistor having one end connected to the inverted input terminal of the differential amplifier and the other end grounded. AC signal is input to the inverting input terminal of the differential amplifier via the first resistor, and the connection between the capacitor and the resistor in the CR circuit or 10. The FM modulator according to claim 9, wherein a connection between the inductor and the resistor in the LR circuit is connected to a non-inverting input terminal of the differential amplifier.
1 3 . 前記第 1および第 2の移相回路の少なくとも一方は、 抵抗値がほぼ等しい 第 1および第 2の抵抗により構成される分圧回路を右しており、 前記分圧回路の 出力端子の電位と前記 C R回路内の前記キャパシ夕および抵抗の接続点の電位と の電位差、 あるいは前記分圧回路の出力端子の電位と前記 L R回路内の前記ィン ダク夕および抵抗の接続点の ¾t位との電位差を
動増幅器により所定の増幅 度で増幅して出力することを特徴とする請求の範囲第 9项記載の F M変調装置。 1 3. At least one of the first and second phase shift circuits is connected to a voltage divider composed of first and second resistors having substantially equal resistance values, and an output terminal of the voltage divider is provided. Or the potential difference between the potential of the capacitor and the connection point of the resistor in the CR circuit, or the potential of the output terminal of the voltage divider circuit and the ¾t of the connection point of the inductor and the resistance in the LR circuit. Potential difference 10. The FM modulator according to claim 9, wherein the output is amplified by a dynamic amplifier at a predetermined amplification factor.
1 4 . 前記第 1および第 2の移相回路を縦続接続して形成される帰還ループの一 部に信号の位相を変えずに出力する非反転回路を接続するとともに、 前記 C R回 路内のキャパシ夕あるいは前記 L R回路內のィンダク夕からなるリアクタンス素 子と前記 C R回路内の抵抗あるいは前記 L R回路内の抵抗との接続順序を前記 2
つの移相问路のそれぞれで同じにしたことを特徴とする請求の範囲第 9項記載の F M変調装置。 14. A non-inverting circuit that outputs a signal without changing the phase of a signal is connected to a part of a feedback loop formed by cascading the first and second phase shift circuits. The connection order of the capacitance element or the reactance element consisting of the inductance element of the LR circuit and the resistance in the CR circuit or the resistance in the LR circuit is defined by 10. The FM modulator according to claim 9, wherein each of the two phase shift circuits is the same.
1 5 . 前 ¾第 1および第 2の移相问路を縦続接続して形成される帰還ループの -- 部に信号の位相を変えて出力する位相反転回路を接続するとともに、 前記 C R回 路内のキャパシタあるいは前記 L R回路内のィンダク夕からなるリアクタンス素 子と前記 C R回路内の抵抗あるいは前記 L R回路内の抵抗との接続順序を前記 2 つの移相回路のそれそれで反対にしたことを特徴とする請求の範囲第 9項記載の F M変調装置。 1 5. A phase inverting circuit for changing the phase of a signal and outputting the signal is connected to the-part of a feedback loop formed by cascading the first and second phase shift circuits, and the CR circuit is provided. The connection order of the capacitor in the LR circuit or the reactance element consisting of the inductor in the LR circuit and the resistance in the CR circuit or the resistance in the LR circuit is reversed for each of the two phase shift circuits. 10. The FM modulator according to claim 9, wherein:
1 6 . 前記 C R回路あるいは d il L R回路の少なくとも一方は、 可変抵抗あるい は可変容量素子を含んでおり、 16. At least one of the CR circuit and the dil L R circuit includes a variable resistor or a variable capacitance element,
前記可変抵抗の抵抗値あるいは前記可変容量素子の静電容量を変えることによ り、 前記 F M変調された信号のキャリア周波数を可変することを特徴とする請求 の範囲第 9項記載の F M変調装置。 The FM modulator according to claim 9, wherein the carrier frequency of the FM-modulated signal is varied by changing a resistance value of the variable resistor or a capacitance of the variable capacitance element. .
1 7 . 人力された交流信号を同相および逆相の交流信号に変換して出力する変換 手段と、 この変換手段によって変換された一方の交流信号を C R回路の一方端を 介して、 他方の交流信号を前記 C R回路の他方端を介して合成する合成手段とを 含む全域通過型の 2つの移相回路と、 17. Conversion means for converting a human-powered AC signal into in-phase and out-of-phase AC signals and outputting the same, and converting one AC signal converted by the conversion means to the other AC through one end of a CR circuit. Two all-pass type phase shift circuits including synthesis means for synthesizing a signal through the other end of the CR circuit;
入力された交流信号の位相を変えずに出力する非反転回路とを備え、 A non-inverting circuit that outputs the input AC signal without changing the phase,
前記 2つの移相回路と前記非反転回路とを所定の順序で縦続接続し、 最終段の 回路の出力を初段の [E]路の人力側に帰還させるとともに、 前記 2つの移相回路の いずれか一方に含まれる前記 C R回路内のキャパシ夕としてコンデンサマイクを 用いることにより、 前記 2つの移相回路のいずれかから F M変調された信 を出 力し、 前記 C R回路内の前記キャパシ夕と抵抗との接続順序を前記 2つの移相回 路で反対にしたことを特徴とする F M変調装置。 The two phase-shift circuits and the non-inverting circuit are cascaded in a predetermined order, and the output of the last-stage circuit is fed back to the human-powered side of the first-stage [E] path. By using a condenser microphone as a capacitor in the CR circuit included in either of the two circuits, an FM-modulated signal is output from one of the two phase shift circuits, and the capacitor and the resistor in the CR circuit are output. Characterized in that the order of connection with the two phase shift circuits is reversed.
1 8 . 前記 2つの移相回路内の前 d変換手段はトランジスタを含んでおり、 前記 トランジスタのソースおよびドレイン、 あるいはェミッ夕およびコレク夕にそれ それ抵抗値がほぼ等しい抵抗を接続し、 前記トランジス夕のゲ一トあるいはベー スに交流信号を入力し、 前記トランジスタのソース ' ドレイン間あるいはェミツ 夕 · コレクタ間に前記合成手段を構成する前記 C R 路を接続したことを特徴と
する請求の範囲第 1 7頌記載の F M変調装置。 18. The pre-d conversion means in the two phase shift circuits includes a transistor, and a source and a drain of the transistor, or a resistor having substantially the same resistance value connected to an emitter and a collector, respectively, are connected to the transistor. An AC signal is inputted to a gate or a base in the evening, and the CR path constituting the synthesizing means is connected between a source and a drain of the transistor or between an emitter and a collector. 17. The FM modulator according to claim 17, wherein
1 9 . 前記 C R回路の少なくとも一方は、 可変抵抗あるいは可変容量素子を含ん でおり、 1 9. At least one of the CR circuits includes a variable resistor or a variable capacitance element,
前記可変抵抗の抵抗値あるいは前記可変容量素子の静電容量を変えることによ り前記 F M変調された信号のキヤリァ周波数を可変することを特徴とする請求の 範囲第 1 7項記載の F M変調装置。 18. The FM modulator according to claim 17, wherein a carrier frequency of the FM-modulated signal is varied by changing a resistance value of the variable resistor or a capacitance of the variable capacitance element. .
2 0 . 入力された交流信号を同相および逆相の交流信号に変換して出力する変換 手段と、 この変換手段によって変換された -方の交流信号を C R回路の一方端を 介して、 他方の交流信号を前記 C R回路の他方端を介して合成する合成手段とを 含む全域通過型の 2つの移相回路と、 20. A conversion means for converting the input AC signal into an in-phase and an in-phase AC signal and outputting the converted signal, and the negative AC signal converted by the conversion means via one end of the CR circuit to the other side. Two all-pass type phase shift circuits including synthesis means for synthesizing the AC signal through the other end of the CR circuit;
入力された交流 ίΓί ^の位相を反転して出力する位相反転回路とを備え、 前記 2つの移相回路と前記位相反転回路とを所定の順序で縦絞接続し、 玆終段 の回路の出力を初段の回路の入力側に 還させるとともに、 前記 2つの移相回路 のいずれか一方に含まれる前記 C R回路内のキャパシ夕としてコンデンサマイク を用いることにより、 前記 2つの移相回路のいずれかから F Μ変調された信号を 出力し、 前記 C R回路内の前 ^キヤパシ夕と抵抗との接続順序を前記 2つの移相 回路で同じにしたことを特徴とする F M変調装置。 A phase inverting circuit for inverting the phase of the input AC ίΓί ^ and outputting the inverted AC, and vertically connecting the two phase shifting circuits and the phase inverting circuit in a predetermined order; To the input side of the first-stage circuit, and by using a condenser microphone as a capacitor in the CR circuit included in one of the two phase-shift circuits, a signal from one of the two phase-shift circuits is obtained. F. An FM modulator, which outputs a modulated signal, and wherein the connection order of the front capacitor and the resistor in the CR circuit is the same in the two phase shift circuits.
2 1 . 前 ^ 2つの移相冋路内の前記変換手段はトランジスタを含んでおり、 前記 トランジスタのソースおよびドレイン、 あるいはエミッ夕およびコレクタにそれ それ抵抗値がほぼ等しい抵抗を接続し、 前記トランジス夕のゲ一卜あるいはべ一 スに交流信^を入力し、 前記トランジスタのソース · ドレイン間あるいはェミツ 夕 · コレクタ間に前記合成手段を構成する前記 C R回路を接続したことを特徴と する請求の範囲第 2 0項記載の F M変調装置。 2 1. The conversion means in the two phase shift circuits include a transistor, and a source and a drain or an emitter and a collector of the transistor are connected to resistors having substantially equal resistance values, and the transistor is connected to the transistor. An AC signal is input to a gate or a base in the evening, and the CR circuit constituting the synthesizing means is connected between a source and a drain of the transistor or between an emitter and a collector of the transistor. FM modulator according to claim 20.
2 2 . 前記 C R回路の少なくとも一方は、 可変抵抗あるいは可変容量素子を含ん でおり、 22. At least one of the CR circuits includes a variable resistor or a variable capacitance element,
前記可変抵抗の抵抗値あるいは前記可変容 素 fの静電容量を変えることによ り、 前記 F M変調された信号のキヤリァ周波数を可変することを特徴とする請求 の範囲第 2 0項記載の F M変調装置。 The FM carrier according to claim 20, wherein a carrier frequency of said FM-modulated signal is varied by changing a resistance value of said variable resistor or a capacitance of said variable capacitor f. Modulation device.
2 3 . 入力された交流 β号を同相および逆相の交流信号に変換して出力する第 1
の変換手段と、 この第 1の変換手段によって変換された一方の交流信号を C R回 路の- -方端を介して、 他方の交流信号を前記 C R回路の他方端を介して合成する 第 1の合成 段とを含む全域通過型の第 1の移相回路と、 2 3. The first to convert the input AC signal into an in-phase and anti-phase AC signal and output it. And one AC signal converted by the first conversion means is synthesized via the-end of the CR circuit, and the other AC signal is synthesized via the other end of the CR circuit. An all-pass type first phase shift circuit including a synthesis stage of
入力された交流信号を同相および逆相の交流信号に変換して出力する第 2の変 換手段と、 この第 2の変換手段によって変換された一方の交流信号を L R回路の 一方端を介して、 他方の交流信号を前記 L R回路の他方端を介して合成する第 2 の合成手段とを含む全域通過型の第 2の移相回路と、 A second converting means for converting the input AC signal into an in-phase and an opposite-phase AC signal and outputting the AC signal; and converting one of the AC signals converted by the second converting means via one end of the LR circuit An all-pass second phase-shift circuit including second synthesizing means for synthesizing the other AC signal via the other end of the LR circuit;
入力された交流信号の位相を変えずに出力する非反転回路とを備え、 A non-inverting circuit that outputs the input AC signal without changing the phase,
前記第 1および第 2の移相回路と前記非反転回路とを所定の順序で縦続接続し、 最終段の回路の出力を初段の回路の入力側に帰還させるとともに、 前記第 1およ び第 2の移相回路のいずれか一方に含まれる前 d C R回路内のキャパシ夕として コンデンサマイクを用いることにより、 前記 2つの移相回路のいずれかから F M 変調された信り-を出力し、 前記 C R回路内のキャパシ夕あるいは前記 L R回路内 のィンダク夕からなるリアクタンス素子と前記 C R回路内の抵抗あるいは前記 L R回路内の抵抗との接続順序を前記 2つの移相回路のそれぞれで同じにしたこと を特徴とする F M変調装置。 The first and second phase shift circuits and the non-inverting circuit are cascaded in a predetermined order, and the output of the last-stage circuit is fed back to the input side of the first-stage circuit. By using a condenser microphone as a capacitor in the previous d CR circuit included in one of the two phase shift circuits, the FM-modulated signal is output from one of the two phase shift circuits, and The order of connection between the capacity element in the CR circuit or the reactance element consisting of the inductance element in the LR circuit and the resistance in the CR circuit or the resistance in the LR circuit is the same for each of the two phase shift circuits. FM modulation device characterized by the following.
2 4 . 前記第 1および第 2の移相回路内の前記変換手段はトランジスタを含んで おり、 前記トランジスタのソースおよびドレイン、 あるいはエミッ夕およびコレ クタにそれぞれ抵抗値がほぼ等しい抵抗を接続し、 前記トランジス夕のゲー卜あ るいはべ一スに交流信号を入力し、 前記トランジスタのソース · ドレイン問ある いはエミッ夕 ·コレクタ間に前記合成手段を構成する前記 C R回路あるいは前記 L R回路を接続したことを特徴とする請求の範囲第 2 3項記載の F M変調装置。 24. The conversion means in the first and second phase shift circuits includes a transistor, and a source and a drain of the transistor, or an emitter and a collector, which are connected with resistors having substantially equal resistances, respectively. An AC signal is input to the gate or base of the transistor, and the CR circuit or the LR circuit constituting the synthesizing means is connected between the source and drain of the transistor or between the emitter and the collector. The FM modulator according to claim 23, wherein the frequency modulation is performed.
2 5 . 前 C R回路の少なくとも一方は、 可変抵抗あるいは可変容量素子を含ん でおり、 25. At least one of the front CR circuits includes a variable resistor or a variable capacitance element.
前記可変抵抗の抵抗値あるいは前記可変容量素 /-の静電容 _¾を変えることによ り前記 F M変調された信号のキヤリァ周波数を可変することを特徴とする請求の 範囲第 2 3項記載の F M変調装置。 The FM carrier according to claim 23, wherein a carrier frequency of the FM-modulated signal is varied by changing a resistance value of the variable resistor or a capacitance _¾ of the variable capacitor element. Modulation device.
2 6 . 入力された交流信 1を同相および逆相の交流信号に変換して出力する第 1 の変換手段と、 この第 1の変換手段によって変換された一方の交流信号を C R「口 I
路の一方端を介して、 他方の交流信号を前記 C R冋路の他方端を介して合成する 第 1の合成手段とを含む全域通過型の第 1の移相回路と、 26. First conversion means for converting the input AC signal 1 into in-phase and opposite-phase AC signals and outputting the same, and converting one of the AC signals converted by the first conversion means into a CR "port I A first phase shift circuit of an all-pass type including, via one end of the path, first synthesizing means for synthesizing the other AC signal through the other end of the CR path;
人力された交流信号を同相および逆相の交流信号に変換して出力する第 2の変 換手段と、 この第 2の変換手段によって変換された一方の交流信号を L R回路の 一方端を介して、 他方の交流信号を前記 L R回路の他方端を介して合成する第 2 の合成手段とを含む全域通過型の第 2の移相回路と、 A second converter for converting the input AC signal into an in-phase and an opposite-phase AC signal and outputting the AC signal; and converting one of the AC signals converted by the second converter into one end of an LR circuit. An all-pass second phase-shift circuit including second synthesizing means for synthesizing the other AC signal via the other end of the LR circuit;
入力された交流信号の位相を反転して出力する位相反転回路とを備え、 前記第 1および第 2の移相冋路と前記位相反転回路とを所定の順序で縦続接続 し、 最終段の问路の出力を初段の回路の入力側に帰還させるとともに、 前記第 1 および第 2の移相冋路のいずれか一方に含まれる前記 C R回路内のキャパシ夕と してコンデンサマイクを用いることにより、 前記 2つの移相回路のいずれかから F M変調された信号を出力し、 前記 C R回路内のキャパシタあるいは前記 L R回 路内のィンダク夕からなるリアクタンス素子と前記 C R回路内の抵抗あるいは前 記 L R回路内の抵抗との接続順序を前記 2つの移相回路のそれそれで反対にした ことを特徴とする F M変調装置。 A phase inverting circuit for inverting the phase of the input AC signal and outputting the inverted signal, cascade-connecting the first and second phase shifting circuits and the phase inverting circuit in a predetermined order, The output of the circuit is fed back to the input side of the first stage circuit, and a condenser microphone is used as a capacity in the CR circuit included in one of the first and second phase shift circuits. An FM-modulated signal is output from one of the two phase shift circuits, and a capacitor in the CR circuit or a reactance element including an inductor in the LR circuit and a resistor in the CR circuit or the LR circuit described above. Characterized in that the order of connection with the internal resistors is reversed for each of the two phase shift circuits.
2 7 . ^記第 1および第 2の移相回路内の前記変換手段はトランジス夕を含んで おり、 前記トランジスタのソースおよびドレイン、 あるいはエミッ夕およびコレ クタにそれそれ抵抗値がほぼ等しい抵抗を接続し、 前記トランジスタのゲートあ るいはベースに交流信号を入力し、 前記トランジスタのソース ' ドレイン間ある いはエミッ夕 ·コレクタ間に前 gd合成手段を構成する前記 C R回路あるいは前記 L R回路を接続したことを特徴とする請求の範囲第 2 6項記載の F M変調装置。 27. The conversion means in the first and second phase shift circuits includes a transistor, and the source and the drain of the transistor or the emitter and the collector have resistances substantially equal to each other. An AC signal is input to the gate or base of the transistor, and the CR circuit or the LR circuit constituting the gd synthesizing means is connected between the source and the drain of the transistor or between the emitter and the collector. 27. The FM modulator according to claim 26, wherein the frequency modulation is performed.
2 8 . 前記 C R回路あるいは前記 L R回路の少なくとも -方は、 可変抵抗あるい は可変容量素子を含んでおり、 28. At least one of the CR circuit and the LR circuit includes a variable resistor or a variable capacitance element,
前記可変抵抗の抵抗値あるいは前記可変容量素子の静電容量を変えることによ り、 前記 F M変調された信号のキヤリァ ;)波数を可変することを特徴とする請求 の範囲第 2 6項記載の F M変調装置。
27. The carrier of the FM-modulated signal is varied by changing the resistance value of the variable resistor or the capacitance of the variable capacitance element, and the wave number is varied. FM modulator.
Priority Applications (1)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
AU61377/96A AU6137796A (en) | 1995-11-15 | 1996-06-20 | Fm modulator |
Applications Claiming Priority (2)
Application Number | Priority Date | Filing Date | Title |
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JP7/322133 | 1995-11-15 | ||
JP32213395 | 1995-11-15 |
Publications (1)
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WO1997018624A1 true WO1997018624A1 (en) | 1997-05-22 |
Family
ID=18140304
Family Applications (1)
Application Number | Title | Priority Date | Filing Date |
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PCT/JP1996/001705 WO1997018624A1 (en) | 1995-11-15 | 1996-06-20 | Fm modulator |
Country Status (3)
Country | Link |
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AU (1) | AU6137796A (en) |
TW (1) | TW297979B (en) |
WO (1) | WO1997018624A1 (en) |
Citations (7)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
JPS5029024A (en) * | 1973-07-18 | 1975-03-24 | ||
JPS52120657A (en) * | 1976-04-02 | 1977-10-11 | Fujitsu Ltd | Voltage control oscillator |
JPS54959A (en) * | 1977-06-06 | 1979-01-06 | Mitsubishi Electric Corp | Phase modulation circuit |
JPS5427306A (en) * | 1977-08-02 | 1979-03-01 | Nec Corp | Instantaneous frequency deviation control circuit |
JPS5947483B2 (en) * | 1974-07-10 | 1984-11-19 | エヌ・ベー・フイリツプス・フルーイランペンフアブリケン | Circuit arrangement that converts bridge unbalance into frequency change |
JPH0575387A (en) * | 1991-09-17 | 1993-03-26 | Sanyo Electric Co Ltd | Variable delay circuit |
JPH05183406A (en) * | 1991-12-27 | 1993-07-23 | Nec Eng Ltd | Automatic phase correction circuit |
-
1996
- 1996-06-20 WO PCT/JP1996/001705 patent/WO1997018624A1/en active Application Filing
- 1996-06-20 AU AU61377/96A patent/AU6137796A/en not_active Abandoned
- 1996-07-11 TW TW85108397A patent/TW297979B/en not_active IP Right Cessation
Patent Citations (7)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
JPS5029024A (en) * | 1973-07-18 | 1975-03-24 | ||
JPS5947483B2 (en) * | 1974-07-10 | 1984-11-19 | エヌ・ベー・フイリツプス・フルーイランペンフアブリケン | Circuit arrangement that converts bridge unbalance into frequency change |
JPS52120657A (en) * | 1976-04-02 | 1977-10-11 | Fujitsu Ltd | Voltage control oscillator |
JPS54959A (en) * | 1977-06-06 | 1979-01-06 | Mitsubishi Electric Corp | Phase modulation circuit |
JPS5427306A (en) * | 1977-08-02 | 1979-03-01 | Nec Corp | Instantaneous frequency deviation control circuit |
JPH0575387A (en) * | 1991-09-17 | 1993-03-26 | Sanyo Electric Co Ltd | Variable delay circuit |
JPH05183406A (en) * | 1991-12-27 | 1993-07-23 | Nec Eng Ltd | Automatic phase correction circuit |
Also Published As
Publication number | Publication date |
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AU6137796A (en) | 1997-06-05 |
TW297979B (en) | 1997-02-11 |
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