WO1996019889A1

WO1996019889A1 - Fek modem for high frequency communications

Info

Publication number: WO1996019889A1
Application number: PCT/GB1995/002999
Authority: WO
Inventors: Robert Graham Wilkinson
Original assignee: The Secretary Of State For Defence
Priority date: 1994-12-21
Filing date: 1995-12-21
Publication date: 1996-06-27
Also published as: AU4270396A; GB9425801D0

Abstract

A communications system for operation in the high frequency spectrum comprising a transmitter comprises a number of tone frequency channels (TFCs) equal to 2P (P=integer), the transmitter being arranged to transmit each character of a digital message as a coded series of bits such that each bit is transmitted Q times where Q=P/n (Q=integer>1; n=1 or 2 or 3 etc.); the TFCs beging arranged in P pairs with a transmission on one TFC of each pair representing a '1' and on the other TFC representing a '0' and the transmissions on each TFC being arranged such that there is a guard period between any two consecutive bit transmissions; and a receiver comprising means to synchronise the receiver; means to periodically measure a known reference tone signal for each of the 2P TFCs so as to determine a reference signal Ri (i=1 to 2P) indicative of the channel signal amplitude appropriate to a received bit; means to periodically determine a noise estimate Ni for each TFC and each received bit in that TFC from measurements before andafter the expected time of the received bit; means to assign a signal threshold T; to each TFC and to each received bit from the measurements Ri and Ni; means to measure the signal amplitude Si of the bit above the threshold Ti; and means to assign a relative quality value to each TFC i and for each bit based on the noise estimate Ni in that channel and the minimum Nm of the measured noise estimates N1 to N2p; the arrangement being that for each of the 2P TFCs a signal estimate is determined from the measured bit signal to noise, the channel signal to noise and the channel relative quality.

Description

FEK MODEM FOR HIGH FREQUENCY COMMUNICATIONS.

The invention relates to high frequency (HF) communications and in particular to modems for producing reliable long range skywave communications in the presence of disturbance in the HF medium without a need to use high transmitter power.

Despite the difficulties to HF communications due to random ionospheric effects and interference from other HF band users the medium is capable of being exploited for communications purposes. HF communications can offer a high performance to price ratio and considerable flexibility of operation to the user. In addition the random interference effects caused by the medium can be exploited where covert long range communication is needed as the effects of the medium which make HF communications difficult also make interception difficult.

In the past Morse signalling has been successful largely because of the ability of skilled operators to "fill in" and interpret corrupted messages and to be able to adjust their receivers to improve reception. It was assumed that an electronic signal processing system could not improve on Morse signalling.

GB Patent number 2092415 describes a high integrity modem for HF communications in which messages are transmitted consecutively at a data rate of about 10 bits per second over a number of narrow band frequency channels. Each of the frequency channels is assessed for quality and the channel signals combined using an 'intelligent' decision-making system to provide an output signal surpassing the quality of Morse signalling. The quality of the frequency channels is established by measurement of the status of a bit determined from bit intervals immediately before the signal bit interval (pre-data bits) and bit intervals after the signal data bit (post-data bits). Each signal data bit with its associated bit status is measured in each of the frequency channels and the decision algorithm applied combines the measured signal bits with their respective bit status. In this HF modem the transmitted data rate requires only low transmitter power because a small detection bandwidth and optimum channel combining is employed to reject in-band interference. Thus the low power leads to improved covert transmission and the narrow band reception improves the rejection of HF interference. The modem of GB 209 415 uses synchronous timing detection for signals and noise measurements are taken at each transmitted channel frequency.

The object of the present invention is to provide an improved HF modem providing higher communications availability and reliability than previously possible. A second object of the invention is to provide a higher data throughput in such a modem.

The invention provides a communications system for operation in a band of the high frequency (HF) spectrum comprising transmitter and receiver wherein the transmitter comprises: a number of tone frequency channels equal to 2P, where P is an integer, within the HF band; the transmitter being arranged to transmit each character of a digital message as a coded series of bits; the transmitter being further arranged such that each bit is transmitted a number of times Q where Q=P/n with Q an integer greater than 1 and n an integer taking the value 1 or 2 or 3 etc; the tone frequency channels being arranged in P pairs with a transmission on one tone frequency channel of each pair representing a logic state "1" and on the other tone frequency channel representing a logic state "0" and the transmissions on each tone frequency channel being arranged such that there is a guard period between any two consecutive bit transmissions; the receiver comprises: means to synchronise the receiver to a received transmission; means to periodically measure a known reference tone signal appropriate to each of the 2P tone frequency channels so as to determine for each tone frequency channel a reference signal R_i (for i = 1 to 2P) indicative of the channel signal amplitude appropriate to a received bit; means to periodically determine a noise estimate N_ι for each tone frequency channel and appropriate to each received message bit in that tone frequency channel from measurements before and after the expected time of the received message bit; means to assign a signal threshold T. to each tone frequency channel and appropriate to each received message bit from the measurements R_ι and

means to measure the signal amplitude S_j of the message bit above the threshold T. ; and means to assign a relative quality value to each tone frequency channel i and for each message bit based on the noise estimate N_α in that channel and the minimum N_m of the measured noise estimates N. to N_2p; the arrangement being that for each of the 2P tone frequency channel a signal estimate is determined from the measured bit signal to noise, the channel signal to noise and the channel relative quality.

The invention uses an adaptive threshold level to be applied to signal measurements, it measures the relative quality of the different tone channels, and it makes use of known bit transmissions to measure each channel signal to noise. The known reference tone bits may be start and stop bits of opposite polarity transmitted respectively before and after each coded character. In an alternative arrangement a recursive method for determining the reference tone bit is adopted where the signal estimates of bit polarities are used to determine when active known bit transmissions occur and measurement of a known bit is taken as the reference signal.

The transmitter may be such that the bit repetitions Q are arranged singly or in equal groups of tone frequency channels for transmission in consecutive time intervals. For example, in an arrangement where n=l the transmitter transmits each bit consecutively P (=Q) times or if P is divisible by 2 then each bit may be transmitted twice in respective groups of P/2 tone frequency channels to represent a logic "1" and in the other respective groups of P/2 tone frequency channels to represent a logic "0". By transmitting a bit on one of a group of tone frequencies a different bit may be transmitted on each group. In a further arrangement, the Q transmissions for each bit are not consecutive but are multiplexed with other bit transmissions for further time diversity. In addition the Q frequency transmissions may use different preselected frequency sets for each bit transmission. With the above implementation of signal processing the performance of the HF communications has been found superior to that described in GB 209 -41 . In the preferred arrangement the channel relative quality value is the quotient N_m/N_i; the measured bit signal to noise is (S_J-T_JJ/N. and the channel signal to noise is R_j/N .

Where stop/start reference bits are transmitted the channel signal R_i is measured just once for each character (7 message bits) while the noise N_j and signal bit S. are measured in each bit time interval. The noise T is taken as the arithmetic mean of measurements in adjacent channel bit periods when there is no bit transmission. Such noise measurements may be taken both during the guard period between successive bit transmissions and also during bit periods when the signal estimate shows there is no bit transmission. The noise measurements can take place over any number of baud periods providing always that they are not made during an active data bit transmission or in the baud period immediately following an active data bit where multipath interference is likely to occur. The more noise samples that can be taken in a given time period the better is the result. The channel signal to noise, being based on a known bit in each channel frequency, is based only on a known signal bit appearing at predetermined times in each logic state "0" or each logic state "1" channel frequency appropriate to the known signal bit. Where there are no stop/start reference bits measurements of every known active data bit are preferably used for the reference signals.

Advantageously the bit signal to noise is made bipolar by subtracting from each measured bit signal S. in a character an appropriate value of the threshold T. for that character. The preferred channel signal estimate Q_i is then given by:

Q. = [ (S. -T. ) /N. ] x [R^N. ] x [N_m/N. ]

The signal estimate Q_i is calculated for each of the 2P channel tone frequencies and appropriate ones of the 2P values Q_: to Q₂ are then summed to give a quality assessment total Z, the sign of which gives the bit value (positive corresponding to logic "1" and negative to logic "0") and the magnitude gives a confidence measure for the bit value.

Advantageously the values of Q. for each message bit are stored such that on re-transmission of the message the stored values can be added to the new values for the re-transmitted message and the resultant values used for assignment of the bit polarity in each case.

In a preferred arrangement the 2P tones are equi-spaced in frequency within a 3kHz bandwidth. In such an arrangement the transmissions can be arranged to fall within a standard 3kHz HF receiver radio channel.

In a slow speed modem there may conveniently be 10 tone frequencies separated by 200 Hz with a bit rate of transmission of 10 bits per sec. In this arrangement each message bit is transmitted as a 20 msec pulse consecutively on 5 different tones in the bit period of 100msec. However, the bandwidth, number of channels and channel spacing are entirely optional.

In modems for operation at higher data rates the 2P tone frequency channels may be arranged for transmission of 2 or more data bits with groups of 2 or more of the tone frequency channels being available for simultaneous transmission. Advantageously there are 12 tone frequency channels arranged for simultaneous transmission on 3 tones.

Preferably the receiver includes 2P band-pass filters each with a centre frequency corresponding to a different one of the transmitted tone frequencies and a pass-band which is as narrow as possible (taking into account Doppler requirements) . In one particular embodiment each filter has a stop band rejection of at least 70dB for frequencies more than the baud rate removed from the centre frequency. Thus for the above example of 10 tones and a bit rate of 10 bits/sec and a baud rate of 0 the filter bandwidth (-70dB frequency) is 50Hz.

The invention will now be described by way of example only with reference to the accompanying Drawings of which:

Figure 1 shows the signal spectrum of a known high frequency transmitter modem which is used in the present invention;

Figure 2 shows the tone frequency to bit mapping scheme for a high frequency (HF) communications system according to the present invention, making use of the Figure 1 signal spectrum;

Figure 3 illustrates the transmission format immediately prior to transmission of a message using the Figure 2 scheme;

Figures 4 and 5 are block diagrams of HF transmitter and receiver according to the invention;

Figure 6 is a block diagram of the Demodulator Algorithm Unit shown in

Figure 5;

Figure 7 is a diagram illustrating the adaptive threshold applied in each of the receiver signal tone channels;

Figure 8 shows a received signal and a tone channel detector timing arrangement;

Figure 9 is a schematic block diagram of a recursive signal reference and noise level arrangement; and Figures 10 11 and 12.

GB Patent No 2092415 describes a high frequency (HF) slow speed communications modem in which a 3kHz audio channel bandwidth at HF is divided so that it contains 10 tones. The modem is optimised for 10 bits per second information transmission. To combat the effects of HF fading each message bit is transmitted five times such that the baud rate is five times greater than the bit rate i.e. 50 bauds/sec. In the normal mode of operation a pair of the 10 tones is selected for transmission, with one of the tones representing logic state "1" and the other tone representing logic state "0". According to the logic state of the bit to be transmitted the appropriate tone of the pair is transmitted for a period of one fifth of the bit period, i.e. 20 msec. A second pair of tones is then selected and the state of the message bit is transmitted on the appropriate one of these two tones for a further one fifth of the bit period. Thus five tones are transmitted sequentially from the selected ten tones during a single bit period. This process is then repeated for the next message bit.

In the prior art receiver, the signal is first digitised and then band pass filtered and detected for each of the ten transmitted tones. Bit synchronisation is then established by means of a detected start of message (S0M) signal. For each message bit, each of the 10 tones is measured at the appropriate 20ms time during the bit period and measurements taken in the 20ms periods immediately before and after the message bit are used as a measure of the noise and hence the quality of the respective tone measurements. The noise and bit measurements for each tone are then combined to give the best estimate of the data bit.

Figure 1 illustrates one arrangement of transmissions as used in the present invention. Ten tones T_i are selected in a 2kHz bandwidth for transmission in the HF band. The tones are spaced at 200 Hz intervals from 600 Hz to 2400 Hz.

Figure 2 shows how the transmitted tone frequencies may be mapped to the bits of a message character. Each message character is transmitted, by example, as a 7 bit ASCII code preceded by a Start Bit (logic 0) and ending with a Stop Bit (logic 1). To combat the effects of HF fading and interference etc. each message bit is transmitted five times during a character period 21. In one mode, a pair of tones, 22 and 23 for example, is selected to transmit the first character bit with one tone 22 representing logic state 1 and the second tone 23 representing logic state 0. The tone appropriate to the first character bit is transmitted for l/5th of the character bit period 21. A second pair of tones at 1000 Hz and 1200 Hz is then selected to transmit the same character bit for a further l/5th of the character bit period 21. In this way five tones are transmitted sequentially during the bit period 21: 22,24,26,28,210 for logic 1 or 23,25,27,29.211 for logic 0. This process is then repeated for the remaining five character bits.

In order to enable a receiver to obtain bit synchronism a 64 bit (for example) semi- orthogonal code having an impulsive autocorrelation function, the Start of Message (SOM) , is transmitted prior to the message text. This 64 bit code is transmitted in entirety as a Frequency Exchange Key (FEK) signal on each of the five tone pairs in sequence. Having transmitted this sequence a fixed three character sequence, the NULL sequence as shown in Figure 3. is transmitted in the message character format, as described above, which allows the receiver to set up various threshold levels. The message text is then transmitted. At the end of the message transmission, a second 64 bit semi-orthogonal code, the End Of Message (EOM) code, is transmitted. At the end of a message transmission a repeat message may optionally be transmitted following a third 64 bit semi-orthogonal code, the ETX code. The SOM transmission is also shown in Figure 3 where the 64 bit SOM code 31 is sequentially transmitted, first on the 600 Hz and 800 Hz tone pair, then the 1000Hz and 1200Hz tone pair and so on. The three characters of the NULL sequence (the first only (32) is shown) are then transmitted on the tone pairs using the ON OFF Key (00K) mode previously described. Instead of sequential transmission on the tone pairs as described the SOM, ETX and EOM signals can use tone pairs randomly selected.

Figure 4 shows a transmitter having a keyboard (4l) and a serial input (42) to provide messages to a Display and Keyboard Processor 3 which controls a display 44. A transmission control 45 receives inputs from a code generator 46 producing the SOM, EOM and ETX codes and from the Display and Keyboard Processor 43- The transmission control 45 provides signals to a tone generator 47 which provides the 10 tones from 600Hz to 2400Hz needed for the FEK semi-orthogonal sequences and for the 00K message transmissions. The output signals from the tone generator 47 are then conditioned in a conditioning unit 48 for HF transmission. Messages, prior to and after transmission, may be stored in the Display and Keyboard Processor 3 and printed as required via an output processor 49.

In the receiver shown in Figure 5 the detected signal, after conversion to an audio baseband, is connected to a conditioning unit 51 where the signal is digitised. The digitised signal is connected to a tone detector block 52 having a bandpass filter and detector appropriate to each of the 10 transmitted tones. Timing information is extracted from the tone detectors by means of a detection synchronisation unit 53 which includes continuously running correlators programmed with the SOM, ETX and EOM sequences. The timing and control information from the detection synchronisation unit 53 are connected to the receiver control 54. The output signals from the 10 tone detectors are also connected to a demodulator algorithm unit 55 which synchronously processes the received message under the control of the control unit 5 . The code, corresponding to the transmitter code, for use by the demodulator algorithm unit 55. is generated by a code generator 56- Demodulated messages are connected via the control unit 54 to an output processor 57 for printing or to a receiver Display and Keyboard Processor 58 for display (59). A keyboard 510 is connected to the Display and Keyboard Processor for providing message signals to the printer and to the control unit 54.

The demodulator algorithm unit 55 is used for the following functions: a) assessment of the relative signal level on each tone frequency using the known start and stop bit intervals and an assessment of the noise on each tone frequency during the expected quiet periods;

b) calculation of quality factors for each tone channel and combination of each of the tone channel data outputs having regard to the quality factors;

c) storage of overall bit quality indication for use in bit comparisons where message repeat is used; and

d) timing correction as required, depending on the stability of an internal reference clock.

Text is transmitted following a 64 bit SOM and NULL preamble or, in the case of a repeat message, immediately following a 64 bit ETX sequence. Each text character is transmitted, as shown in Figure 2 as a 9 bit unit with a logic 0 Start Bit followed by 7 ASCII coded data bits and a logic 1 stop bit.

Figure 6 shows the various stages of the demodulation process. For convenience the Figure shows the processing of a single tone frequency Fi (the 600Hz tone) and the combination of all the processed tone frequencies Fi to Fio.

The output 60 from the Fi tone detector is connected to a delay line 61 which can hold 1 second of the Fi signal. Thus at a transmission rate of 10 bits per second 10 bits are held and this therefore can include a complete character and its associated start and stop bits. The signal in the delay line 61 of a logic 1 tone (such as the 600Hz illustrated in this example) will consist of a number of positive pulse-like signals corresponding to 20msec tone transmissions representing the logic 1 levels in the character and also in the stop bit. These pulse-like signals will occur in any of the 100msec time intervals according to the specific bit pattern of the character. Although a given tone is trans¬ mitted for 20msec at the transmitter the tone detector filtering and multipath propagation effects broaden and delay the detected tone pulse and smear it into the next 20msec period. After this next 20msec period there will be three further 20msec periods before the next bit time when the tone may be transmitted again. These three periods, where the tone detector receives general noise and other channel interference, are used to estimate the noise of the particular tone channel. In a 1 second period at 10 bits/sec, the delay line 61 will contain 30 20msec time slot samples that can be used to estimate the tone channel noise for the Fi tone and 30 samples for each of the other tones F2 to Fio. The 30 noise samples available at any time in the delay line 61 for a particular frequency are combined in a channel noise combiner unit 62 where the 30 noise samples are collected symmetrically in time before and after the current character bit under consideration. As shown in Figure 6 this combination of noise samples is used to give a noise estimate Ni for the Fi tone. This process, repeated for each of the noise estimates N1..N10 for each channel, requires that bit synchronisation has been achieved and hence that the position in time of a given tone transmission is known, and also that it is known which tones represent logic level 1 and which logic level 0. The 10 noise estimates N1..N10 are connected to respective inputs of a comparator 63 which provides as an output signal, Nm, the minimum of the noise estimates N1..N10.

Given that the current tone channel, Fi (say), represents logic 1 then the last bit of the 9 bit character sequence is known to be a logic 1 (the stop bit). Hence the measured signal amplitude in the current channel in that 20msec time slot can be used as a reference for that channel for the character period, i.e. for the 9 bits. Thus for each bit interval within a character sequence the signal Si is measured and also a reference signal Ri for the character. As shown the signals Si and reference Ri are provided at the output 64 from the delay line 6l and are switched (65) such that the signals Si are connected to the positive input of a subtractor 66 and the reference signal Ri is simultaneously connected to a divider 67 and a threshold signal circuit 68. The tone noise output signal Ni from the combiner 62 is connected to respective second inputs to the divider 67 and the threshold signal circuit 68. The signal at the output of the divider 67 is equal to R1/N1 (the ratio of the reference signal to the tone noise estimate) and the output signal from the threshold signal circuit 68 is a threshold signal Ti given by:

This is illustrated in Figure 7 where the measured tone reference signal above the average tone background noise is (R1-N1). The tone threshold is selected to be at a level above Ni which is a pre-selected factor K multiplied by half the signal amplitude above the background. K would normally be within the range 0.5 to 1 so as to produce a bi-polar signal, positive for logic 1 and negative for logic 0.

The threshold signal Ti is connected to the negative input of the sub- tractor 66 so that the difference signal (Si-Ti) is connected to one input of a second divider 69■ The tone noise estimate Ni from the com¬ biner 62 is connected to the second input of the divider 69 to provide an output signal (Sι-Tι)/Nι which is equivalent to the signal to noise ratio for the measured bit. This signal to noise output signal from divider 69 is multiplied (610) by the output signal Ri/Ni (representing the signal bit quality) from the first divider 67 to give a product equal to (Sι-Tι)Rι/Nι². Finally this product is multiplied (611) by a factor Nm/Ni (the minimum tone noise to the current tone noise) obtained from a third divider 612 and representing the relative tone quality.

The output signal Ql from the multiplier 611 is thus given by:

Qi = [Sι-Tι]/Nι x [R./N.] x [N./N

- [SNR] x [bit quality]x[channel quality]

Where SNR is the signal to noise ratio and Qi is a signal estimate for the first tone Fi and can take negative as well as positive values. The signal estimates Q1...Q10 for all ten tones F1...F10 are added in adder 612 with due regard to the polarity of the Q values to give an output signal Z, called here the bit quality assessment. The sign of Z gives the bit value (positive is logic 1 and negative is logic 0) and the magnitude of Z gives a measure of the confidence in the bit value. There will be ten values Q....Q₁₀ for each bit period of 100msec. Since these are formed at different times, appropriate delays must be provided before summing in the adder 612. The Z values are stored such that they may be used in a repeat message mode: for one repeat transmission the higher of the two Z values is used for each bit of the message.

Although the invention has been described in relation to a system with five frequency pairs and a transmission rate of 50 bauds (10 bits per second) it will be apparent to those skilled in the art that the invention can be applied to systems with differing numbers of transmission frequencies and at different transmission rates. The frequencies chosen for the system can be based on prediction i.e. those frequencies most likely to provide high availability and reliability.

In a modem according to the invention, arrival at the receiver of more than one transmitted signal due to multipath transmission can have a constructive effect, unlike its adverse effect on other HF systems. This is because there is a high probability that in at least one channel the multipath with add constructively to the zeroth mode, thus causing an increase in signal to noise in that channel. The decoding algorithm detects this and gives that channel more significance than for the zeroth mode alone and also switches off those channels in which the multipath modes have added destructively.

Applications of the invention particularly include battery operated equipment where only low power is available for communications. Examples include: a) expendable buoys; b) remote sensor communications; c) man-pack radios; d) radios for helicopters and fixed wing aircraft; and e) low probability of intercept (LPI) communications.

The invention may also be used in HF communications in general as it improves reliability and availability of the communications medium. It does not require the use of error detection and correction (EDAC) which necessitates parity redundancy. Even when used in a repeat mode, the first transmission can be stored and displayed immediately for the operator before repeat processing to improve the message quality. By re-transmitting a message, any received errors can be corrected by making use of the quality values Z. In this repeat process, each received repeat data bit is compared with the original data bit and the quality values are summed to give an aggregate value which improves the assignment of bit polarity. Under severe propagation conditions the message could be re-transmitted more than once to successively improve the received message quality. Trials have shown that modems according to the invention out-perform Morse signalling under all conditions.

In the arrangement described thus far each signal bit is transmitted sequentially five times on one set of 5 frequencies or another in dependence on the bit polarity. In this arrangement a requirement for 1 watt power transmission per tone (for example) would be met by a 1 watt transmitter. The duty cycle for each channel tone, however, is just 1 in 5 and providing there is no conflict with the transmitter power this duty cycle could be increased to 1 in 2, remembering that a guard period must be retained. Also it may be desirable to time multiplex the (ⁱⁿ the example) signal bit transmissions and the channel frequencies so that burst interference and fading induced errors may be avoided by spreading the 5 transmissions for one bit over several bit periods. The channel frequencies assigned to the 5 transmissions may also be scrambled.

Increasing the duty cycle from 1 in 5 to 1 in 2 increases the data bit rate in each tone channel and reducing the number of tone transmissions for any data bit to Q (= P/n where n>l) with simultaneous channel transmission leads to an increased data rate. In these arrangements, the data rate can be increased by dispensing with the start and stop reference bits for measurement of the channel reference signals R_i and instead making use of measurements of known data bit transmissions as determined by the decoding algorithm i.e. a recursive technique as previously discussed. The initial transmission of the known characters of the Null sequence is used to establish the first channel reference signals and thresholds and thereafter the recursive method is used to sample known channel signals for the reference signals R_j as well as to sample known non-transmission periods for the noise measurements Ni .

The use of a recursive method for measuring channel noise and reference signals is illustrated in Figures 8 and 9- In Figure 8 there is shown a 1 in 3 duty cycle channel tone signal 80 (polarity 1) which is provided as one of the channel tone inputs 90 to a decoder detector 9 as seen in Figure 9- The channel tone signals are successively stored in channel detector unit 92 and the decoder detector 91 measures and stores the reference and noise signal levels for each channel tone. From the determined quality assessment totals Z as previously described the receiver is able to assign a data stream 1,1,0,1,0,0,1 (as shown) at the output 93 of the decode detector 91-

These decoded data values 1,1,0,1,0,0,1 are shown at 81 above the received signal 80. Also indicated along the line 82 are the data bit time slots D interspersed with two guard slots G. and G₂. Knowing the positions of the active transmissions as indicated by D. ie the positions of the "l"s which are fed back to the channel detector unit 92, enables the channel tone reference signal R_£ to be measured at each of these transmissions. The noise Ni is determined from measurements in the G₂ time slots (the G. time slot is not used because of possible multipath corruption) and also the D_Q time slots where there is a data '0' and hence no '1' transmission at this frequency. The N_f values for each channel tone, as before, are used to determine the mean noise N_m, and the values of Nm,*Nl. and Rl. are used to set the channel thresholds and produce the channel quality values for the signal measurements Si . This channel weighting is indicated by the unit 94. Knowledge of each transmitted bit enables measurements to be made on the known received bits in place of the start and stop bits of the first example of the invention.

Figures 10-12 show alternative transmitted waveforms for achieving higher data rates than the 10 bps of Figure 1,2 arrangement. Each of these arrangements uses a baud rate of 100 i.e. a bit length of 10msec and has 12 tones spaced at 200Hz intervals from 600Hz to 2800Hz with a maximum of 3 tones transmitted per bit period. In Figure 10 a data "1" is indicated by 6 10msec transmissions on tone frequencies 1-6 and a data "0" by 6 10msec transmissions on tone frequencies 7-12. The duty cycle in each tone channel is 1 in 2 so that any two consecutive 10msec data transmission periods are separated by one 10msec guard time period to overcome the multipath interference problem on reception. The 6 "1" tones are divided into two groups of 3 with the first group of frequencies 1-3 transmitted in the first 10msec time period 100 and the second group of frequencies 4-6 transmitted in the second 10msec time period 101 corresponding to the guard time period for the frequencies 1-3. For clarity the "0" frequency time channels 7~12 are shown slipped 20msec in time (to the right) as would result from transmission of 1,0 (i.e. "1" followed by "0"). These 6 "0" frequency tone channels are also divided in similar manner into 2 groups with tones 7^_9 being transmitted together in a first 10msec time period 102 and the tones 10-12 being transmitted together in the second time period 103- The data bit period, ie the total time to transmit a bit, in this arrangement is thus 20msec and the data rate is 50 bps.

In the 4 time periods shown, 100-103, as for the transmission of 1,0, noise samples can be taken and used for channel noise assessment in periods 102,103 for tones 1-3 since 101 is a guard time period and we will have been able to deduce from operation of the data decoder detector in Figure 9 that the time slot 102 is inactive because a "0" was being transmitted. In similar manner, noise samples can be used from 103,104 in tones 4-6, 100,101 in tones 7-9, and 101,102 in tones 10-12. In this arrangement in a first 10msec time period tones 1-3 are transmitted for a "1" or 7"9 for a "0" and in a second consecutive time period tones 4-6 are transmitted for a "1" or tones 10-12 for a "0". Figure 11 shows a second transmission format, again with 12 tone channels, where the data rate is 100 bps. In this arrangement 3 tones are transmitted simultaneously to represent a data bit but the frequency redundancy is halved. The 12 tone channels are divided into 2 sets and each set into 2 groups. One set transmits an A data bit, a "1" or "0", while the second set transmits a B data bit. Each group of the set has 3 frequencies. Considering the A set: as before, transmission of 1,0 is represented by transmissions in a 1st 10msec time period 110 on tones 1-3 followed by transmissions in a 3rd 10msec time slot 112 on tones 4-6. The B set transmissions occur during the guard time periods 111,113 of the A set "1" and "0" transmissions. Thus as shown for a B set 1,0 data transmission, a B set "1" is represented by 3 transmissions on tones 7-9 in the 10msec time period 111 and the "0" is represented by 3 transmissions on tones 10-12 in the 10msec time period 113•

In the Figure 12 arrangement the frequency redundancy is reduced to 2. The 12 frequency tones are divided into 3 sets A-C of 4 channels each where 2 channels of each set represent a "1" and the others represent a "0". Three data bits are transmitted in the respective three channel sets A-C and the transmissions are as shown for a 1,0 in each of the three sets A-C. In the set A a "1" is represented by transmissions in successive 10msec time periods 120,121 on tones 1 then 2, while the "0" is represented by successive transmissions in time periods 122,123 on tones 3 then 4. The time periods for sets B and C are as stated for set A with the frequencies 1-4 replaced respectively by 5"8 and 9~12. Thus in time period 120 for transmission of 1,0 in each of the 3 sets, three frequencies are transmitted on tones 1,5 and 9- In this arrangement 3 data bits are transmitted every 20msec giving a data rate of 150 bps. In the arrangements shown in Figures 11 and 12 the considerations applying to channel noise sampling are the same as discussed with respect to the Figure 10 arrangement. Where three tones, maximum, are transmitted simultaneously the transmitter needs to be capable of handling more than the mean power.

It has been found that a significant improvement in message decoding occurs via the repeat message mode, through the addition of Z quality assessment values for each bit of a message. Advantage can be taken of this, for example, in the Figure 12 arrangement by transmitting the message three times with each bit being transmitted via the A,B and C tone sets. The quality assessment for a message bit is then taken as the sum of the three Z values for each sets.

Use of the measurements during active signal transmissions to derive the channel reference values R_χ and during inactive periods to measure the noise N. means that measurements are made every 10msec. This leads to a high update rate on channel information and noise. As in the first 10bps modem example the modem can be arranged to store 1 seconds worth of signal samples for analysis of signal and noise. In all cases, the signal can be analysed during active transmissions (as determined by the soft decision decoder) or additionally making use of start/stop bits) and the noise measurements by analysing any of the signals, except where there is active transmission or in the time period immediately following an active transmission.

Claims

1. A communications system for operation in a band of the high frequency (HF) spectrum comprising transmitter and receiver:

wherein the transmitter comprises: a number of tone frequency channels equal to 2P, where P is an integer, within the HF band; the transmitter being arranged to transmit each character of a digital message as a coded series of bits; the transmitter being further arranged such that each bit is transmitted a number of times Q where Q=P/n with Q an integer greater than 1 and n an integer taking the value 1 or 2 or 3 etc; the tone frequency channels being arranged in P pairs with a transmission on one tone frequency channel of each pair representing a logic state "1" and on the other tone frequency channel representing a logic state "0" and the transmissions on each tone frequency channel being arranged such that there is a guard period between any two consecutive bit transmissions;

and the receiver comprises: means to synchronise the receiver to a received transmission; means to periodically measure a known reference tone signal appropriate to each of the 2P tone frequency channels so as to determine for each tone frequency channel a reference signal R (for i = 1 to 2P) indicative of the channel signal amplitude appropriate to a received bit; means to periodically determine a noise estimate N_i for each tone frequency channel and appropriate to each received message bit in that tone frequency channel from measurements before and after the expected time of the received message bit; means to assign a signal threshold T. to each tone frequency channel and appropriate to each received message bit from the measurements R_i and N_i5 means to measure the signal amplitude S. of the message bit above the threshold T_i ; and means to assign a relative quality value to each tone frequency channel i and for each message bit based on the noise estimate N. in that channel and the minimum N of the measured noise estimates N.1 to N2p;' the arrangement being that for each of the 2P tone frequency channel a signal estimate is determined from the measured bit signal to noise, the channel signal to noise and the channel relative quality.

2. A HF communications system as claimed in claim 1 wherein the known reference tone bits are start and stop bits of opposite polarity transmitted respectively before and after each coded character.

3. A HF communications system as claimed in claim 1 wherein there is provided a computing means programmed with a recursive method for determining the reference tone bit, the receiver being arranged such that the signal estimates of bit polarities are used to determine when active known bit transmissions occur and a measurement of a known bit is taken as the reference signal.

4. A HF communications system as claimed in any one preceding claim wherein the transmitter is arranged such that the bit repetitions Q are made singly or in equal groups of tone frequency channels for transmission in consecutive time intervals.

5. A HF communications system as claimed in claim 4 wherein P is selected to be divisible by 2 and each bit is transmitted twice, in one respective group of P/2 tone frequency channels to represent a logic "1" and in the other respective group of P/2 tone frequency channels to represent a logic "0".

6. A HF communications system as claimed in any one of claims 1 to 5 wherein the Q transmissions for each bit are multiplexed with other bit transmissions for time diversity.

7. A HF communications system as claimed in any one preceding claim wherein the Q frequency transmissions use different preselected frequency sets for each bit transmission.

8. A HF communications system as claimed in any one preceding claim wherein the channel relative quality value is the quotient N_m/N_i ; the measured bit signal to noise is (S.-Tj/N_j and the channel signal to noise is R./N..

10. A HF communications system as claimed in claim 2 wherein the channel signal R is measured just once for each character (7 message bits) while the noise N. and signal bit S_j are measured in each bit time interval.

11. A HF communications system as claimed in claim 10 wherein the noise N. is taken as the arithmetic mean of measurements in adjacent channel bit periods when there is no bit transmission.

12. A HF communications system as claimed in any one of claims 1 to 10 wherein noise measurements N_d are taken both during guard periods between successive bit transmissions and also during bit periods when the signal estimate shows there is no bit transmission.

13. A HF communications system as claimed in claim 12 wherein the noise measurements are not made in the baud period immediately following an active data bit where multipath interference is likely to occur.

14. A HF communications system as claimed in claim 1 wherein every known active data bit as predicted by the signal estimate is used for the reference signals.

15. A HF communications system as claimed in any one preceding claim wherein the bit signal to noise is made bipolar by subtracting from each measured bit signal, S_ι , in a character an appropriate value of the threshold T for that character.

16. A HF communications system as claimed in claim 1 wherein the channel signal estimate Q_j is then given by:

the arrangement being such that the signal estimate Q^ is calculated for each of the 2P channel tone frequencies and appropriate ones of the 2P values Q. to Q₂ are then summed to give a quality assessment total Z, the sign of which gives the bit value (positive corresponding to logic "1" and negative to logic "0") and the magnitude gives a confidence measure for the bit value.

17. A HF communications system as claimed in claim 16 wherein the values of Q. for each message bit are stored such that on re-transmission of the message the stored values can be added to the new values for the re-transmitted message and the resultant values used for assignment of the bit polarity in each case.

18. A HF communications system as claimed in any one preceding claim wherein the 2P tones are equi-spaced in frequency within a 3kHz bandwidth.

19. A HF communications system as claimed in claim 18 wherein the transmissions are arranged to fall within a standard 3kHz HF receiver radio channel.

20. A HF communications system as claimed in any one preceding claim wherein the transmitter has 10 tone frequencies (P = 5) separated by 200 Hz with a bit rate of transmission of 10 bits per sec.

21. A HF communications system as claimed in claim 20 wherein each message bit is transmitted as a 20 msec pulse consecutively on 5 different tones in the bit period of 100msec.

22. A HF communications system as claimed in any one of claims 1 to 19 wherein the 2P tone frequency channels are arranged for transmission of 2 or more data bits with groups of 2 or more of the tone frequency channels being available for simultaneous transmission.

23. A HF communications system as claimed in claim 22 wherein there are 12 tone frequency channels arranged for simultaneous transmission on 3 tones.

24. A HF communications system as claimed in any one preceding claim wherein the receiver includes 2P band-pass filters each with a centre frequency corresponding to a different one of the transmitted tone frequencies and a pass-band which is as narrow as possible (taking into account Doppler requirements) .

25. A HF communications system as claimed in claim 24 wherein each filter has a stop band rejection of at least 70dB for frequencies more than the baud rate removed from the centre frequency.