WO1994023488A1

WO1994023488A1 - Snubber

Info

Publication number: WO1994023488A1
Application number: PCT/AU1994/000157
Authority: WO
Inventors: Nigel Charles Machin
Original assignee: Rectifier Technologies Pacific Pty. Ltd.
Priority date: 1993-04-06
Filing date: 1994-03-31
Publication date: 1994-10-13
Also published as: CN1120875A; CN1035972C

Abstract

A substantially non-dissipative current snubber and a substantially non-dissipative current and voltage snubber for a power converter. In one form of current snubber the circuit comprises a switch means (SWB), a diode (DB) which exhibits reverse recovery and a means for controlling the rate of rise of current (L1) with the improvement comprising connecting the controlling means (L1) to an energy storage means (CB out). The circuit is of particular application in switched mode power converters.

Description

SNUBBER FIELD OF INVENTION

The present invention relates to the field of power converters in general, snubbers therefore. The present invention relates, in particular but not exclusively to switched mode power converters, and has equal application in and to other forms of power converters. In one form, the present invention provides a substantially non-dissipative current snubber or a substantially non- dissipative current and voltage snubber. BACKGROUND In the field of switched mode power converters, pulse width modulation

(PWM) circuits are often used. A voltage or current is controlled by varying the pulse width (or duty cycle) of one or more switches which usually operate at a constant frequency.

Other converters are the so-called Resonant circuits. These have a large resonant tank and vary the frequency of excitation of that tank to regulate voltage or current. There are also Quasi-Resonant converters which have small resonant networks which provide a current or voltage limiting action. Regulation is usually achieved by frequency modulation (FM).

Yet another class of converters are the so called Soft-Switched converters. These types use a multiple of switches (2 or more) and small resonant networks to achieve a current or voltage limiting action while providing a constant frequency PWM-like control.

These other classes of converters have been developed to alleviate some of the problems of PWM, but they still exhibit a number of drawbacks. Notably, at low to medium switching frequencies, efficiency is lower than that of equivalent PWM circuits. PWM circuits and some circuits from other classes exhibit two key problems.

Problem 1 : Diode reverse recovery in switched mode power converters causes relatively high peak current and power stress in the switching device during turn-on of the switch, which results in a relatively high average power loss in the switching device. Significant EMI is also generated during turn-on, and there is considered to be a resultant low overall efficiency. Figure 1 shows one type of prior art snubber, being a dissipative current snubber in a boost circuit. The components of the snubber are drawn with bold lines and the path of the reverse recovery current is shown with an arrow. The snubber serves to alleviate most of the aspects of the problems noted above, but transfers the energy stored by the snubber inductor to a snubber resistor. This results in a power loss, and thus there is still a low overall efficiency. In practice, the power loss in the snubber resistor is comparable to the reduction in power loss gained by the switching device, so the overall efficiency of the switched mode converter is not improved. Figure 2 shows another type of prior art snubber, being a non-dissipative current snubber. The snubber shown does alleviate aspects of the problem noted above, but creates yet another problem. The overall efficiency is improved by returning the energy stored by the snubber inductor back to the power supply. However, in so doing, it creates a relatively high peak voltage stress for the switching device. This is due to the leakage inductance added by this snubber between the switch and the clamping voltage.

Problem 2: Inductive load present while turning off the switching device causes relatively high peak power stress in the switching device and a consequential high average power loss in the switching device, which leads to an overall low efficiency.

The prior art fails to provide a current snubber that solves this problem.

Yet another problem associated with the prior art is that the prior art also fails to provide a combined current and voltage snubber which has a relatively small number of components. OBJECT OF INVENTION

An object of the present invention is to alleviate at least one problem associated with the prior art.

Another object of the present invention is to provide a current snubber and / or a current and voltage snubber for use with power converters. SUMMARY OF INVENTION

The present invention is predicated on the principle of returning energy, stored in component(s) during a switching action of a switching means in a power converter, to at least one energy storage element associated with the power converter. The energy storage element may be in circuit, at an input and/or at an output of the power converter, or combination thereof.

The present invention contemplates, in one form, that the energy from or in the component(s) is returned by substantially direct connection between the component(s) and the energy storage element(s) of the power converter.

The energy storage element(s) may be any form of energy storage element. The connection is preferably by connecting a conduction element(s) adapted to conduct in at least one direction between the component(s) and storage element(s). A diode or MOSFET is most preferred as the conduction element(s) although they are not to be considered as the only type of conduction elements.

The present invention results from the realisation that component(s) or a snubber can preferably be incorporated in, or added to the normal componentry of, a power converter to control, including limiting or reducing, the rate of rise of current and/or voltage during switching of the switch means and which stores energy in the component(s) for return to energy storage elements. The return of energy by direct connection alleviates high peak voltage stress in the switch element as the passage of the stored energy to the energy storage element by way of the direct connection serves to clamp, remove, reduce or substantially eliminate an additional voltage which would otherwise cause the high peak voltage stress.

It is preferred for the control component or snubber to be placed in series connection with or directly connected to the power converter component exhibiting reverse recovery. This component of the power converter may be a diode.

The present invention advantageously incorporates one or more components which may be inductor, capacitor, diode, resistor, or combination thereof in a power converter which serve to substantially limit or reduce the rate of rise of current through the switch means during turn-on and / or which limit or reduce the rate of rise of voltage across the switch means during turn-off. This limiting action is achieved substantially without the necessary dissipation of the energy stored in the limiting elements as occurs in the prior art, but rather the stored energy is returned by connection to energy storage means associated with the converter. The return of some or substantially all the stored energy enables a power converter incorporating the present invention to have improved efficiency. The elements which form the snubber and / or the snubber itself can be considered substantially non-dissipative or lossless due to the return of stored energy to the storage element(s).

The present invention provides in a power converter including, in a loop, a switch means, a diode which exhibits reverse recovery, and a means for controlling the rate of rise of current, the improvement comprising connecting the controlling means to an energy storage means.

The connection can be considered, in one form, direct and provided by a diode or other suitable element. The provision of the yet further feature of a combination current and voltage snubber may be embodied by the provision of at least one extra capacitive element to the embodiment of the current (only) snubber as disclosed above.

The additional capacitive means is used in voltage snubbing. A capacitor is added to achieve control or a limiting action over the rate of rise of voltage across the converter switch means. Both the current and the current and voltage snubbers utilise the relative direct connection feature disclosed above. Additional control or limiting of the rates of rise of voltage across the switch means may be delivered by the capacitive element . The energy stored in the capacitive element is returned using the direct connection of the current (only) snubber. The combination of two substantially lossless snubbers into the one circuit addresses the problems raised above and uses less components than prior art solutions. A second embodiment of the present invention illustrates the current and voltage combination snubber. Resultant from the direct connection feature, a relatively reduced or limited peak voltage stress as compared to non-dissipative prior art current snubber arrangements, and thus lower voltage and / or more efficient switching devices can be employed in the circuit arrangements of the present invention. There may also be a cost saving achieved due to the reduced number of components used in the circuits according to the present invention.

The present invention also provides an improved current snubber circuit for controlling the rate of change of current during diode reverse recovery in a power converter, comprising : an inductive element inserted in series with an existing diode in the power converter which is subject to reverse recovery, in such a manner that the rate of change of diode current during reverse recovery is limited by the inductive element with a direct connection made between one terminal of the existing diode and one terminal of the inductive element; two series connected diodes themselves connected between the outer terminals of the series connection of the inductive element and existing diode so as to form an alternative or parallel current path with like polarity; and a capacitive element connected between the junction of the two series connected diodes and the junction of the inductive element and existing diode, in order to receive energy from the inductive element via the interconnecting diode after the completion of reverse recovery and in order to return that energy to the power converter via both diodes before the existing diode again conducts. Preferred embodiments of the present invention are now described with reference to the accompanying drawings.

In each drawing the components of the snubber are drawn with bold lines and the path of the reverse recovery current is shown with an arrow.

Figures 1 and 2 show prior art circuit arrangements as applied to a boost converter;

Figure 3 illustrates a boost circuit with a substantially lossless current snubber;

Figure 4 illustrates timing waveforms for the circuit of Figure 3;

Figure 5a illustrates a boost circuit incorporating a substantially lossless current and voltage snubber in accordance with a second embodiment. The embodiment essentially uses an additional capacitance to give effect to a voltage snubbing action further to the snubber of Figure 3; Figure 5b illustrates a boost circuit incorporating a substantially lossless current and voltage snubber of Figure 5a with an additional diode inserted to prevent continuous ringing occurring during part of the switching cycle;

Figure 5c illustrates a boost circuit incorporating a substantially lossless current and voltage snubber of Figure 5b with additional diodes to decrease conduction loss in the snubber;

Figure 6 illustrates waveforms associated with the circuit of figure 5a;

Figure 7 illustrates a boost circuit with another substantially lossless current and voltage snubber. Several other components are added to improve the performance of the circuit illustrated as compared to Figure 5b under different and non-ideal operating conditions; and

Figures 8 to 15 show embodiments of the present invention as applied to other forms of power converters, without limitation.

To simplify the explanation of the operation of the circuit, the forward voltage drop of diodes as well as of the power switch SWB has been assumed to be negligible. In addition, second and third order effects, such as the ringing which occurs when diodes stop conducting, have been left out of the operating waveforms shown in Figs. 4 and 6. The switch may be any type of contact or electronic switch, for example a transistor, MOSFET, IGBT, or other combination of switches.

Referring to Figure 3, it can be seen that inductor LB, Diode DB, Switch

SWB and capacitors CBIN and CBOUT are the elements which normally comprise a boost power converter. In one form of this invention, inductor Li is connected in series with SWB and determines the rate of current increase through SWB at turn-on (To) where: d ISWB(TO) _ VOUT dt Li _{(1 )}

Since the voltage across SWB falls rapidly to zero while the current in SWB is low, the power loss during turn-on of SWB is small. Here it is assumed that at switch-on of SWB, current IIN is flowing through

DB to the load. As current In builds up through Li, it correspondingly decreases through DB , eventually becomes zero at Ti and then further decreases due to the reverse recovery of DB to a negative value IR at time T2 as shown in Figure 4.

At this point in time, DB reverts to a high impedance state (recovers) and voltages V1 and V2 fall as shown in Figure 4.

Since the current through Li is now IB plus IIN, and the current through LB is still IIN (assuming LB » Li) the excess current through Li, IR, begins to flow in the loop formed by Dι, C1 and Li and resonantly reduces to substantially zero as shown in Figure 4 between T2 and T3.

In this period the excess energy stored in Li, EXLI(T2) is thus transferred to C1 where

and E_C1(T3) = -C₁V₁(T3)² (3)

hence V_I(T3) = /^^» (2II IR+ IR²) _≡ VE (⁴)

Eci(T3) is energy stored in C1 at T3 and VE is the voltage on C1 at T3.

The above equations do not take into account the finite energy loss in Li, D1 and Cι, so in practice VE will be smaller than given by (4).

When switch-off occurs (T4), the initial rate of rise of voltage across SWB is limited by its own rate of decrease of current in combination with the total capacitance of the switch and other components and stray capacitance.

When voltage V2 attains a value VOUT, D2 will conduct the current IIN. Initially IIN flows through L1 and D1 into D2, but since there is a voltage -VE on Li at this point in time, the current through Li will begin to decrease. The difference in current between In and IIN will flow through Cι, decreasing the voltage on C1 until at T5, substantially all the energy in Li has transferred to C1 so that the current through C1 is now IIN , that through D1 is substantially zero and the voltage on Ci is substantially VEL The constant current in Ci then discharges it at a constant rate given by:

d Vi d t Ci

(5)

When Ci is substantially completely discharged (Tβ) DB turns on and the snubber is ready for a new cycle.

In this embodiment of the invention, it can been seen that due to the provision of a substantially direct connection between the switch SWB and energy storage element CB, that:

1. the maximum voltage that appears across SWB is substantially equal to VOUT,

2. the maximum voltage that appears across DB is substantially VOUT + VE,

3. the maximum voltage across D2 and D1 is substantially VOUT,

4. the turn-on switching loss of SWB is low since the voltage across SWB falls rapidly to substantially zero before the current through it has risen to its full value, and

5. the snappiness and peak reverse recovery current of the diode DB is substantially reduced due to the low and controlled dl/dt.

Figure 5a shows another form of the invention in which a capacitor C2 is added to the circuit of Figure 3 to achieve a controlled rate of rise of voltage across the power switch SWB at turn-off.

Reference is made to Figure 6 which shows various waveforms at different phases of a switching cycle. The starting point is with current IIN flowing through DB to the load. The current through Li , Dι , D2 and SWB is substantially zero as is the voltage across Ci . The voltage V2 on C2 is approximately equal to VOUT. It is further assumed that Ci » C2; e.g. Ci = 10 C2.

When SWB turns on at To, the current through it will increase from zero at a controlled or limited rate given by: d ILI(Ε)) _ VOUT d t Li ₍₆₎

Since the voltage across SWB falls rapidly to zero, its turn on power loss is low.

The current through DB will linearly fall to substantially zero at time Tι and continue to decrease until reverse recovery current IR flows at T2.

Voltages V1 and V2 begin to fall after DB turns off at T2, thus beginning the discharge of C2 towards substantially zero volts.

The current causing C2 to discharge is substantially equal to the difference between IIN and In which substantially equals the reverse recovery current IR to begin with, but in the time period T2 to T3, it increases to a substantially higher value due to the increase in current through Li caused by the positive voltage V1 applied to it. At T3, V1 is zero volts but because of the charge attained by Ci in this period, the voltage V2 is positive.

From T3 to T4 the voltage on Li becomes negative, so the current through it begins to decrease. However, the net current flowing through C2 and Ci is still equal to the difference between IIN and III

At T4 the voltage on C2 (V2) is substantially just less than zero and D1 begins to conduct, thus clamping V2 to approximately zero volts (ignoring SWB forward voltage and D1 voltage drop).

Between T4 and Ts, a resonant 1/₄ cycle ring occurs during which the excess energy stored in Li due to both the reverse recovery of DB and the discharge of C2, is transferred to Ci so that the voltage V1 is given by:

By inspection, it can be seen that even if IR = 0, VE (Vi(T5)) will still have a positive and finite value, ensuring a "resetting" of Li during the turn-off phase. At Tδ the diode Di turns off. The presence of capacitor C2 prevents the voltage V1 from immediately resetting to zero at this point, as occurs in the current (only) snubber, and instead a continuous ringing occurs between the components Li, Ci and C2. This ring is small in amplitude but does have an effect on the operation of the snubber. An improvement to the snubber in order to remove this ringing is suggested in Figure 5b.

At time T6 the switch SWB is turned off. At this time the voltage V2 will be within the range of OV to twice VE, depending on the part of the resonant cycle in which turn-off occurs. Assuming that Cι»C2 then most of the resonating voltage will appear across Ci and the voltage across C2 will remain at approximately VE. If V2 is at OV at Tβ, as shown in Figure 6, then operation will be as follows:

When SWB is turned off at Tε, the current IB flowing through Li will flow through D1 and C2 thus causing the voltage across SWB to increase from zero at a rate given by:

d VSWB(T6) _ IIN (9) d t C2

Since the current in SWB falls rapidly to substantially zero, its turn-off power loss is low.

At time Tε if V2 is not at OV then the voltage across SWB will not start from zero and consequently the power loss in the switch at turn-off will not be as low. The interval of time from Tε to T7 will become shorter as V2(T6) increases. The extreme case occurs when V2(T6) is twice VE, but it should be noted that since VE is a small fraction of VOUT (assuming Cι»C2), the reduction in performance of the snubber compared to the best case is marginal. At T7, D2 also becomes forward biased and clamps the switch voltage and C2 voltage to VOUT, the output voltage. At this point, current begins to flow in the loop formed by Li, D1 and Ci, and current in C2 drops to zero. The voltage across Li at this time is -VE SO current in it begins to drop from its value IIN. The increasing difference current between IIN and the current through Li flows out of Ci and begins to discharge it as shown in Figure 8 between T7 and Tβ. At Tβ, Di turns off and Li is fully reset with current through it being zero. If the voltage across Ci is still not zero, the current IIN through Ci, D2 will continue the discharge of Ci until at T9 its voltage is zero and DB is thenceforth forward biased and conducts current IIN . At this point Li and Ci are reset and the circuit is ready for a new cycle.

Some of the advantages of this embodiment as provided by the relative direct connection of the junction of Li and SWB , via D1 and D2, and the connection of C2, via D2 to the energy storage element CBOUT are:- 1. The maximum voltage of SWB is substantially VOUT, 2. The maximum reverse voltage on DB the main boost diode, is substantially VOUT + VE and is substantially well defined by the relative values of IR, LI , Ci and C2, 3. The maximum rate of rise of current through SWB is substantially well defined by Li and VOUT and its turn-on power loss and stress is low, 4. The maximum rate of rise of voltage across SWB is substantially well defined by C2 and IIN and the turn-off power loss and stress is low, 5. The energy stored in Li and C2 during the switching cycle in order to achieve control of the rate of rise of current and voltage is substantially returned to the output supply thus enabling a substantially lossless operation.

Another form of the invention is shown in Figure 5b where a diode D3 is added to the circuit of Figure 5a in order to reduce oscillation between the series combination of capacitors Ci and C2 with inductor Li which would otherwise occur after time Tδ of Figure 6. The reduction of this oscillation enables the states of the snubber components to be more accurately predicted at time Tε and thus the operation of the snubber to be more consistent. Power loss in the switch is improved since the voltage on capacitor C2 is always near zero at time Tε. However an additional loss occurs due to the extra conduction loss of D3. Another form of the invention is shown in Figure 5c where two diodes D4 and Dδ are added to the circuit of Figure 5b in order to avoid the additional loss that the voltage drop of diode D3 brings to the circuit of Figure 5a. Either or both diodes may be added, depending on the improvement in efficiency desired.

Another form of the invention is shown in Figure 7 in which several components are added to the circuit shown in Figure 5b in order to improve the behaviour of the invention where it is used in a non-ideal environment.

Rι is added to dampen resonant oscillations which occur at times Tδ and

T7 respectively (Figure 6), R2 and C3 are added to substantially dampen oscillations which occur at time T2 (Figure 6), and Z1 (zener diode(s)) is added to substantially prevent excessive voltage build-up on Ci when current in LB becomes discontinuous at light loads.

Figure 8 shows the current snubber applied to the boost converter in four different arrangements. Each arrangement produces a similar result in terms of current snubbing but places slightly different requirements on the components. For example, the peak current requirement for the snubber choke L1 in (1 ) is greater than in (2) - (4) because it must carry the input current added to the reverse recovery current of the boost diode DB, whereas in (2) - (4) it carries either the input current or the reverse recovery current but not both at once. Other aspects of the different component requirements are left to the interested reader to delve into, however it is worth noting that arrangement (4) is not useful since it produces voltage spikes at the output.

Note however that in each case the switch and diode of the boost converter have their maximum voltages limited by the direct diode connection to the output capacitor of the boost converter.

Figure 9 shows the current snubber applied to the Buck converter, again in four different arrangements. Each arrangement again produces a similar current snubbing result and again places slightly different requirements on the components. However, unlike the application to the Boost converter, all four arrangements are useful.

In each case the switch and diode of the Buck converter have their maximum voltages limited by the direct diode connection to the input capacitor.

Figure 10 shows the current snubber applied to some common converters, namely the Buck-Boost, Cuk, Sepic and Zeta converters. Each of these converters can have four variations of snubber similar to the examples given above, though each schematic shows only one variant. The reverse recovery current path in each example is shown with an arrow.

The current snubbers in these converters make direct diode connection to various energy storage capacitors - the buck-boost converter has direct connection to the output capacitor, the cuk converter has connection to the intermediate capacitor, the sepic and zeta converters connect to the intermediate capacitor and the output and input capacitors respectively.

Figure 11 shows the current snubber applied to some common bi- directional converters. These each have alternate reverse recovery current paths depending on which way power is flowing in the circuit.

Only one of the two current snubbers would normally be active depending on the direction of power flow. The inactive snubber would not transfer energy since the diode that it snubs would not carry current. Figure 12 shows the current snubber applied to some common isolated converters: the Flyback, Forward and isolated Cuk. The leakage inductance of the isolating transformer in each case provides a degree of current snubbing. However, in applications where this is insufficient the current snubber will be useful. The directness of the diode connection is compromised by the leakage inductance of the isolation transformer, but the contribution of the snubber inductor to the voltage stress of the converter switch and diode is minimal due to the direct diode connection around it.

Figure 13 shows the current snubber applied to some isolated converter secondaries. The primaries are not shown but could be half bridge, full bridge or push pull arrangements operating as current fed or voltage fed inverters. As in the previously described isolated converters the leakage inductance of the isolation transformer provides a degree of current snubbing. In applications where this is insufficient the current snubber will be useful. Note that in some arrangements the one current snubber can be used to snub two existing diodes by adding a third diode to the snubber - this is possible because only one of the two existing diodes recovers depending on the polarity of the transformer output.

Figure 14 shows the current and voltage snubber of Figure 5a applied to the boost converter in six different arrangements. In each case the additional capacitor couples across the switching device of the converter, either directly as in (1 ) and (4), through the output capacitor as in (2) and (5), or through the input capacitor as in (3) and (6). As with the current (only) snubber the different arrangements produce a similar result in terms of the current and voltage snubbing function but place slightly different requirements on the components. Figure 15 shows the current and voltage snubber applied to various converters. These are a small selection of converters, each of which shows only one of many possible implementations of the current and voltage snubber of Figure 5a. The additional components of Figures 5b , 5c and 7 can be applied to all implementations of the Figure 5a snubber in each converter. In isolated converters the leakage inductance of the isolation transformer reduces the effectiveness of the voltage snubbing capacitor because it reduces the closeness of the coupling between this capacitor and the switching device(s). The reduction in effectiveness results in a smaller but still useful efficiency improvement compared to that which would occur in a similar non-isolated converter.

Although a number of embodiments have been described, the feature of the present invention as applied to a current snubber or a current and voltage snubber being the connection to an energy storage element is applicable to any power converter.

Claims

THE CLAIMS DEFINING THE INVENTION ARE AS FOLLOWS:

1. In a power converter including, in a loop, a switch means, a diode which exhibits reverse recovery, and a means for controlling the rate of rise of current, the improvement comprising connecting the controlling means to an energy storage means.

2. An improvement as claimed in claim 1 , wherein the controlling means is connected directly to the reverse recovery exhibiting diode.

3. An improvement as claimed in claim 1 or claim 2, wherein the loop is primarily determined by a path of reverse recovery current flow.

4. An improvement as claimed in claim 1 , claim 2 or claim 3, wherein the controlling means is additional to the power converter.

5. An improvement as claimed in any one of claims 1 to 4, wherein the connection is substantially direct.

6. An improvement as claimed in any one of claims 1 to 5, wherein the connection is via diode or MOSFET.

7. An improvement as claimed in any one of claims 1 to 6, wherein the controlling means is an inductive element.

8. An improvement as claimed in any one of claims 1 to 7, wherein the energy storage means is connected to the input of the power converter.

9. An improvement as claimed in any one of claims 1 to 7, wherein the energy storage means is connected to the output of the power converter.

10. An improvement as claimed in any one of claims 1 to 9, wherein at least one diode interconnects the controlling means and the energy storage means.

11. A current snubber including the improvement as claimed in any one of claims 1 to 10.

12. A power converter including a current snubber as claimed in any one of claims 1 to 11.

13. A power converter as claimed in claim 12, being any one of Buck, Boost, Cuk, Buck-Boost, Sepic, Zeta, Forward, Flyback, Isolated Cuk.

14. A voltage and current snubber including the improvement as claimed in any one of claims 1 to 10.

15. An improved current snubber circuit for controlling the rate of change of current during diode reverse recovery in a power converter, comprising : an inductive element inserted in series with an existing diode in the power converter which is subject to reverse recovery, in such a manner that the rate of change of diode current during reverse recovery is limited by the inductive element with a direct connection made between one terminal of the existing diode and one terminal of the inductive element; two series connected diodes themselves connected between the outer terminals of the series connection of the inductive element and existing diode so as to form an alternative or parallel current path with like polarity; a capacitive element connected between the junction of the two series connected diodes and the junction of the inductive element and existing diode, in order to receive energy from the inductive element via the interconnecting diode after the completion of reverse recovery and in order to return that energy to the power converter via both diodes before the existing diode again conducts.

16. A current snubber as claimed in claim 15, wherein an additional capacitor is placed between the junction of the series connected diodes and a point which effectively couples the capacitor across the switch means of the power converter, said capacitor thereby limiting the rate of change of voltage across the switch means of the converter at turn-off, thereby providing an additional voltage snubbing action.

17. A current snubber as claimed in claim 16, wherein an additional diode is placed between the voltage snubbing capacitor and the capacitor of the current snubber in order to alleviate ringing between the two capacitors and the inductive element.

18. The combined current and voltage snubber of claim 17 wherein one or two additional diodes are placed in parallel with one or both series connected pairs of diodes previously described in order to offer lower voltage drops to currents carried by those diodes and thus result in improved efficiency for the circuit.

19. A current snubber as herein disclosed.

20. A current and voltage snubber as herein disclosed.