WO1990013113A1

WO1990013113A1 - Process for positioning a magnetic-disc memory, and device for carrying out the process

Info

Publication number: WO1990013113A1
Application number: PCT/EP1989/001091
Authority: WO
Inventors: Heinz-Dieter Maier; Werner Spengler; Heriberto Unger
Original assignee: Siemens Aktiengesellschaft
Priority date: 1989-04-27
Filing date: 1989-09-20
Publication date: 1990-11-01
Also published as: EP0468962A1

Abstract

The invention concerns a process for positioning a positioning unit (2), in which the current for the positioning unit drive (201) is controlled by continuous comparison of the instantaneous position (xa) of the positioning unit as measured value with the target position (xn) as theoretical value, using a theoretical curve of the distance between start and target track. The theoretical curve is drawn, taking into account the back-emf of the drive unit, as a time-optimal step function. Sample values (ik and xk) corresponding to points on this theoretical curve are fed continuously at predetermined sampling intervals (Δ) to an FIR filter (e.g. 52). Their filter coefficients (c0... cn) are chosen so that the points of inflexion of the theoretical curve are preferably cos-curve-shaped. The sampled values (ink and xnk) filtered in this way are converted into a control signal (is(t)) for the positioned unit drive. This process is carried out in real time, preferably by a signal processor.

Description

Positioning method for a magnetic disk memory and device for carrying out the method

The invention relates to a method for positioning a positioning unit of a magnetic disk memory according to the preamble of claim 1 and to a device for performing this method.

In the case of magnetic disk memories, the average access time is known to be one of the essential performance criteria in addition to the storage capacity. The track density and the average access time, which may include the pure positioning time and the settling time on a selected track, determined the structure of the positioning unit and the positioning method. For example, in the group of powerful 5 1/4 "magnetic disk memories, the conventional positioning methods are similar in that a speed profile dependent on the track distance, ie a specific speed as a function of the distance traveled during the positioning process, is specified as a setpoint curve and the sequence of the positioning process is then regulated An analysis of the positioning process on this basis is described, for example, in "Magnetic Recording", Vol. II, published by CDMee and EDDaniel, Mc Graw Hill Verlag, 1988 in Section 2.3.5. With this setpoint specification in the form of a speed profile, access times are around 20 ms still controllable.

If, on the other hand, you want to significantly reduce the positioning times, ie reach an average of 14 ms and less, you have two options: You can set up the positioning unit so that higher maximum speed and greater acceleration are possible. With conventional positioning methods, however, these possibilities are also limited. One skilled in the art immediately understands that the goal, despite the higher maximum speed of the positioning unit, it is only possible to approach the target track as precisely as with conventional magnetic disk memories, and can only be achieved with a corresponding increase in the "dynamic range" of the speed control. However, conventional standard circuits that were developed for use in the positioning process of magnetic disk memories are no longer suitable. In addition, there is a further limitation that an increase in the permissible acceleration is not readily achievable. Initially, this would mean what seems reasonable in view of a time-optimal solution to the positioning task of applying the current as quickly as possible to accelerate the drive of the positioning unit. However, if one tries to achieve the goal in this way, one will see that sudden changes in the acceleration of the positioning unit directly excite mechanical vibrations, i. H . extend the settling time and z. B. also cause operating noise.

For this purpose, it is already known from international patent application WO 88/02913 to specify an acceleration profile for the positioning unit in spite of the goal of time-optimal positioning instead of a speed profile in such a way that the acceleration or deceleration preferably changes only linearly.

In this known positioning method, three groups of positioning processes are distinguished depending on the track distance that occurs. In the case of positioning operations below a certain track distance, the positioning unit does not yet reach the maximum predetermined acceleration. In this case, the positioning unit is linearly accelerated in the first quarter of the positioning time, then the change in the acceleration reverses its direction, so that with three quarters of the positioning time the greatest deceleration occurs, which then occurs near drops again until the end of the positioning process. The course of acceleration over time therefore corresponds to two triangles which are symmetrical to the time axis and are placed directly next to one another.

In a second group of positioning processes over medium track distances, the maximum acceleration or deceleration has not yet been reached, but the maximum speed of the positioning unit has been reached. The acceleration profile therefore also contains sections with constant acceleration or deceleration and corresponds to two trapezoids that are symmetrical with respect to one another with respect to the time axis and are placed directly next to one another. In the third group of positioning operations over large

Track distances are finally reached in the meantime, the limit speed of the positioning unit. The acceleration profile then corresponds to two trapezoids lying symmetrically to the time axis, which are spaced apart from one another in time.

In all of these cases, the acceleration component of the overall profile with the opposite sign is identical to the deceleration component, i. H. the setpoint curve is point symmetrical. This symmetry considerably facilitates the determination of the respective setpoint profile under real-time conditions, as does the requirement that the acceleration or deceleration change linearly with a predetermined gradient in all positioning processes.

It can be seen that this procedure contributes to reducing the operating noise, since extreme values for changes in the acceleration are avoided as required. It can also be seen that settling times of the positioning unit above the target track are shortened with the known method and therefore average access times no longer take time despite the positioning methods maximized in terms of speed optimal setpoint can be achieved, which are the same or even better than with these older conventional methods. On closer inspection, however, it is immediately clear that even with this known method the task of reducing the access time while reducing the operating noise has not yet been optimally solved. Each of the described acceleration profiles contains more or less discontinuities in the temporal derivation of this acceleration function, which are still the reason for the excitation of mechanical vibrations and thus for the generation of operating noises, although due to the gradual increase and decrease in the acceleration or deceleration, an extension of the positioning time is accepted. The present invention is therefore based on the object of creating a method of the type mentioned which, with a further reduction in the average access time, eliminates the causes of the generation of undesired mechanical vibrations even more than hitherto, and also to provide a device for carrying out this method which inexpensive structure has a high flexibility to adapt to the respective application.

In a method of the type mentioned, this task is solved by the features described in the characterizing part of patent claim 1.

A major advantage of the solution according to the invention is that it succeeds in "smoothing" the acceleration for driving the positioning unit throughout the course of the setpoint curve, i. H. in other words, without specifying rounded points of discontinuity. However, a linear increase in the acceleration or deceleration is not accepted, but rather the aim is to optimize the time, to achieve maximum acceleration or deceleration values as quickly as possible by means of a corresponding current specification. This helps to reduce the average access time as well as the Rounding of the acceleration profile at the end of a positioning process, which allows the positioning unit to enter the target track as precisely as possible with an optimized settling behavior. In other words, the positioning method is based on a setpoint curve which, apart from the rounding, is preset in a time-optimized manner and is nevertheless also optimized with regard to the settling on the target track.

As further developments of the invention show, according to which the positioning process is always divided into three time intervals, regardless of the respective track distance, it is possible to regulate all positioning processes without case differentiation according to a uniform positioning process. It can be seen immediately that the distance from the respective route, i.e. H. the determination of a certain profile as a setpoint dependent on the track distance is to be decisively simplified. This increases the flexibility of the method according to the invention with reduced expenditure and contributes to a reduction in the computing time, which is of crucial importance in processes that run under real-time conditions.

Another important advantage of the method according to the invention is that it is also able to use the reserves in the control for driving the positioning unit, ie. H. make full use of the influence of the back EMF. In other words, this means detaching the setpoint profile from the point symmetry specified in conventional positioning methods for reasons of simplification. The detachment from point symmetry practically results in a deceleration interval that is shorter with respect to the acceleration interval and thus uses a further influencing variable that contributes to reducing the average access time.

As a further development of the invention according to claim 7 shows, this method using a commercially available signal processor is able to carry out the control process under real-time conditions without that of the respective track distance-dependent courses of the setpoint profile are already predetermined, for example in a setpoint memory. For this reason, a device for carrying out this method does not require any specialized integrated circuits that were previously not available on the market. Rather, the method can be implemented solely with the aid of conventional components, with a conventional signal processor with its properties being optimized with regard to certain addition and multiplication processes.

Embodiments of the invention are explained in more detail below with reference to the drawing. 1 shows schematically in a block diagram a control unit for a magnetic disk memory with an actual value generator and a setpoint generator, in which setpoints are generated for the positioning for each positioning process, FIGS. 2a to 2c schematically in the form of pulse diagrams the principle of a speed profile, a corresponding current profile for driving the positioning unit and a profile for the terminal voltage on a drive coil of the positioning unit,

3 shows an equivalent circuit diagram for the drive of the positioning unit,

4a and 4b, 4c and 4d or 4e and 4f in each case the comparison of a specific acceleration profile and a corresponding amplitude spectrum on the basis of a typical mechanical transfer function to explain the excitation of mechanical vibrations and operating noises, FIG. 5 shows the equivalent circuit diagram for an FIR filter,

FIG 6 in the form of pulse diagrams a step function and a filter response of an FIR filter corresponding to this step function,

7 shows a block diagram for a setpoint generator according to FIG. 1,

8 shows an example of a in the form of a pulse diagram

Current or acceleration profile, as it is initially determined, for example, for all positioning processes, FIG. 9 shows an embodiment for a setpoint generator in hard-wired circuit technology,

10a to 10c each in different pulse diagrams, embodiments for an acceleration profile, a speed profile or a path profile, which is derived with the setpoint generator from FIG. 9 after specification of an acceleration profile according to FIG. 8,

11 shows schematically in a block diagram a further embodiment of a device for carrying out the positioning method using a signal processor,

12 shows schematically in a block diagram the processes under control of the signal processor for converting the acceleration or current profile shown in FIG. 8 into a profile according to FIGS. 10a and

13 shows a flowchart for the control or computing processes carried out by the signal processor.

In FIG. 1, to explain a positioning method for a magnetic disk memory, a device control 1 is shown schematically in a block diagram, via which the magnetic disk memory contacts its environment. The device controller controls the data transfer from and to the magnetic disk storage and - which is of more important interest in the present case - also triggers a lane change if one Writing or reading in another track of the magnetic disk memory is to be continued. Such device controls are generally known. There is therefore no need for any further explanation here, because when a lane change is to be carried out, this device control merely provides a distance address ADR which outputs a corresponding jump distance in the form of the number of recording tracks to be overflowed and the respective direction, and a start signal START which defines the start of the lane change operation.

FIG. 1 also schematically shows a positioning unit 2 for the magnetic disk memory, which is to be designed as a rotary positioner. Rotary positioners for magnetic disk storage are particularly common for 5 1/4 "or 3 1/2" disk storage, so that no detailed description appears necessary for this unit. Therefore, it should only be pointed out here that conventional rotary positioners have a swivel piece which is rotatably mounted in the disk storage housing. On the one hand, the swivel piece carries, via a plurality of radially projecting support arms, a corresponding plurality of magnetic head systems which are exactly aligned with one another and which, when the swivel piece swivels, pivot in together over the surfaces of the storage disks of the magnetic disk storage assigned to them. The swivel is driven by a magnetic drive coil, to which a corresponding coil current is fed via a controlled power controller. The magnetic drive coil is surrounded by a system of permanent magnets so that it moves depending on the coil current and thus deflects the swivel piece. Such a drive device 201 of the positioning unit 2 is only indicated schematically in FIG. Typically, read / write electronics 202, which contains, in particular, write or read amplifiers assigned to the magnetic head systems, are also axially arranged in the positioning unit 2. It is assumed here that this read / write electronics 202 additionally also has an evaluation electronics for recording riding, rectification and selective comparison of servo signals. It is thus able to feed these prepared read signals RS to a phase-locked loop 3, which is used to derive a dependent internal clock pulse sequence CLK from these read signals RS in magnetic disk memories, which forms the time basis for all internal control processes and, for reasons of simplicity, for example also for the control of the control processes when positioning is used.

In addition, this read / write electronics 202 in the form of the evaluation electronics should also have switching devices which generate position error signals PES from the read servo signals. These usually form the control signals for tracking in magnetic disk memories. As indicated schematically, the position error signals here have a sawtooth-shaped course, the zeros of which occur when the positioning unit 2 continuously crosses recording tracks in the course of a track change.

All of the units described above are used in conventional magnetic disk memories in a wide variety of embodiments and are therefore generally known in the art. Beyond the above introduction, there is therefore no need for any further detailed description, which could only complicate the clarity.

In the following, however, the positioning process in a magnetic disk memory is to be considered in principle with regard to the positioning method to be described. The magnetic disk memory is a memory with random access, ie the positioning system must be adjustable to any track on the magnetic memory disks each time a track is changed. Furthermore, every positioning process should always run optimally, since the average positioning time is a key performance criterion in magnetic disk storage. One projects in the drive device 201 of the positioning unit 2 given supply voltage V is available, which is to be used optimally, in principle in the acceleration phase of the magnetic drive coil of the positioning device 2 to supply maximum acceleration current until the predetermined maximum speed of the positioning unit 2 is reached. This speed is maintained until the braking phase begins, in which the positioning unit 2 is braked with the maximum braking current, until it enters the target track as precisely as possible. Naturally, the braking phase begins with short distances, ie short track distances, before the maximum speed is reached.

This theoretical consideration makes it clear that a specific speed profile v. (T) can be used as a basis for each track distance in order to optimally carry out the positioning process.

A speed profile v _i (t) is shown in FIG. 2a, FIG. 2b shows a corresponding current curve i (t), ie a curve which corresponds to the time profile of the current supplied to the magnetic drive coil in the drive device 201 as a control variable, and FIG. 2c illustrates the corresponding one Course of the terminal voltage V (t) on the drive device 201 of the positioning unit 2.

In connection with an equivalent circuit diagram for the drive device 201 shown in FIG. 3, the influences which result from the structure of the drive device in practice are to be explained first. 3 shows a series circuit of a current source 203 and a magnetic drive coil 204 of the drive device 201 arranged between the supply voltage V _s and ground. In practice, the current source 203 is implemented by a power amplifier, to which a control voltage for setting the coil current i (t) is supplied. This power amplifier has a predetermined own requirement, which is effective as a voltage drop. You also need a certain span to be able to regulate during the positioning process reserve. These two influencing variables are shown in FIG. 2c as partial voltage V ₁ . The equivalent circuit diagram of the magnetic drive coil 204 clarifies the ohmic resistor 205 and the inductive resistor 206 of the magnetic drive coil, which are predetermined by the dimensioning of the coil 204.

Finally, the influence of the back emf is indicated in the form of a voltage source 207, which builds up as a voltage contribution that is linearly dependent on the speed and the motor constant K of the magnetic drive coil 204.

If you want to keep the terminal voltage at the magnetic drive coil 204 constant despite the increasing speed of the positioning unit 2, you have to specify a coil current i (t), as shown in FIG. 2b. The coil current i (t) follows an exponential drop over time according to relationship 1:

(1) i (t) = I _o exp (-K ² t / R),

where I _o = (V _s - V ₁ ) / R where K is the motor constant, R is the ohmic resistance of the magnetic drive coil 204, V _{s is} the supply voltage and V _{1 is} the partial voltage which corresponds to the self-consumption of the controlling power amplifier 203 and the control reserve. In the positioning procedure described here, this influence of the back EMF is used to advantage. The back EMF has a braking effect when accelerating, but can then be used to support the braking phase. However, this has the consequence that the setpoint curve for the coil current i (t) becomes asymmetrical, ie that the braking phase becomes shorter than the acceleration phase.

FIGS. 2a to 2c also illustrate, as an example, that in order to control the positioning process, a specific profile must be specified for each possible track distance in order to carry out the positioning process in a time-optimized manner. In a majority of conventional magnetic disk storage, a trapezoidal one is used Velocity profile v _i (t) is used. But if, for example, with 5 1/4 "magnetic disk storage media today you are aiming for average positioning times of less than 15 ms, this means speeds that can only be mastered by control technology with considerable effort due to the additional constraint that the target track should be approached as precisely as possible. In the present case, therefore, a path profile x _n (t) is used in connection with a current specification in (t) for the control of the positioning process, the latter determining the acceleration of the positioning unit 2 in the initial phase or its deceleration in the final phase of a positioning process.

So far, another essential problem has not been considered in detail, namely the problem of excitation of mechanical vibrations that result from the oscillating behavior of the positioning unit 2. Theoretically, it is desirable to apply maximum current as quickly as possible to the drive device 201 of the positioning unit 2, in order to allow the positioning process to be carried out in a time-optimal manner at high accelerations. This would be possible with a current profile i (t) or a corresponding acceleration profile a (t) with a stepped course, as shown by way of example in FIG. 4a. In a typical mechanical transfer function according to the positioning model considered here, such a current curve causes mechanical vibrations whose amplitude spectrum, i. H. the vibration amplitudes A plotted against the frequency f, qualitatively results in the image shown in FIG. 4b. The amplitude spectrum drops exponentially from a maximum value occurring at low frequencies, but still has relatively high maximum values even in the range of 10 kHz.

A current curve i (t) or the corresponding acceleration curve a (t) is shown schematically in FIG. 4c, which has a linear course pieced together. The associated amplitude spectrum A shown in FIG. 4d with the same transfer function shows a significantly more favorable course compared to the illustration in FIG. 4b, but always appreciable amplitude values even at frequencies above 5 kHz.

A current and acceleration curve i (t) or a (t) is shown in FIG. 4e, the edges of which rise or fall relatively steeply, but which is rounded, for example, according to a cosine function in the transition regions. The associated amplitude spectrum A shown in FIG. 4f shows the best values in a comparison of all FIGS. 4b, 4d and 4f. This shows convincingly that it is very important to design the current curve i (t) in such a way that all transitions run as smoothly as possible in order to avoid the excitation of mechanical vibrations, thus disturbances in the positioning process, such as long settling times and high operating noises. In other words, if only changes in the current or acceleration over time excite the mechanics to vibrate and then it is possible to specify the coil current i (t) with as few harmonics as possible, higher resonances in the amplitude spectrum can be avoided. In this case, the transfer function for the assumed positioner model, expressed mathematically in relation (2), applies:

(2) F (s) = x (s) / i (s) = K (Θs ² + ßs + d) ^-1

Here x is the deflection related to the servo head, Θ the moment of inertia of the positioning unit 2, 0 the speed proportional friction and d is a restoring force of the positioning unit, which can be attributed, for example, to the flexible cable feeds. Now you can assume that positioning times in a range from 1 ms to a maximum of 50 ms occur. Expressed in the frequency domain, this means fundamental frequencies from 200 Hz to about 1 kHz in the excitation spectrum. For these frequencies, the relationship (2) can be approximately expressed by a relationship (2a): (2a) F (s) = K / Θ s ²

This approximate relationship corresponds to the fact that the displacement of the positioning unit 2 is proportional to the double integration of the current in the magnetic drive coil 204, as long as the coil is not oversaturated.

It must now be made clear that such a current curve i (t), which differs from linear sections, must be made available during the positioning process. This is a task that is hardly feasible analytically under real time conditions with such a complex profile. As stated above, the current curve i (t) cannot be rounded off. Now analog technology generally offers to round a pulse train by pulse shaping, ie filtering. If this were also possible here, an unrounded pulse sequence could be specified as a "hard" current profile i (t) and rounded off by filtering. The above-mentioned requirements may apply, namely that the path x can be represented as a double integral of the acceleration a and that the acceleration a is directly proportional to the coil current i, ie that there is linear, saturation-free behavior. Furthermore, it is assumed that an analog control variable i _s (t) fed to the drive device 201 of the positioning unit 2 is linearly proportional to the current curve i (t). Under these conditions, it would be irrelevant whether this control variable is based on an already rounded current profile with the matching path x or whether one first specifies a hard current profile i (t), as indicated for example in FIG. 2b, with the associated path and then both in electrical Sends functions expressed by signals through the same filter and thus rounded.

This is opposed by the fact that defined starting and end conditions apply to exact positioning, namely that exactly defined positions are assumed by the positioning unit 2 at the beginning or at the end of each positioning process and at the same time there are at least the first and second derivatives directions of time must be zero. If you think of the filtering under consideration as being implemented by means of a low-pass filter, it can immediately be seen that the end condition cannot be met in any case, since an exponential entry into the end position is systematically predetermined according to the transfer function of such a filter. In theory, this means that the end position would only be taken exactly after an infinitely long time. Common analog filters are therefore not an acceptable solution to this problem.

However, finite impulse response filters are known from digital filter technology, which are usually abbreviated as FIR filters. A circuit design of such a filter is such. B. in IEEE Journal of Solid-State Circuits, Vol. 23, No. 2, April 1988, pages 536 to 542 and discussed in detail. Knowledge of this type of filter can therefore be assumed here. In FIG. 5, an equivalent circuit diagram for such an FIR filter is therefore shown schematically only for better understanding. Since the rules of the z-transformation are to be applied to digital filters, each block z in this equivalent circuit diagram means the shift of the input signal SI by one sampling period

Δt. The filter coefficients are denoted by c _o to c _n , so that according to relationship (3):

(3) Ftz ^-1 ) = c _o + c ₁ z ^-1 + c ₂ z ^-2 + ... + c _n z ^-n the transfer function F (z ^-1 ) of the FIR filter for a filter n-th Order can be specified.

6 schematically illustrates this filter function for a 5th order FIR filter. A broken line pulse is shown with broken lines as the input signal SI. The filter response, ie the output signal SO of the FIR filter, is the step-shaped signal curve which, after a number n of sampling periods corresponding to the order n of the filter, corresponds exactly to the end value of the input signal SI. This illustrates the decisive property of the FIR filter, according to which the FIR filter has "settled" precisely on the input signal after a finite, predetermined time. Furthermore, it can be seen immediately that by choosing the appropriate filter coefficients c _o to c _n , it is possible to round the filter response, ie the output signal SO, in any way desired between the start and the end value. Using an FIR filter, it is therefore possible to specify the current profile "hard" and still achieve a "soft" setpoint value with an adjustable rounding curve while observing the described boundary conditions.

This explains the basics for the positioning procedure here. Now, returning to the illustration in FIG. 1, the implementation of the principles described above can be explained. The block diagram shows an actual value transmitter 4, to which the sawtooth-shaped position error signal PES is fed. This signal has a period that is identical to the track grid and has a stroke that fully utilizes the available supply voltage for a single track distance. The device address 1 supplies this actual value transmitter 4 with the distance address ADR, which defines the jump distance for the upcoming lane change. Triggered by the START signal START, which is also supplied by the device controller 1, the actual value transmitter 4 evaluates the supplied position error signal PES in such a way that it reduces the distance address ADR by the value "1" for each detected track crossing. The respectively remaining current remainder is output as the actual path value x _a for the deflection of the positioning unit 2 which is still to be covered.

A possible implementation for this actual value transmitter 4 is shown in FIG. 7. Thereafter, it has a loadable counter 41 designed as a down counter, at whose data inputs DI the distance address ADR is present and is loaded when the start signal START occurs. For the evaluation of the position error A zero crossing detector 42 is provided for the signal PES. This derives a rectangular pulse sequence from the position error signal, which is supplied to the counter 41 as a clock signal. After each leading edge of this clock signal, the counter 41 thus outputs a distance address reduced by "1" via data outputs DO. This is fed to an adder 43, at whose second inputs the position error signal PES converted into a binary value via an analog-digital converter 44 is present. The actual path value xa output by the adder 43 is therefore composed of the distance still to be covered, expressed in the number of tracks still to be crossed, and a fine adjustment value which corresponds to the current offset from the middle of the track of the last crossed track. This already indicates that, in contrast to many conventional designs in magnetic disk memories, the positioning method here does not switch between control systems for tracking and control systems for changing lanes. FIG. 1 also schematically indicates a setpoint generator 5, the nominal values. x _n and in for the still to be covered

Distance or generated for the given coil current. This setpoint generator 5 is - as a possible embodiment - constructed in hard-wired circuit technology and has a function generator 51 which, depending on the value of the distance address ADR supplied to it after the start signal START has been supplied, continuously generates the non-rounded current specification curve i (corresponding to the respective positioning process) t) generated. This unrounded current specification curve is fed to a FIR filter 52 in the form of binary values. This filter should be of a sufficiently high order, for example 20th order, in order to generate a filter response which is rounded in the desired manner, preferably cosine, and which is output in binary form as the nominal current specification i _n .

Under the conditions described above, the nominal current specification i _n corresponds linearly to a nominal acceleration tion, ie the second derivative of the remaining nominal

Path x _n according to time. Therefore, the nominal current specification i _{n is} fed to an integration stage 53, which forms twice the integral of the current value and outputs it as the nominal displacement specification x _n . The processes in the setpoint generator 5 are controlled by a z. B. generated by the phase locked loop 3 internal clock pulse sequence CLK.

The calculation basis on which the function generator 51 is based is explained below with reference to FIG. 8. In order to minimize the circuitry and to be able to carry out the current specification in real time during the lane change, this basis should be as simple as possible and apply to the full range of all possible lane changes over short and long distances. The diagram shown in FIG. 7 for an unrounded, "hard" current specification curve i (t) applies to all lane changes over longer distances with speed limitation of the positioning unit 2. This current specification curve takes into account the described influence of the back emf. However, to simplify the calculation, the exponentially falling branches are the

Current curve i (t), as clearly shown in FIG. 2b, linearized, so that the hard current specification curve is composed of straight sections. As shown in FIG. 8, three time intervals δ t1 to δt3 are distinguished. In order to clearly define the linearly specified current specification curve i (t), it is only necessary to define the basic values for the current at the beginning and at the end of these time intervals dt1 to δt3. Here i _{o is} the instantaneous value for the current at the start time to. The acceleration phase ends at time t1 with a current value i ₁ , which jumps to the value zero. At time t2, the braking phase begins with a current value -i ₂ and at time t3 the braking phase ends with a current value -i ₃ , which in turn jumps to zero. The linear sections of the current specification curve i (t) during the time intervals δt1 and δt3 are described as gradients m ₁ and m ₃ by the current drop di / dt approximating the back emf. The parameters i ₀ , i ₃ , m ₁ and m ₃ are each independent of the respective track distance, ie they are device constants. This makes it clear that the individual positioning processes differ only in the length of the time intervals δt1 to δt3.

A special case of this hard current specification curve i (t) shown in FIG. 8 applies to lane changes over short lane distances up to a single lane change. In these cases, the positioning unit 2 does not reach its predetermined limit speed, i. H. this means that - considered in principle - the time interval δt2 = 0.

Under these conditions, all variable parameters for all track distances can be calculated in advance, ie specified in a suitable manner. FIG. 9 shows an embodiment for the function generator 51 which is embodied in a hard-wired circuit and is shown as a block diagram on which this calculation basis is based. The variable parameters, but expediently also the constants, are stored in a programmable read-only memory 510 for all track distances and can be selected by the distance addresses ADR. At the start time for a positioning process, triggered by the start signal START, the current start value i _{0 is} loaded into a first counter 511. This counter is designed as a down counter, which is controlled by the internal clock pulse sequence CLK. Its counter reading decreases in time with this control signal and is continuously read out to a multiplexer 512. Set by the START signal, this multiplexer switches this variable current value during the time interval

£ tl through to the output of the function generator 51 and thus to the FIR filter 52 connected to it. This outputs the rounded nominal current control voltage value i _n derived therefrom via a delay element 54, the function of which will be explained below.

A first digital comparator 513 is also provided The current value i _{1 is} supplied to the first inputs from the programmable read-only memory 510 and the second inputs of the current value are also connected to the outputs of the first counter 511. This comparator thus continuously compares the counter reading of the first counter 511 with the final value of the current at the end of the first time interval δt1 of the unrounded current specification curve i (t). The first comparator thus determines the expiry of the first time interval δt1 when the signals supplied to it are identical. At this time, the multiplexer 512 is switched by the output signal of this first comparator 513 and at the same time a second counter 514 is loaded with a value that corresponds to the length of the second time interval δt2. According to the circuit principle described above, the outputs of this second counter 514 are connected to inputs of a second digital comparator 515. This digital comparator is also supplied with a value "0", preferably from the programmable read-only memory 510, so that it determines when the second time interval δt2 has expired. When the second time interval δt2 has elapsed, the second comparator 515 outputs an output signal which switches the multiplexer 512 and at the same time controls the loading of a third counter 516 with the braking current value -i ₂ at the beginning of the third time interval δt3. Again, in the same way in the course of the third time interval δt3, the counter reading of this third counter 516, which is continuously reduced by the internal clock pulse sequence CLK, is fed to the multiplexer 512 and, at the same time, inputs of a third comparator 517. This comparator is also offered the final value -i ₃ for the braking current from the read-only memory 510. At the end of the third time interval δt3, this comparator 517 establishes the identity of the input signals fixed and emits a corresponding output signal. This control signal STOP indicates the end of the unrounded current specification for the positioning process and is supplied to the device controller 1 and the integrator 53 shown in FIG. An embodiment for this integrator 53 is also shown in FIG. Under the conditions described above for a linear, ie saturation-free, operation, as stated, the output signal emitted by the FIR filter 52 represents the second derivative of the nominal path specification x _n . Since the nominal current specification i _{n is available} as a digital value, the integration of this value can be attributed to a continued addition. Therefore, the integrator 53 is composed of two adders 530 and 531. The first adder is connected to the FIR filter 52 with first signal inputs. Its outputs are both connected to first inputs of the second adder 531 and also fed back to its second inputs. The second adder 531 is connected in a corresponding manner, so that both adders carry out continuous additions of the input signals supplied in each case and together double integration of the output signals of the FIR filter

52 perform. The integrator 53 thus generates the nominal one

Path specification x _n . 10a to 10c show pulse diagrams which exemplify the effect of the FIR filter arrangement 52. A current or acceleration profile a (t), i (t) is shown with broken lines in FIG. 10a, which corresponds to the unrounded specification of FIG. With solid lines, the associated filtered default curve is a (t); i _n (t) shown. In addition to a certain delay in the ms range compared to the non-rounded default curve, the rounding effect of the FIR filter 52 can be seen in particular. It is particularly important that the defined final state is reached exactly after a finite delay predetermined by the order of the FIR filter 52 is. In FIGS. 10b and 10c, speed profiles v (t), v _n (t) and path profiles x (t) are analogous to the acceleration profile of FIG. 10a; x _n (t) not rounded and filtered. The three time segments of the setpoint curve are illustrated there analogously by the times t ₀ , t ₁ , t ₂ and t ₃ .

This describes in detail how the actual and target values can be obtained for the control of the positioning process. Returning to the block diagram of FIG. 1, it is shown there that the actual travel value x _a and the nominal travel specification x _n are fed to a digital comparator 6 which, as a reference variable, outputs a digital signal corresponding to the deviation between the actual travel value x and the nominal travel specification x _n . On the basis of the control principle, this control deviation should ideally be zero during the positioning process, but in the practical case it should represent a finite small value that only takes into account faults or influencing factors of the mechanics not detected by the calculation of the setpoints.

The essential reference variable for the positioning process, on the other hand, is the nominal current specification i _n , to which, delayed by the delay element 54, the output signal of the digital comparator 6 is added in an adder 7. The delay element 54 compensates for transit times in the integrator 53 and in the digital comparator 6, so that the input variables offered to the adder 7 are phase-synchronized. The output signal of this adder 7 thus represents the actual reference variable for the positioning process, which is converted in a digital / analog converter 8 into the analog control variable i _s (t). This control variable is supplied to the drive device 201 of the positioning unit 2 as a control signal for the power amplifier 203 forming the switched current source.

A first exemplary embodiment has been described above which allows the positioning method on which the underlying method is based to be carried out in hard-wired circuit technology. Despite all the optimization of the method steps described, such a hard-wired solution is always relatively complex and also not very flexible with regard to adaptations, for example changing the device constants. A way out is a programmable controller. Today, digital signal processors are available that are suitable for this application, since they are optimized for filter calculations in which multiplications and additions predominate. The A person skilled in the art is, for example, known from the publications "Digital Signal Processing Applications With The TMS 320 Family", 1986 and "First Generation TMS 320 User's Guide", 1987, both published by Texas Instruments, an extensive range of instruments, such as how to use the TMS 320 CIO signal processor ^{® can be used} in a wide variety of applications. The processor mentioned has a cycle time of 200 ns with a word length of 16 bits, the accumulator being 32 bits wide and the multiplier delivering a 32 bit result with a thickness of 16 x 16 bits. With reference to the second of the above-mentioned publications, an explanation of the performance of this signal processor is given here by way of example to an instruction sequence MPY and LTD, which is described there on pages 4/49 and 4/41. In a total of only two clock cycles, ie in 400 ns, this command sequence results in a 16 x 16-bit multiplication, a 32-bit addition to the accumulator, the loading of a RAM cell into the multiplier and the shifting of the contents of this RAM cell to the next higher address.

In FIG. 10, as a further embodiment, an intelligent positioning control unit 9 is shown schematically in a block diagram, which is connected to the device controller 1. In addition to a digital signal processor 91 of the type described above, an 8-bit wide analog / digital converter 92 and a 10-bit wide digital / analog converter 93 are provided in this positioning control unit. The above-mentioned units are connected to one another via an interface 94 which, as in many conventional microprocessor applications, has simple logic circuits which enable the signal processor 91 to exchange data with the connected units.

The analog / digital converter 92 and the digital / analog converter 93 can be implemented by an AD7569 module from Analog Devices®, which has an 8-bit analog / digital converter and two 8-bit digital / analog converters and corresponding inputs /Output and contains selection logic. This module can be used in such a way that its two digital / analog converters together form the digital / analog converter 93. For this purpose, they are switched in such a way that one only outputs the signal corresponding to the current specification curve i _n during a positioning process and the other outputs the actual control signal for tracking. The gain of the first digital / analog converter is four times higher than that of the second. The result is that the resolution for the nominal current command i _n corresponds to a 10-bit value. Because of the availability of a corresponding fast module, this division is more cost-effective and possibly also more expedient in view of the real-time conditions than a correspondingly broad individual digital / analog converter, as is indicated schematically in FIG. 10.

The design of the positioning control unit 9 is based on the principles which have already been explained in detail in connection with the first exemplary embodiment. A significant difference in the implementation here is, however, that the default values, made possible by the performance of the digital signal processor 91, are determined in real time during a positioning process, which is particularly advantageous with regard to a possible adaptability in this embodiment for carrying out the positioning method . Since the calculation methods for determining the time intervals δt1 to δt3 are optimized with regard to the properties of the digital signal processor 91 used, they are explained below. Here, too, it is assumed that the following basic conditions apply to each positioning process. The final speed v (t ₃ ) at the end of the positioning process must be zero, the positioning distance, ie the distance x (t ₃ ) covered at time t3 corresponds to the nominal path specification x _n and the following relationship applies to the limit speed during the positioning process (4): ⁽⁴⁾ v _lim = German

The validity of the following relationship (5) can also be derived from FIG. 8 for positioning processes without speed limitation in which the second time interval t ₂ is actually zero:

(5) 2a ₀ δt ₁ + m ₁ δt ₁ ² = -2a ₃ δ t ₃ + m ₃ dt ₃ ² This relationship (5) says nothing more than that because of the energy theorem the areas of the two parts of the electricity or

Acceleration specification curve according to FIG 8 must be the same.

Finally, the following relationship applies to the distance over the full track distance (6):

(6) (1/2) a _n δt ₁ ² + (1/6) m ₁ δt ₁ ³ (1/2) a ₃ δt ² +

+ (1/6) m ₃ δt ₃ ³ = x _n The relationships (5) and (6) form a system of equations with two unknowns, namely δt ₁ and δt ₃ depending on the distance x _n for the full track distance. However, these relationships cannot be solved explicitly, since no analytical functions δt ₁ = f ₁ (x _n ) or δt ₃ = f ₃ (x _n ) can be established.

This problem is solved by an estimate for the length of the first time interval δt ₁ as a function of the respective track distance. For this estimate δt ₁ 'it is only assumed that the condition δt ₁ '<δt _{1 is} fulfilled. An estimated value δt ₃ 'can then be determined on the basis of the following relationship (7), which is derived from relationship (5):

(7)

With these estimates, the relationship (6) gives an estimated positioning range xs', which must be smaller than because of the requirements δt ₁ '<δt ₁ and δt ₃ '<δt ₃ the full positioning range xs. The difference between the actual positioning range xs and the estimated positioning range xs' is compensated for by introducing the time interval δt ₂ . During this time interval, the positioning unit 2 is constantly moved at the speed v, at the end of the time period δt ₁ '. With reference to relationship (5), this velocity v _{1 is} calculated according to the following relationship (8): (8) v ₁ = a ₀ δt ₁ '+ (1/2) m ₁ δt ₁ ' ²

The length of the time interval (δt ₂ ) can be determined from the following relationship (9): (9) δt ₂ = (xs - xs') / v ₁

The derivation described above initially has the advantage that the estimate for the time interval δt ₁ 'is based on a simple relationship which, in contrast to an exact calculation, requires considerably less programming effort, computing time and storage space requirement. Since the time interval δt ₁ 'is estimated anyway, it can always be selected as an integer value of sampling periods Δt, thereby avoiding correction calculations of the specification curve in the transition area between the time intervals δt ₁ and δt ₂ . Another very important advantage is that, regardless of the track distance, a time interval δt _{2 is} introduced with each positioning process or, in other words, there is no need to distinguish between positioning processes over short or long track distances.

Provided that saturation-free operation is used, a speed limitation of the positioning unit 2 must of course be specified for longer track distances, but this results automatically by limiting the first time interval δt ₁ . The maximum length of this time interval results from the following relationship (10):

the result being positive since m. is a negative quantity. This maximum value for the first time interval δt1 is an operating constant which is calculated once during a restart of the magnetic disk storage in a start-up routine. In the explanation of the first exemplary embodiment, it has already been described that for determining the default values for current and distance, because of the assumed linearity of the transfer functions, only one of the two default curves is calculated without rounding and then filtered for rounding, it being irrelevant whether one is interested in the current - or acceleration values or the values for the path. In the first exemplary embodiment, the current specification was used as an example and the path specification was obtained from it by integration. Here, in the second exemplary embodiment, a path specification is used, from which the rounded nominal current specification is determined by double differentiation after the filtering.

As explained, the non-rounded default curve is to be shaped by filtering in such a way that it specifies the current for exciting the magnetic drive coil of the positioning unit 2 as harmless as possible. This goal can only be adequately achieved with a higher order FIR filter. The basis here is an FIR filter, for example of the 20th order, which is implemented by a filter program of the signal processor 91. Because of the high order of the FIR filter, a corresponding amount of computing time arises, which should be minimized as far as possible, since the determination of the default values for current and distance takes place in a time-critical area.

In order to relieve the load on the digital signal processor 91, one is therefore first used to calculate the unfiltered route specification Extrapolation routine for the route is used, which determines a current route setpoint x _k for the unrounded route specification from the three route setpoints of the previous sampling periods Δt according to the following relationship (11):

(11) x _k = 3x _k-1 - 3x _{k-2 +} x _{k-3 +} m _i Δt ³ i is one of the values 1, 2 or 3 that relates to one of the time intervals δt ₁ , δt ₂ or refer to δt ₃ . The last term from relationship (11), namely m _i Δt ^3, is constant within these time intervals. The relationship (11) thus corresponds formally to a filter operation that can be processed by the signal processor 91 in a time-optimized manner. In this extrapolation routine it should be noted that when moving from a period, e.g. B. δt ₁ for the next period, e.g. B. δt ₂ does not have any corresponding previous travel setpoints per se. For these transitions in the unrounded path specification curve, these three previous path setpoints must therefore be simulated, ie calculated before the actual execution of the positioning process and made available in a memory. It must also be clarified that the path information, for example a current path setpoint x _{k, is} absolute information which relates to the width of the entire data band on the magnetic storage disks. For accuracy reasons, a data format of 32 bit positions for the x-variables is therefore advisable for a 5 1/4 "magnetic disk memory. This format corresponds to two data words for a 16-bit signal processor. This extrapolation from three previous setpoint values could also be avoided. One could namely, instead of determining the path setpoint x _{k of} the current sampling period from the last path setpoint and its two derivatives, as will be shown, as can easily be seen from the previous considerations.

After continuous calculation of the current travel setpoints x _k , the corresponding rounded quantity is filtered using Using the FIR filter to determine. It has already been indicated above that this filter should be of the 20th order. 5 and 6 it has already been explained that the filter order results from the ratio of smoothing time to the sampling period. If the sampling period is 60 ms and a 20 th order filter is used, the result is a smoothing time of 1.2 ms, which is still acceptable in the present application.

However, if the route information is in a 32-bit format, the selected high-performance digital signal processor 91 would still result in an excessively long computing time despite the algorithms described. One solution to this problem is not to filter the absolute value for the current travel setpoint x _k , but only to use its travel increment δx _k to the previous travel setpoint x _k-1 . This path increment can be represented with sufficient accuracy in a 16-bit format. In other words, one forms the path increment δx _k before filtering and then filters it. The filtered path increment thus obtained is added to the previous absolute value of the nominal path specification x _{nk -1} and thus the absolute size for the filtered path specification, ie the nominal path specification x _nk for the current sampling period. In this exemplary embodiment, the rounded current specification i _{n is} derived from the nominal path specification x _n by double differentiation. In the present case, this can be reduced to the formation of the second difference according to the following relationship (12):

(12) i _nk = (x _nk - 2x _nk-1 + x _nk-2 ) / Δt ²

However, since, as explained above, the path increments δx _k are already used as the basis for filtering the path specification, this differentiation is further simplified, as is expressed in the following relationship (12a): (12a) i _nk = (δx _nk - δx _{nk- 1} ) / Δt ²

This relationship obviously illustrates how easily the nominal current specification i _n can be obtained. This applies in particular if the sampling period Δt is nominally set to "1". This also eliminates the division according to relationship (12) or (12a) and the nominal current specification i _n results from a simple subtraction of the corresponding path increments. The above calculation basis is based on the assumption of ideal mechanics. However, the losses that occur in practice with a positioning unit 2 can possibly be taken into account if the relationship (12a) is expanded to the following relationship (12b):

(12b) i _nk = (δx _nk - δ x _nk-1 ) ₊ α ₀ + α ₁ . x _nk + α ₂ . δx _nk

The expansion term α ₀ corresponds to a constant acceleration, which compensates for an unbalance of the positioning unit 2, for example. The second expansion term α ₁ . x _nk corresponds to a path-proportional acceleration that takes into account a force that acts proportionally on the positioning unit 2 and compensates, for example, a restoring force caused by connecting lines. The third extension term α ₂ . δx _nk corresponds to an acceleration proportional to speed and thus takes into account, for example, a friction in the positioning unit 2. This shows that with the help of the relationship (12b) it is also possible to control a positioning unit 2 which deviates to a considerable extent from the quantities on which the ideal mechanics are based.

The above-described calculations to be carried out by the digital signal processor 91 are shown in FIG. 11 for illustration and in summary form. Block 110 shows an equivalent circuit diagram for the path extrapolation described. Line 111 shows basic values x _k-2 , x _k-2 and x _k-3 or m _i Δ t ³ , each of which begins at the beginning of a new time interval δt ₁ , δt ₂ and δt ₃ are loaded into registers of the signal processor 91 in order to suitably initialize the path extrapolation. As a variant, the current path setpoint x _k can already be loaded in a register 112 and the simulated previous path setpoints x _k and x _k-2 in corresponding registers. This saves the calculation of the first extrapolation value, ie the current travel setpoint x _k, and thus compensates for the time required to load the basic values. The three path parameters for one of the time intervals δt _i result from the parameters a ₀ , v ₀ and x _{0 of} this time interval according to the following relationships (13), (14) and (15):

(13) x _k = x ₀ (14) x _k-1 = x ₀ - v ₀ Δt + (1/2) a ₀ Δt ² - (1/6) m _i Δt ³

(15) x _k-2 = x ₀ - 2v ₀ Δt, + 2a ₀ Δt ² - (8/6) m _i Δt ³ where relationships (14) and (15) are derived from relationship (6). These relationships also make it clear that - as indicated as a variant above - the current travel setpoint x _{k could also be determined} from the previous travel setpoint x _k-1 and its two derivatives. As illustrated in FIG. 11 with dash-dotted lines, the content of the register 112 for the path setpoint x _{k is} continuously loaded into a register 113 as the calculation progresses, which register thus contains the old path setpoint x _{k-1 of} the previous sampling period Δt. In addition, with the content of the register 112 with progressive path extrapolation, the contents of the registers for the basic values are successively updated.

As stated above, the first time interval δt _{1 is} always an integer multiple of the sampling period. With that lies the

Transition from the sampling interval δt ₁ to the sampling interval δt ₂ always in synchronism with the sampling grid. It is different with the transition from the time interval δt ₂ to the time interval δt ₃ , because - as explained above - the second time interval δt _{2 is} not an integer multiple of a sampling period Δt. This must be taken into account when calculating the basic values for the beginning of the time interval δt ₃ . If the period of time between the start of the third time interval δt ₃ or the next sampling period Δt is designated as Δt _f, the following corrected values based on the relationships (16), (17) result for the first sampling period Δt occurring after the beginning of the third time interval ) or (18):

(16) a _e3 = a ₃ + m ₃ Δt _f

(17) v _e3 = v ₁ + a ₃ Δt _f + (1/2) m ₃ 4t _f ²

(18) x _e3 = x ₁ + v ₁ (δt ₂ + Δt _f ) + (1/2) a ₃ Δt _f ² + (1/6) m ₃ Δ t _f ³

This defines all boundary conditions for the path extrapolation, which is illustrated in FIG. 11 by the equivalent circuit diagram 110.

The equivalent circuit diagram 114 reproduces the path increment described, in which the path setpoints x _k and x _{k-1 of} the current or the previous sampling period are subtracted from one another and result in the path increment δx _k loaded in a register 115.

Block 116 represents an equivalent circuit diagram for the FIR filter, in which path increments x _ki , evaluated with corresponding filter coefficients c _i , are added up. Here i means an integer value from 0 to n, where n indicates the order of the FIR filter.

The result of this filtering is a value loaded in a register 117 for the rounded path increment δx _nk , which progressively moves into a further register 118 after each sampling period as a path increment of the previous sampling period is saved. Block 119 thus designates the equivalent circuit diagram for the described two-fold differentiation of the path increments, so that the current value i _nk for the nominal current specification i _n results therefrom. The value temporarily stored in the register 117 for the rounded path increment δx _nk is - as indicated by a delay element 121 - added to the content of a register 120 with a delay, which contains the absolute value for the path specification x _{nk-1 of} the previous sampling period. The current absolute value x _nk for the nominal path specification x _n is thus continuously output in synchronism with the current value for the current specification i _n .

FIG. 12 shows a flow chart for the entire positioning routine. After the start of the positioning process, the values for the time intervals δt ₁ , δ t ₂ and δt _{3 are first} calculated in method step 1201. With these values obtained in this way, the basic parameters for each time interval, as indicated in method step 1202, are determined before the actual positioning process is carried out. With zeroing the values for the time interval δt and the

Time lapse t in method step 1203, the positioning control unit 9 and thus its signal processor 91 is set to a defined initial state at the beginning of the actual positioning process. Then, in order to initiate the first time interval δt _{1 in} accordance with method step 1204, the value for the current time interval is incremented by "1". After the initiation of this new time interval, the basic values assigned to it are first loaded in accordance with method step 1205. In the course of the first time interval δt ₁ , the path increments δx _k are then successively calculated in accordance with method step 1206 and filtered in accordance with method step 1207. The current path increment output at the output of the FIR filter is, generally considered, added to the absolute size of the previous path setpoint and the current absolute size x _{nk of} the nominal path specification x _n is thus specified in method step 1208. In step 1209, this is followed by the calculation of the corresponding current variable i _nk for the nominal current specification i _n . The absolute values x _nk thus calculated for the

The path specification and i _nk for the current specification are output, ie fed to the comparator 6 according to FIG. 1 or to the digital / analog converter, as indicated in method step 1210.

The time sequence is then incremented by the value "1" in method step 1211 and, according to branch 1212, the condition is queried as to whether t is already greater than the calculated total time for the positioning process. If this is the case, the positioning process is finished. Otherwise, a further condition is queried in method step 1213 as to whether the current time interval δt ₁ , δt ₂ or δt _{3 has} expired. If this is not the case, the current setpoint value x _k for the applicable time period is calculated in accordance with method step 1214 and the program sequence continues with the calculation of the next path increment δx _{k in} accordance with method step 1206. However, if the branching condition in method step 1213 is fulfilled, ie the current time interval has been completed, then the program flow continues according to method step 1204 with the incrementing of the time interval.

Finally, it should be pointed out that for the positioning control unit 9 there is no difference between a positioning process with an adjustment to a new track and a control process for tracking a selected track. There are therefore no switching processes between a control for tracking and a control for positioning on a new track.

During the control processes for tracking a selected track, two variables, namely the actual travel value x _a and the nominal travel specification x _{n, are managed} as absolute values in the memory of the signal processor 91. During the control process, only the difference between these two variables is evaluated, each of which has a 32-bit format. With these two data words The higher-order data word contains the information about the track number in whole numbers, the lower-order data word, on the other hand, the deviation from the track center in the form of fractions of a track. With this format, the entire width of the data tape on the magnetic storage disks can be represented linearly with high accuracy.

Any nominal track and additionally a selectable deviation from this track can thus be specified via the nominal path specification x _n . As was explained on the basis of the description of FIG. 1, the actual path value x _{a is} continuously updated via the position error signals PES which are read in, track crossings being automatically taken into account. The next expected value is extrapolated from the course of this path actual value in the past and the corresponding edge of the position error signal PES is selected. As is known, in the case of magnetic disk memories, adjacent information tracks are distinguished by the track type which is defined in the servo cells containing the track information on the servo surface of the magnetic disk memory. In connection with this differentiation, the described method can ensure correct track selection even if, at the limit speed of the positioning unit 2 of, for example, more than 1.5 m / s, up to six tracks are "blindly" flown over in a sampling period Δt. Due to the separate management of the actual path value x _a and the nominal path specification x _n , any path errors are permitted in principle, this applies both to the positioning process and when settling onto a selected track. This principle also allows, if expedient, positioning operations over short track distances to be brought about solely by the sudden change in the nominal path specification x _n , although a corresponding settling time must be expected. Reference list

1 device control

ADR distance address

START start signal

2 positioning unit

201 drive device for 2

202 read / write electronics

RS servo signals

3 phase locked loop

CLK internal clock pulse train

PES position error signals

V _s supply voltage of 201

v _i (t) speed profile

i (t) current specification curve

V (t) terminal voltage at 201

203 power amplifier (current source) from 201

204 solenoid drive coil of 201

V ₁ partial voltage from control reserve and own consumption of 203

205 ohmic resistance from 204

206 inductive resistance of 204

207 back emf (voltage source) of 204

K motor constant of 204

x (t) path over track distance

a (t) acceleration of 2

A amplitude spectrum

z ^-1 filter block of the FIR filter

Δt sampling period

c ₀ ... c _n filter coefficients

SI input signal

SO output signal

4 actual value transmitter

x ^a actual position value

41 counters of 4

DI data inputs from 41

42 zero crossing detector D0 data outputs from 41

43 adders of 4

44 analog / digital converter

5 setpoint adjuster

51 function generator

52 FIR filters

i _n nominal current specification

x _n nominal path specification

53 integrator stage

δ t1 ... δt3 time intervals of the current specification curve t0 start time

t1 end of acceleration phase

t2 start of braking phase

t3 end of braking phase

i ₀ ... i ₃ current values at times t0 ... t3 m1, m3 current drop in δt1 or δt3

510 programmable read-only memory

511, 514, 516 counters of 51

512 multiplexers of 51

513, 515, 517 compressors from 51

530, 531 adders of 53

6 digital comparator

7 adders

8 analog / digital converters

9 positioning control unit

91 digital signal processor

92 analog / digital converters

93 digital / analog converter

94 Connection logic

v _lim limit speed

δt ₁ 'Estimated value for time interval δt ₁ xs positioning values

xs' estimated positioning values

δxs difference xs - xs'

x _k current travel setpoint

δx _k path increment

110 Equivalent circuit for path extrapolation 111 Equivalent circuit for basic values

112 Register for travel setpoint x _k

113 Register for travel setpoint x _k-1

Δt _f Time period between the beginning of δt ₃ and the subsequent sampling period Δ t

114 Equivalent circuit for path increment formation

115 register for path increment

116 Equivalent circuit for FIR filters

117 Register for rounded path increment

118 register for path increment of the previous

Sampling period

119 Equivalent circuit for double differentiation of the path increments

120 registers for distance setpoint of the previous ones

Sampling period

121 delay element

1201 ... 1214 procedural steps in the flow chart

Claims

1. Method for positioning a positioning unit (2) of a magnetic disk storage from a start track on a magnetic storage disk to a target track by continuously comparing the current position (xa) of the positioning unit as the actual value with the target position (xn) as the target value using one of the distance between the start - and target track dependent setpoint curve for the current supplied to a drive (201) of the positioning unit, characterized in that a setpoint curve designed as a step function is assumed, according to which the positioning unit with a maximum nominal current specification (i _n ) at the beginning of the positioning process, if necessary, until it is reached a predetermined limit speed (v _lim ) is accelerated or decelerated at the end of the positioning process that corresponding sample values with a predetermined sampling period (Δt) are continuously fed to this setpoint curve, an FIR filter arrangement (e.g. 52) whose filter select coefficients (co ... cn) are selected such that the kinks in the setpoint curve are rounded and that a control voltage (i _s (t)) is derived from these sample values (i _nk or x _nk ) thus filtered, which

Drive (201) of the positioning unit (2) is supplied as a control variable.

2. Positioning method according to claim 1, d a d u r c h g e k e n n z e i c h n e t that a positioning process is divided into three time intervals (δt1 to δt3) regardless of the respective track distance, that the current supplied to the drive (201) of the positioning unit (2) in the first time interval

(δt1) is controlled in such a way that, taking into account the back emf with rounded transitions, it runs linearly from an initial maximum to approximately the end of this time interval and then drops steeply, that this current is essentially zero in the second time interval (δt2) and that this current at the beginning of the third time interval (δt3) using the back emf of the drive (201) of the positioning unit (2) steep increase quickly reaches an excessive negative extreme value, from which it initially decreases linearly and then steeply towards the end of the third time interval.

3. Positioning method according to claim 2, characterized in that the length of the first time interval (δt1) for a positioning process is always chosen such that this length corresponds to an integral multiple of the sampling period (Δt), being shorter than the corresponding length at an absolutely optimal time designed target value curve and even with large track distances does not exceed a predetermined maximum value (δ tlmax), which corresponds to a limit speed (V _lim ) of the positioning unit (2), at which it is ensured that the drive (201) of the positioning unit still works in a saturation-free range.

4. Positioning method according to claim 3, d a d u r c h g e k e n n z e i c h n e t that for the nominal path and

Current specification (x _n or i _n ) only one of these variables is explicitly derived from a common setpoint curve and then filtered, and that in the case of a nominal path specification (x _n ) derived in this way, the nominal current specification (i _n ) is derived from this by double differentiation or in the case of a nominal current specification (i _n ) derived in this way, the nominal path specification (x _n ) is obtained by double integration.

5. Positioning method according to one of claims 1 to 4, characterized in that the samples (x _k ) in the filter arrangement (z. B. 52) are rounded according to a cos function.

6. Positioning method according to one of claims 1 to 5, characterized in that the actual value for the current position of the positioning unit (2) as a distance already traveled over the respective track distance (x _a ) from position error signals (PES) in the form of the number of during the positioning process already crossed tracks as well a current storage from the middle of the last crossed track is continuously determined and compared with the nominal path specification (x _n ) as a setpoint that the deviation of the nominal path specification (x _n ) from the actual value (x _n ) of the nominal current specification (i _n ) superimposed and from this the control signal supplied to the positioning unit (i _s (t)) for setting the current amplitude for the drive (201) of the positioning unit is derived, and that the nominal current specification (i _n ) essentially regulates the control process via the track distance and the deviation of the nominal current specification (x _n ) from the actual value (x _a ) determines the position control for tracking.

7. Positioning method according to one of claims 4 to 6 using a positioning control unit (9) equipped with a digital signal processor (91), characterized in that first the calculation of the sample values (x _k ) of the setpoint curve, then the filtering of these sample values and finally the determination the nominal path and current specification (x _n or i _n ) is carried out during the positioning process under real-time conditions with the aid of the signal processor (91).

8. Positioning method according to claim 7, characterized in that the length of the first time interval (δt1) is derived from the respective track distance (x _s ) before starting each positioning process, from this taking into account the opposite in the first time interval and in the third time interval (δt3) supporting effective influence of the back emf of the drive (201) of the positioning unit (2) determines the length of this third time interval and finally the length of the second time interval (δt2) is determined such that it is the remaining difference between the full track distance and that in the first and Third time interval covered distances of the positioning unit compensated.

9. Positioning method according to one of claims 7 or 8, characterized in that within one of the time intervals (δt1, δt2 or δt3) for each sampling period (Δt) a new sample value (x _k ) of the setpoint curve by extrapolation from sample values (e.g. x _k-1 , X _{k -2} , x _k-3 ) of previous sampling periods is determined, and for this purpose, starting values corresponding to the time intervals and geometrically derived from the setpoint curve are specified as key parameters.

10. Positioning method according to one of claims 6 to 9, in which to filter the samples of the setpoint curve according to the order (n) of the selected FIR filter arrangement (n + 1) successive signal values (z. B. x _kn ... x _k ) are evaluated in each case predetermined by a corresponding to the transfer function of the FIR filter the filter coefficients (c _o ... c _n) and summed up, characterized in that for the filtering of the samples (x _k) instead of their absolute values, only the increment (z. B. x _k - x _k-1 ) to the previous sample (x _k-1 ) and that the filtered absolute size (x _nk ) from the corresponding filtered increment (δx _nk ) with addition to the absolute size (x _nk-1 ) of the previous sampling period ( Δt) is obtained.

11. Positioning method according to claim 10, in which the filtered sample values (x _nk ) derived from the setpoint curve represent the nominal path specification (x _n ), characterized in that to derive the corresponding nominal current specification (i _n ), the difference from the filtered increments (δx _nk , δx _nk-1 ) of the samples of the current or the previous sampling period (Δt) is formed.

12. Arrangement for performing the positioning method according to one of claims 1 to 11, characterized by an actual value transmitter (4) for deriving digital actual values (x _a ) from the sequence of analog position error signals (PES) output by the positioning unit (2), by a setpoint generator (e.g. 5), for a positioning process an address (ADR) of a target track or a variable corresponding to the distance between the starting track and the target track is supplied and which is constructed in such a way that it continuously derives a digitized value for the nominal path and current specification (x _n or i _n ) therefrom , by a digital comparator (e.g. 6), to which the actual value given by the actual value transmitter (x _a ) and that of the

Setpoint generator generated value of the nominal path specification (x _n ) are fed as input variables and which, as a control deviation, outputs a value corresponding to the difference between these input variables by an adder connected to the digital comparator (e.g. 7), which also has the digitized value for the nominal value Current specification (i _n ) is supplied and by a digital / analog converter (e.g. 8) which is connected to the output of the adder and the control signal (i _s (t)) for the drive (201) of the positioning unit ( 2) delivers.

13. The arrangement according to claim 12, characterized by an actual value transmitter (4) with a zero crossing detector (42) and an analog / digital converter (44) to which the position error signals (PES) are supplied in the form of a sawtooth pulse sequence, further with a down counter (41) , which can be loaded at the beginning of a positioning process with a digitized value (ADR) corresponding to the respective track distance and is triggered by output signals from the zero crossing detector, and with an adder (43) which is sent to both the outputs of the down counter and that of the analog / digital converter is connected and outputs the digitized actual value (x _a ) as the sum of its input variables.

14. Arrangement according to claim 12 or 13, characterized by a setpoint generator (5) with a memory (51) in which, for each track distance, a set of the basic values defining the course of the associated setpoint curve composed of linearly running sections (e.g. i ₀ , δ t2) and can be selected by an address (ADR) that corresponds to the respective track distance and is provided with switching networks (eg 511, 512, 513) Generation of consecutive, possibly linearly decremented, sample values in a predetermined time grid as a function of the basic values of the setpoint curve stored and read out in the switching networks, the setpoint generator also containing the FIR filter arrangement (52), to which these generated sample values (x _k ) are supplied.

15. An arrangement according to one of claims 12 to 14, in which the filtered sample values output by the FIR filter arrangement (52) represent a nominal current specification (i _n ), characterized by an integrator (52) consisting of two series-connected, in each case for the continuous addition fed back adders (530 or 531), which is connected to the output side of the FIR filter arrangement (52) and outputs the digitized nominal path specification (x _n ).

16. Arrangement according to one of claims 12 to 14, in which the filtered sample values output by the FIR filter arrangement (52) represent a nominal path specification (x _n ), characterized by a differentiating unit, consisting of two series-connected units, for continuous subtraction feedback subtraction stages, which is connected to the output side of the FIR filter arrangement (52) and outputs the digitized nominal current specification (i _n ).

17. Arrangement for performing the positioning method according to one of claims 7 to 11, characterized by an analog / digital converter (92) to which the position error signals (PES) emitted by the positioning unit (2) are supplied as a saw-shaped pulse sequence for digitization and by a Digital / analog converter unit (93) constructed from two digital / analog converters with outputs connected in parallel, via which the analog control voltage (i (t)) for the drive (201) of the positioning unit (2) can be output, one of the two Digital / analog converter with a gain higher than the other by a predetermined factor nominal current specification (i _n ) and the other part of this control signal (i _s (t)) corresponding to the control deviation of the actual position value (x _a ) from the nominal displacement specification (x _n ).