WO1983001157A1 - Efficient current modulator useful with inductive loads - Google Patents

Efficient current modulator useful with inductive loads Download PDF

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Publication number
WO1983001157A1
WO1983001157A1 PCT/US1982/001243 US8201243W WO8301157A1 WO 1983001157 A1 WO1983001157 A1 WO 1983001157A1 US 8201243 W US8201243 W US 8201243W WO 8301157 A1 WO8301157 A1 WO 8301157A1
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WO
WIPO (PCT)
Prior art keywords
coil
load
current
switching device
power supply
Prior art date
Application number
PCT/US1982/001243
Other languages
French (fr)
Inventor
Inc. Gould
Joe E. Deavenport
Original Assignee
Gould Inc
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Gould Inc filed Critical Gould Inc
Priority to AU89579/82A priority Critical patent/AU8957982A/en
Priority to GB08312199A priority patent/GB2116787A/en
Publication of WO1983001157A1 publication Critical patent/WO1983001157A1/en

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Classifications

    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03K—PULSE TECHNIQUE
    • H03K17/00—Electronic switching or gating, i.e. not by contact-making and –breaking
    • H03K17/51—Electronic switching or gating, i.e. not by contact-making and –breaking characterised by the components used
    • H03K17/56—Electronic switching or gating, i.e. not by contact-making and –breaking characterised by the components used by the use, as active elements, of semiconductor devices
    • H03K17/687—Electronic switching or gating, i.e. not by contact-making and –breaking characterised by the components used by the use, as active elements, of semiconductor devices the devices being field-effect transistors

Definitions

  • the present invention pertains to current modulators is particularly directed to enhancing the utility and improving the efficieny of current modulators that are used for modulatin the current flowing through a load from a DC power supply to circuit ground.
  • a switching device is connected between the load and circuit grou for varying the current through the load in response to a modu ⁇ lating signal applied to the switching device.
  • An overshoot clipping diode is connected between the switching device and th power supply side of the load for protecting the load and the switching device and the power supply side of the load for protecting the load and the switching device from excessively high currents.
  • Such prior art current modulator cannot be operated t produce an alternating current through an inductive load. Be- cause the voltage at the switching device side of the load woul exceed the power supply voltage during each half cycle, whereby the diode would become forward biased and effectively cause a sh circuit across the load.
  • the present invention provides an efficient current modulator that is useful with inductive loads.
  • the present invention is a current modulator for modu ⁇ lating the current flowing through a load from a DC power supply to circuit ground, including a switching device connected betwee the load and circuit ground for varying the current through the load in response to a modulating signal applied to the switching device; a first inductance coil connected between the switching device and the load; a second inductance coil mutually coupled to the first coil for receiving energy induced in the first coil by the current variations; and a diode coupled between the secon coil and thepower supply side of the load for transferring the received energy to the power supply side of the load.
  • the secon inductance coil is conductively isolated from the first inductan coil.
  • the current modulator of the present invention is more efficient.
  • the efficiency of -the current modulator is further in ⁇ creased by using a high frequency switching device. Less power lost in a high frequency switching device because it has shorter turn-on time.
  • the efficiency of the current modulator is still furth enchanced by providing that the mutually coupled inductance coil have an air core.
  • the combination of air core inductance coils switching at a high frequency reduces heat loss in the inductanc coils.
  • Figure 1 is a combination block and schematic circuit diagram of a preferred embodiment of the current modulator of t present invention.
  • Figure 2 is a schematic circuit diagram of the modula signal generator included in the current modulator of Figure 1.
  • Figures 3A through 3E illustrate waveforms of signals produced in the modulating signal generator of Figure 2.
  • Figure 4 is a combination block and schematic circuit diagram of another preferred embodiment of the current modulato of the present invention.
  • Figure 5 is a schemtatic circuit diagram of the modu ⁇ lating signal generator included in the current modulator of Figure 4. *
  • FIGS 6A through 6F illustrate waveforms of signals produced in tire modulating signal generator of Figure 5.
  • a preferred embodiment of the current modulator of the present invention includes a high freq switching device QI, a first inductance coil Ll, a second in ⁇ ductance coil L2, two diodes Dl and D2, three capacitances Cl, C and C3 and a modulating signal generator 10.
  • the switching devi Ql is a VMOS FET power transistor which is capable of switching at high frequencies.
  • the transistor Ql is connected between a load 12 and circuit ground for varying the current through the load 12 in response to a modulating signal applied on line 14 to the gate of the transistor Ql from the modulating signal generator 10.
  • a DC power supply terminal V g is coupled to the other side of the load 12 by a protective diode Dl.
  • the load 12 isolated from circuit ground by the capacitances Cl and C2.
  • the first inductance coil LI is connected between the transistor Ql and the load 12.
  • the second inductance coil L2 is mutually coupled to the first inductance coil Ll for receiving energy that is induced in the first inductance coil L by variations in the current through the load 12.
  • the first an second inductances Ll, L2 are tightly wound on an air-filled core by a bifilar, or preferably a coaxial winding.
  • the first and second inductances Ll, L2 are conductivel iso ated from each other, i.e. there is no DC current path between the coils Ll and L2. However, the coils Ll and L2 are coupled by the capacitance C3 to minimize any problems that cou arise due to poor mutual coupling- between the coils Ll and L2.
  • the diode D2 is connected between the second inductance coil L2 and the power supply side of the load 12 for transferri the received energy from the second coil L2 to the power supply side of the load 12.
  • the modulating signal generator 10 is adopted for apply a high frequency pulsed modulating signal to he gate of the switching transistor Ql.
  • the high frequency chosen in this preferred embodiment is 100 kHz.
  • a preferred embodiment of the modulating signal generat 10, as shown in Figure 2, includes a free running oscillator consisting of a Schmitt inverter II, a resistance Rl and a capacitance C4 for providing a high frequency pulsed square wav signal on line 16, as illustrated in Figure 3A.
  • Rl is.connected between the input and output of the inverter II the capacitance C4 is connected between the input of the invert II and circuit ground.
  • the modulating signal generator 10 further includes a differentiating circuit consisting of a capacitance C5 and a resistance R2; an npn transistor Q2; a current limiting resistance R3; a capacitance C6 and a comparator consisting of a high gain amplifier Al.
  • the square wave signal on line 16 is differentiated b the differentiator circuit C5, R2 to cause the transistor Q2 to be turned on each time there is a positive going transition in the square wave on line 16.
  • the capacitance C6 is discharged each time the transistor Q2 is turned on, but then is charged by the ' current through the resistance R3 from a DC supply volta terminal V_ while the transistor Q2 is turned off.
  • a resul a sawtooth waveform is provided on line 18 to one input termina of the comparator Al, as shown by the solid line in Figures 3B and 3D.
  • a low frequency reference signal V_ is applied to the other input terminal of the comparator Al for comparison with the sawtooth'signal on line 18.
  • the frequency of the reference signal V_ is chosen in accordance with what predetermined low frequency the current through the load 12 is to be varied. Typically, the predetermined low frequency is 50 or 60 Hz.
  • the amplitude of the reference signal V_ varies at th predetermined low frequency.
  • a pulsed high frequency modulating signal having a relatively short duty cycle (as shown in Figure 3C) is provided on line 14 from the output of the comparator Al to the gate of the switching transistor Ql ( Figure 1) .
  • a pulsed high frequency modulating signal having a relatively long duty cycle (as shown in Figure 3E) is provided on line 14 from the o put of the comparator Al.
  • the amplitude of the current through the load 12 is proportional to the amount of time that the switching transisto Ql is turned on, which is in turn proportional to the duty cycle of the pulsed high frequency modulating signal. Accordingly, th current through the load 12 is varied at the predetermined low frequency of the reference signal V R.
  • the inductance value of the first inductance coil Ll i chosen to have enough inductance to allow only negligible curren to build up when the pulsed modulating signal on line 14 has a minimum discrete duty cycle, and to allow a relatively large current to build up when the pulsed modulating signal on line 14 has a maximum discrete duty cycle.
  • Overshoot is minimized in the current modulator of the present invention by coupling induced energy from the first coil Ll to the second coil L2 and returning the energy received by the seco coil L2 back to the power supply side of the load 12 via the diod D2. If overshoot was not minimized, it typically would damage t switching transistor Ql and the first coil Ll.
  • the energy that returned back to the power supply side of the load 12 via the diode D2 reduces the current that must be supplied from the powe supply terminal V g through the diode Dl by an amount equal to the average current through the diode D2.
  • the average power lost in the transistor Ql may be minimized during its "on" time and if the transistor Ql is turned off with a very short turn-off time, th losses across the transistor Ql during turn-off may be kept quite small. Therefore, switching at a high frequency is prefe
  • the coils Ll and L2 have a very low resistance, whereby the heat losses in these coils Ll and L2 are quite smal
  • a preferred embodiment of the current modulator of th present invention that is particularly adapted for use with a transformer as a load is described with reference to Figures 4, and 6..
  • the load is the primary winding S of the transformer Tl is connected between an output terminal V ⁇ and circuit ground.
  • the current modulator includes first and second high frequency switching devices Qll and Q12. As in the preferred embodiment of Figure 1, the switching devices Qll and Q12 are
  • VMOS FET power transistors which are capable of switching at high. frequencies, such as 100 kHz.
  • the current modulator of Figure 4 further includes first, second, third and fourth inductance coils Lll, L12, L13 and L14, first and second diodes Dll and D12, capacitances Cll and C12 and a modulating signal generator 24.
  • the primary winding P has a center tap 26 that is connected to a DC power supply terminal V c .
  • the first switching transistor Qll is connected . betwe one side 28 of the primary winding P and circuit ground for ' varying the current through the primary winding P in response to the first modulating signal applied on line 30 to the gate of th switching transistor Qll from the modulating signal generator 24
  • the first inductance coil Lll is connected between the first transistor Qll and the one side 28 of the primary winding
  • the second inductance coil L12 is mutually coupled to the first inductance coil Lll for receiving energy induced in the first coil Lll by the current variations.
  • the first and second coils Lll, L12 are tightly wound on an air-filled core by a bifilar or preferably a coaxial winding.
  • the first and second coils Lll, L12 are conductively isolated from each other.
  • the first diode Dll is coupled between the second coil
  • the second switching transistor Q12 is connected betwe the other side 32 of the primary winding P and circuit ground for varying the current through the primary winding P in respons to a second modulating signal applied on line 34 to the gate of the second switching transistor Q12 from the modulating signal generator 24.
  • the third inductance coil Ll ' 3 is connected between the second transistor Q12 and the other side 32 of the primary winding P.
  • the fourth inductance coil L14 is mutually coupled to the third inductance coil L13 by the current variations.
  • the third and fourth coils L13, L14 are rightly wound on an air fill core by a bifilar or preferably a coaxial winding.
  • the third an fourth coil are conductively isolated from each other.
  • the second diode D12 is coupled between the fourth coi L12 and the one side 28 of the primary winding P for transferrin the received energy from the fourth coil L14 through the one sid 28 of the primary winding P to the power supply terminal V .
  • the modulating signal generator 24 is adapted for
  • first and second modulating signals are applied in such a manner as to cause only one side o the primary winding P to be connect j 5 to circuit ground at any one time.
  • I 24, as shown in Figure 5, includes an oscillator circuit con- i sisting of a first hysteresis inverter Gl, a resistance Rll j connected between the input and the output of the inverter Gl an : • 10 a capacitance Cll connected between the input of the inverter Gl ' and circuit ground.
  • the modulating signal generator 24 further includes a second hysteresis inverter G2 having its input connected to the output of the inverter Gl; an input amplifier A2, and 15 resistances R12, R13, R14 and R15.
  • A-reference signal terminal V R is connected through the resistance R12 to the inverting input terminal of the amplifierA2.
  • the inverting input terminal is also connected through the resistance R13 to a power supply terminal V_.
  • the non-inverting input terminal of the amplifier 20 A2 is connected to circuit ground.
  • The, resistance R14 is connec between the inverting input and the output of the amplifier A2.
  • the output of the amplifier A2 is connected through the resistanc R15 to the input of the first hysteresis inverter Gl.
  • the oscillator circuit Gl, Rll, Cll provides a square 25 wave pulsed signal on line 30, as shown in Figure 6A.
  • the inverter G2 inverts the signal on line 30 to provide a complement square wave pulsed signal on line 34. If the signals on lines 30 and 34 each have the same duty cycle, the average current through the primary winding P of the transformer Tl in the current 30 modulator of Figure 4 will be zero.
  • the duty cycle of the pulsed signal provided on lines and 34 from the respective outputs of the inverters Gl and G2 is varied by varying the rate at which the capacitance Cll is charged or discharged. This is accomplished by varying the current through the resistance R15 to the input terminal 36 of the inverter Gl. Such current variation is caused in response t a reference signal having a predetermined low frequency -that is applied to the reference signal terminal V R .
  • the frequency of reference signal is chosen in accordance with what predetermined low frequency the current through the primary winding P of the transformer Tl is to be varied. Typically, the predetermined low frequency is 50 or 60 Hz.
  • the amplifier A2 inverts the reference ' signal applied to the terminal V-. and causes a current to be provided through the resistance to R15 the inverter input terminal 36 that either aids or opposes the current through the resistance Rll in charging or discharging the capacitance Cll. '
  • the imbalance in the duty cycles of the pulsed modula signals applied to the respective gates of the switching transistors Qll and Q12 causes the average current flow through the primary winding P of the transformer Tl to be other than ze
  • the amount of current flow through the primary winding P is proportional to the difference ' between the duty cycles of the pulsed modulating signals applied on lines 30 and 36 to the respective gates of the switching transistors Ql and Q2.
  • the variation in the amplitude and direction of curre flow through the primary winding P is in response to the variation of instantaneous voltage at the amplifer output terminal 38 in the modulating signal generator 24.
  • a predetermined low frequency reference signal to the reference signal terminal V R , an output signal having the prefetermined low frequency is produced at the output termin V Q of the secondary winding of the transformer Tl.
  • the current modulator of Figure 4 is particularly useful for providing a low distortion sine wave low frequency output signal Basically, the operation and specifics of construction of the respective combinations of the switching transistor ,Q11, the inductance coils Lll, L12 and the diode Dll and the switching transistor Q12, the inductance coils L13, L14 and the diode D12 in the current modulator of Figure 4 are the same as those of the combination of the switching transistor Al, the inductance coils Ql, Q2 and the diode Dl in the current modulato of Figure 1.
  • a unique feature of the current modulator of Figure 4 is the use of the energy recovery path through the primary windi P to provide additional efficiency. This may be seen as follows If the two diodes, Dll and D12 had their cathodes connected to the power supply terminal V g . " The received energy from the coil L2 and L4 still would be returned to the power supply. However, when these two diodes Dll and D12 are cross-connected to the opposite sides 28, 32 of the primary winding P, as shown in
  • the recovered energy flows not only back to the power supply, but returns to the power supply through the opposite non-conducting side of the primary winding, and thereby provides added volt-ampere turns to the transformer Tl. Since the recovered energy from the collapsing field of the inductance coils tries to generate a high voltage "inductive kick" that is clipped by diodes Dll and D12, the current through these diodes Dll, D12 tends to be relatively constant for different voltages during the energy recovery period.

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
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Abstract

A current modulator for modulating the current flowing through a load (12) from a DC power supply (Vs) to circuit ground. A switching device (Q1) is connected between the load (12) and circuit ground for varying the current through the load (12) in response to a modulating signal (14) applied to the switching device (Q1). A first inductance coil (L1) is connected between the switching device (Q1) and the load (12). A second inductance coil (L2) is mutally coupled to the first coil (L1) for receiving energy induced in the first coil (L1) by the current variations. The second coil (L2) is conductively isolated from the first coil (L1). A diode (D2) is connected between the second coil (L2) and the power supply side of the load (12) for transferring the received energy back to the power supply side of the load (12).

Description

"EFFICIENT CURRENT MODULATOR USEFUL WITH INDUCTIVE LOADS" BACKGROUND OF THE INVENTION
The present invention pertains to current modulators is particularly directed to enhancing the utility and improving the efficieny of current modulators that are used for modulatin the current flowing through a load from a DC power supply to circuit ground.
In prior art current modulators of this type, a switching device is connected between the load and circuit grou for varying the current through the load in response to a modu¬ lating signal applied to the switching device. An overshoot clipping diode is connected between the switching device and th power supply side of the load for protecting the load and the switching device and the power supply side of the load for protecting the load and the switching device from excessively high currents.
Such prior art current modulator cannot be operated t produce an alternating current through an inductive load. Be- cause the voltage at the switching device side of the load woul exceed the power supply voltage during each half cycle, whereby the diode would become forward biased and effectively cause a sh circuit across the load.
SUMMARY OF THE INVENTION
The present invention provides an efficient current modulator that is useful with inductive loads.
-BϋRtA
OMPI The present invention is a current modulator for modu¬ lating the current flowing through a load from a DC power supply to circuit ground, including a switching device connected betwee the load and circuit ground for varying the current through the load in response to a modulating signal applied to the switching device; a first inductance coil connected between the switching device and the load; a second inductance coil mutually coupled to the first coil for receiving energy induced in the first coil by the current variations; and a diode coupled between the secon coil and thepower supply side of the load for transferring the received energy to the power supply side of the load. The secon inductance coil is conductively isolated from the first inductan coil. As a result when the diode becomes forward biased for transferring energy to the power supply side of the load, it nevertheless does not cause a short circuit across an inductive load.
Because energy is returned to the power supply side of the load by the diode, the current modulator of the present invention is more efficient. The efficiency of -the current modulator is further in¬ creased by using a high frequency switching device. Less power lost in a high frequency switching device because it has shorter turn-on time.
The efficiency of the current modulator is still furth enchanced by providing that the mutually coupled inductance coil have an air core. The combination of air core inductance coils switching at a high frequency reduces heat loss in the inductanc coils.
Additional features of the present invention are descr in the description of the preferred embodiments.
-Ψ i , °-MPI BRIEF DESCRIPTION OF THE DRAWINGS
Figure 1 is a combination block and schematic circuit diagram of a preferred embodiment of the current modulator of t present invention.
Figure 2 is a schematic circuit diagram of the modula signal generator included in the current modulator of Figure 1. Figures 3A through 3E illustrate waveforms of signals produced in the modulating signal generator of Figure 2. Figure 4 is a combination block and schematic circuit diagram of another preferred embodiment of the current modulato of the present invention.
Figure 5 is a schemtatic circuit diagram of the modu¬ lating signal generator included in the current modulator of Figure 4.*
Figures 6A through 6F illustrate waveforms of signals produced in tire modulating signal generator of Figure 5.
DESCRIPTION OF THE PREFERRED EMBODIMENTS
Referring to Figure 1, a preferred embodiment of the current modulator of the present invention includes a high freq switching device QI, a first inductance coil Ll, a second in¬ ductance coil L2, two diodes Dl and D2, three capacitances Cl, C and C3 and a modulating signal generator 10. The switching devi Ql is a VMOS FET power transistor which is capable of switching at high frequencies. The transistor Ql is connected between a load 12 and circuit ground for varying the current through the load 12 in response to a modulating signal applied on line 14 to the gate of the transistor Ql from the modulating signal generator 10. A DC power supply terminal Vg is coupled to the other side of the load 12 by a protective diode Dl. The load 12 isolated from circuit ground by the capacitances Cl and C2.
-BUREΛ
OΛ.PI The first inductance coil LI is connected between the transistor Ql and the load 12. The second inductance coil L2 is mutually coupled to the first inductance coil Ll for receiving energy that is induced in the first inductance coil L by variations in the current through the load 12. The first an second inductances Ll, L2 are tightly wound on an air-filled core by a bifilar, or preferably a coaxial winding.
The first and second inductances Ll, L2 are conductivel iso ated from each other, i.e. there is no DC current path between the coils Ll and L2. However, the coils Ll and L2 are coupled by the capacitance C3 to minimize any problems that cou arise due to poor mutual coupling- between the coils Ll and L2.
The diode D2 is connected between the second inductance coil L2 and the power supply side of the load 12 for transferri the received energy from the second coil L2 to the power supply side of the load 12.
The modulating signal generator 10 is adopted for apply a high frequency pulsed modulating signal to he gate of the switching transistor Ql. The high frequency chosen in this preferred embodiment is 100 kHz.
A preferred embodiment of the modulating signal generat 10, as shown in Figure 2, includes a free running oscillator consisting of a Schmitt inverter II, a resistance Rl and a capacitance C4 for providing a high frequency pulsed square wav signal on line 16, as illustrated in Figure 3A. The resistance
Rl•is.connected between the input and output of the inverter II the capacitance C4 is connected between the input of the invert II and circuit ground.
The modulating signal generator 10 further includes a differentiating circuit consisting of a capacitance C5 and a resistance R2; an npn transistor Q2; a current limiting resistance R3; a capacitance C6 and a comparator consisting of a high gain amplifier Al.
OMPI
. y ■ . — The square wave signal on line 16 is differentiated b the differentiator circuit C5, R2 to cause the transistor Q2 to be turned on each time there is a positive going transition in the square wave on line 16. The capacitance C6 is discharged each time the transistor Q2 is turned on, but then is charged by the' current through the resistance R3 from a DC supply volta terminal V_ while the transistor Q2 is turned off. As a resul a sawtooth waveform is provided on line 18 to one input termina of the comparator Al, as shown by the solid line in Figures 3B and 3D.
A low frequency reference signal V_ is applied to the other input terminal of the comparator Al for comparison with the sawtooth'signal on line 18. The frequency of the reference signal V_ is chosen in accordance with what predetermined low frequency the current through the load 12 is to be varied. Typically, the predetermined low frequency is 50 or 60 Hz.
The amplitude of the reference signal V_ varies at th predetermined low frequency. When the reference signal V has a relatively high amplitude (as shown in Figure 3B) , a pulsed high frequency modulating signal having a relatively short duty cycle (as shown in Figure 3C) is provided on line 14 from the output of the comparator Al to the gate of the switching transistor Ql (Figure 1) . When the reference signal has a relatively low amplitude (as shown in Figure 3D) , a pulsed high frequency modulating signal having a relatively long duty cycle (as shown in Figure 3E) is provided on line 14 from the o put of the comparator Al.
The amplitude of the current through the load 12 is proportional to the amount of time that the switching transisto Ql is turned on, which is in turn proportional to the duty cycle of the pulsed high frequency modulating signal. Accordingly, th current through the load 12 is varied at the predetermined low frequency of the reference signal V R.
Figure imgf000007_0001
The inductance value of the first inductance coil Ll i chosen to have enough inductance to allow only negligible curren to build up when the pulsed modulating signal on line 14 has a minimum discrete duty cycle, and to allow a relatively large current to build up when the pulsed modulating signal on line 14 has a maximum discrete duty cycle. When the current through the first inductance coil Ll is terminated during each pulse upon the switching transistor Ql being turned off, the stored energy in the two mutually coupled' inductance coils Ll and L2 results in a high positive voltage at point 20 on the switching transist side of the first coil Ll. This is commonly known as overshoot. Overshoot is minimized in the current modulator of the present invention by coupling induced energy from the first coil Ll to the second coil L2 and returning the energy received by the seco coil L2 back to the power supply side of the load 12 via the diod D2. If overshoot was not minimized, it typically would damage t switching transistor Ql and the first coil Ll. The energy that returned back to the power supply side of the load 12 via the diode D2 reduces the current that must be supplied from the powe supply terminal Vg through the diode Dl by an amount equal to the average current through the diode D2. In the current modulator of Figure 1, when the instantaneous voltage at point 22 on the switching transistor si of the load 12 exceeds the power supply voltage V_, the diode D2 does not short circuit the load 12 because the inductance coi Ll and L2 are not conductively coupled to each other, whereby there is no DC current path from the point 22 through the diode D2 back to the power supply side of the load 12. As a result, the instantaneous voltage at the point 22 may swing to as much as twice the power supply voltage Vς.
As the switching transistor Ql is switched on with a short turn-on time, there will be minimum loss in the transistor Ql during the turn-on time because there is low current in the inductance Ll for a fast step change. If the saturation voltage
-~ -~ on the transistor Ql is low, the average power lost in the transistor Ql may be minimized during its "on" time and if the transistor Ql is turned off with a very short turn-off time, th losses across the transistor Ql during turn-off may be kept quite small. Therefore, switching at a high frequency is prefe The coils Ll and L2 have a very low resistance, whereby the heat losses in these coils Ll and L2 are quite smal
Typically, the current modulator of Figure 1 will deliver
96 percent of input power into the load with four percent being dissipated in the combination of the coils Ll and L2, the switching transistor Ql and the diode D2.
A preferred embodiment of the current modulator of th present invention that is particularly adapted for use with a transformer as a load is described with reference to Figures 4, and 6.. The load is the primary winding S of the transformer Tl is connected between an output terminal V~ and circuit ground.
The current modulator includes first and second high frequency switching devices Qll and Q12. As in the preferred embodiment of Figure 1, the switching devices Qll and Q12 are
VMOS FET power transistors which are capable of switching at high. frequencies, such as 100 kHz.
The current modulator of Figure 4 further includes first, second, third and fourth inductance coils Lll, L12, L13 and L14, first and second diodes Dll and D12, capacitances Cll and C12 and a modulating signal generator 24. The primary winding P has a center tap 26 that is connected to a DC power supply terminal Vc.
The first switching transistor Qll is connected.betwe one side 28 of the primary winding P and circuit ground for' varying the current through the primary winding P in response to the first modulating signal applied on line 30 to the gate of th switching transistor Qll from the modulating signal generator 24
_ OMPI The first inductance coil Lll is connected between the first transistor Qll and the one side 28 of the primary winding
The second inductance coil L12 is mutually coupled to the first inductance coil Lll for receiving energy induced in the first coil Lll by the current variations. The first and second coils Lll, L12 are tightly wound on an air-filled core by a bifilar or preferably a coaxial winding. The first and second coils Lll, L12 are conductively isolated from each other. The first diode Dll is coupled between the second coil
L12 and the other side 32 of the primary winding P for trans¬ ferring-the received energyfrom the second coil Ll-2 through the other side 32 of the primary winding P to the power supply terminal Vς. The second switching transistor Q12 is connected betwe the other side 32 of the primary winding P and circuit ground for varying the current through the primary winding P in respons to a second modulating signal applied on line 34 to the gate of the second switching transistor Q12 from the modulating signal generator 24.
The third inductance coil Ll'3 is connected between the second transistor Q12 and the other side 32 of the primary winding P.
The fourth inductance coil L14 is mutually coupled to the third inductance coil L13 by the current variations. The third and fourth coils L13, L14 are rightly wound on an air fill core by a bifilar or preferably a coaxial winding. The third an fourth coil are conductively isolated from each other.
The second diode D12 is coupled between the fourth coi L12 and the one side 28 of the primary winding P for transferrin the received energy from the fourth coil L14 through the one sid 28 of the primary winding P to the power supply terminal V .
SURE
_OMP
A. wip i " The modulating signal generator 24 is adapted for
• applying the first and second modulating signals to the j respective gates of the switching transistors Qll and Q12. The
* first and second modulating signals are applied in such a manner as to cause only one side o the primary winding P to be connect j 5 to circuit ground at any one time.
I A preferred embodiment of the modulating signal genera
I 24, as shown in Figure 5, includes an oscillator circuit con- i sisting of a first hysteresis inverter Gl, a resistance Rll j connected between the input and the output of the inverter Gl an :• 10 a capacitance Cll connected between the input of the inverter Gl ' and circuit ground.
' The modulating signal generator 24 further includes a second hysteresis inverter G2 having its input connected to the output of the inverter Gl; an input amplifier A2, and 15 resistances R12, R13, R14 and R15. A-reference signal terminal VR is connected through the resistance R12 to the inverting input terminal of the amplifierA2. The inverting input terminal is also connected through the resistance R13 to a power supply terminal V_. The non-inverting input terminal of the amplifier 20 A2 is connected to circuit ground. The, resistance R14 is connec between the inverting input and the output of the amplifier A2. The output of the amplifier A2 is connected through the resistanc R15 to the input of the first hysteresis inverter Gl.
The oscillator circuit Gl, Rll, Cll provides a square 25 wave pulsed signal on line 30, as shown in Figure 6A. The inverter G2 inverts the signal on line 30 to provide a complement square wave pulsed signal on line 34. If the signals on lines 30 and 34 each have the same duty cycle, the average current through the primary winding P of the transformer Tl in the current 30 modulator of Figure 4 will be zero.
BVR
OM W When the output of the inverter Gl on line 30 switches high, the capacitance Cll is charged through the resistance Rll to provide a signal at the input terminal 36 of the inverter Gl having a waveform as shown in Figure 6B. When the voltage of the signal at input terminal 36 rises to the upper switching threshold V, of the inverter Gl, the output of the inverter Gl switches low as shown in Figure 6A. The capacitance Cll is then discharged through the resistance Rll until the voltage of the signal at the input terminal 36 drops to the lower switching threshold V. of the inverter Gl, at which point, the inverter Gl again switches high.
The duty cycle of the pulsed signal provided on lines and 34 from the respective outputs of the inverters Gl and G2 is varied by varying the rate at which the capacitance Cll is charged or discharged. This is accomplished by varying the current through the resistance R15 to the input terminal 36 of the inverter Gl. Such current variation is caused in response t a reference signal having a predetermined low frequency -that is applied to the reference signal terminal VR. The frequency of reference signal is chosen in accordance with what predetermined low frequency the current through the primary winding P of the transformer Tl is to be varied. Typically, the predetermined low frequency is 50 or 60 Hz. *
The amplifier A2 inverts the reference' signal applied to the terminal V-. and causes a current to be provided through the resistance to R15 the inverter input terminal 36 that either aids or opposes the current through the resistance Rll in charging or discharging the capacitance Cll.'
When the voltage at the output terminal 38 of the amplifier A2 is negative with respect to the average value of -the voltage at the inverter input terminal 36, the capacitanc Cll is charged slowly and discharged quickly, as shown in Figure 6C. This causes the first inverter output signal on line 30 to have a long duty cycle, as shown in Figure 6D. Conversely, the pulsed output signal of the second inverter G2 simultaneously has a short duty cycle. When the voltage at the output terminal 38 of the amplifier A2 is positive with respect to the average value of t voltage at the inverter input terminal 36, the capacitance Cll is charged quickly and discharged slowly, as shown in Figure 6E This causes the first inverter output signal on line 30 to have a short duty cycle as shown in Figure 6F. Conversely, the pulsed output signal of the second inverter G2 simultaneously h a long duty cycle.
The imbalance in the duty cycles of the pulsed modula signals applied to the respective gates of the switching transistors Qll and Q12 causes the average current flow through the primary winding P of the transformer Tl to be other than ze The amount of current flow through the primary winding P is proportional to the difference' between the duty cycles of the pulsed modulating signals applied on lines 30 and 36 to the respective gates of the switching transistors Ql and Q2.
Current flow through the primary winding P is greater in the direction from the center tap 26 toward the one side 28 when the duty, cycle of the pulsed modulating signal on line is greater than the duty cycle of the pulsed modulating signal line 34.
The variation in the amplitude and direction of curre flow through the primary winding P is in response to the variation of instantaneous voltage at the amplifer output terminal 38 in the modulating signal generator 24. Thus by providing a predetermined low frequency reference signal to the reference signal terminal VR, an output signal having the prefetermined low frequency is produced at the output termin VQ of the secondary winding of the transformer Tl. Accordingly, the current modulator of Figure 4 is particularly useful for providing a low distortion sine wave low frequency output signal Basically, the operation and specifics of construction of the respective combinations of the switching transistor ,Q11, the inductance coils Lll, L12 and the diode Dll and the switching transistor Q12, the inductance coils L13, L14 and the diode D12 in the current modulator of Figure 4 are the same as those of the combination of the switching transistor Al, the inductance coils Ql, Q2 and the diode Dl in the current modulato of Figure 1.
A unique feature of the current modulator of Figure 4 is the use of the energy recovery path through the primary windi P to provide additional efficiency. This may be seen as follows If the two diodes, Dll and D12 had their cathodes connected to the power supply terminal Vg." The received energy from the coil L2 and L4 still would be returned to the power supply. However, when these two diodes Dll and D12 are cross-connected to the opposite sides 28, 32 of the primary winding P, as shown in
Figure 4, then the recovered energy flows not only back to the power supply, but returns to the power supply through the opposite non-conducting side of the primary winding, and thereby provides added volt-ampere turns to the transformer Tl. Since the recovered energy from the collapsing field of the inductance coils tries to generate a high voltage "inductive kick" that is clipped by diodes Dll and D12, the current through these diodes Dll, D12 tends to be relatively constant for different voltages during the energy recovery period.

Claims

1. A current modulator for modulating the current flowing through a load from a DC power wupply to circuit grou comprising a switching device connected between the load and circuit ground for varying the current through the load in response to a modulating signal applied to the switching devi a first inductance coil connected between the switc device and the load; a second inductance coil mutually coupled to the fi coil for receiving energy induced in the first coil b said current variations, the second coil being conductively isolat from the first coil; and a diode coupled between the second coil and the pow supply side of the load for transferring the received energy the power supply side of the load.
2. A current modulator according to claim 1, where switching device is a high frequency switching device.
3. A current modulator according to claim 2, furth comprising means for applying a high frequency pulsed modulati signal to the switching device.
4. A current modulator according to claim 3, where the switching device is a MOS semiconductor device.
5. A current modulator according to claim 4, wherei the switching device is a VMOS FET power transistor.
.. OM
6. A current modulator according to claims 3 or 5," wherein the means for applying the modulating signal to the switching device comprises first means for generating a high frequency pulsed signal; and second means for varying the duty cycle of the pulsed signal at a predetermined low frequency to thereby provide the high frequency pulsed modulating signal for varying the current through the load at the predetermined low frequency.
7. A current modulator according to claim 6, wherein the second means is adapted for varying said duty cycle in response to a reference signal having said predetermined 'low frequency.
8. A current modulator according to claim 3, wherein the inductance coils have an air core.
9. A current modulator for modulating the current flowing through a load to circuit ground from a DC power supply connected to the load, comprising a first switching device connected between one side of the load and circuit ground for varying the current through the load in response to a first modulating signal applied to the first switching device; a first inductance coil connected between the first switching device and the one side of the load; a second inductance coil mutually coupled to the firs coil for receiving energy induced in the first coil by said current variations, the second coil being conductively isolated from the first coil; a first diode coupled between the second coil and the power supply for transferring the received energy from the second coil to the power supply; a second switching device connected between the other side of the load and circuit ground for varying the current through the load in response to a second modulating signal applied to the second switching device; a third inductance coil connected between the second switching device and the other side of the load; a fourth inductance coil mutually coupled to the thir coil for receiving energy induced in the third coil by said current variations, the fourth coil being conductively isolated from the third coil; a second diode coupled between the fourth coil and the power supply for transferring the received energy from the fourth .coil to the power supply; and means for applying the modulating signals to the switching devices to cause only one side of the load to be connected to ground at any one time.
10. A current regulator according to claim 9, wherein the first diode is coupled between the second coil and the other side of the load for transferring the received energy from the second coil through the other side of the load to the power supply; and wherein the second diode is coupled between the fourt coil and the one side of the load for transferring the received energy from the fourth coil through the one side of the load to the power supply.
11. A current regulator according to claim 10, for use with a load having a center tap, further comprising means for connecting the center tap to the power supply.
12. A current regulator according to claim 10, wherei the switching devices are high frequency switching devices.
13. A current modulator according to claim 12, wherein the switching devices are MOS semiconductor devices.
14. A current modulator according to claim 13, wherei the switching device is a VMOS FET power transistor.
15. A current modulator according to claim 12, wherei the means for applying the modulating signals .to the switching devices are adapted for applying high frequency pulsed modulating signals.
16. A current modulator according to claims 14 or 15, wherein the means for applying the modulating signals to the switching devices comprises a first means for generating first and second high frequency pulsed signals that are complementary to each other; a second means for varying the duty cycles of the first and second high frequency pulsed signals at a predetermined low frequency to thereby provide the first and second modulating signals for varying the current through the load at the pre- determined low frequency.
BURE
OMP
17. A current modulator according to claim 16, wher the second means is adapted for varying said duty cycles in response to a reference signal having said predetermined low frequency.
18. A current modulator according to claim 15, wher the inductance coils have air cores.
OMPI wipo
PCT/US1982/001243 1981-09-16 1982-09-14 Efficient current modulator useful with inductive loads WO1983001157A1 (en)

Priority Applications (2)

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AU89579/82A AU8957982A (en) 1981-09-16 1982-09-14 Efficient current modulator useful with inductive loads
GB08312199A GB2116787A (en) 1981-09-16 1982-09-14 Efficient current modulator useful with inductive loads

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US30249281A 1981-09-16 1981-09-16
US302,492810916 1981-09-16

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FR (1) FR2513045A1 (en)
GB (1) GB2116787A (en)
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US5982639A (en) * 1997-11-04 1999-11-09 Power Integrations, Inc. Two switch off-line switching converter
US6107851A (en) * 1998-05-18 2000-08-22 Power Integrations, Inc. Offline converter with integrated softstart and frequency jitter
US6226190B1 (en) 1998-02-27 2001-05-01 Power Integrations, Inc. Off-line converter with digital control
US6876181B1 (en) 1998-02-27 2005-04-05 Power Integrations, Inc. Off-line converter with digital control
US7719243B1 (en) 2007-11-21 2010-05-18 Fairchild Semiconductor Corporation Soft-start system and method for power converter
US7723972B1 (en) 2008-03-19 2010-05-25 Fairchild Semiconductor Corporation Reducing soft start delay and providing soft recovery in power system controllers
US7872883B1 (en) 2008-01-29 2011-01-18 Fairchild Semiconductor Corporation Synchronous buck power converter with free-running oscillator
US8018694B1 (en) 2007-02-16 2011-09-13 Fairchild Semiconductor Corporation Over-current protection for a power converter
US9484824B2 (en) 2005-08-26 2016-11-01 Power Integrations, Inc. Method and apparatus for digital control of a switching regulator

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US4692643A (en) * 1983-10-28 1987-09-08 Hitachi, Ltd. Semiconductor switching device having plural MOSFET's, GTO's or the like connected in series
SE461626B (en) 1988-07-06 1990-03-05 Philips Norden Ab POWER SUPPLY CIRCUIT IN MICROWAVE OVEN

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* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US5982639A (en) * 1997-11-04 1999-11-09 Power Integrations, Inc. Two switch off-line switching converter
US6005781A (en) * 1997-11-04 1999-12-21 Power Integrations, Inc. Two switch off-line switching converter
US7477534B2 (en) 1998-02-27 2009-01-13 Power Integrations, Inc. Off-line converter with digital control
US7248029B2 (en) 1998-02-27 2007-07-24 Power Integrations, Inc. Off-line converter with digital control
US6226190B1 (en) 1998-02-27 2001-05-01 Power Integrations, Inc. Off-line converter with digital control
US6297623B1 (en) 1998-02-27 2001-10-02 Power Integrations, Inc. Off-line converter with digital control
US6414471B1 (en) 1998-02-27 2002-07-02 Power Integrations, Inc. Off-line converter with digital control
US6608471B2 (en) 1998-02-27 2003-08-19 Power Integrations, Inc. Off-line converter with digital control
US6747444B2 (en) 1998-02-27 2004-06-08 Power Integrations, Inc. Off-line converter with digital control
US6876181B1 (en) 1998-02-27 2005-04-05 Power Integrations, Inc. Off-line converter with digital control
US7038439B2 (en) 1998-02-27 2006-05-02 Power Integrations, Inc. Off-line converter with digital control
US6229366B1 (en) 1998-05-18 2001-05-08 Power Integrations, Inc. Off-line converter with integrated softstart and frequency jitter
US6107851A (en) * 1998-05-18 2000-08-22 Power Integrations, Inc. Offline converter with integrated softstart and frequency jitter
US9484824B2 (en) 2005-08-26 2016-11-01 Power Integrations, Inc. Method and apparatus for digital control of a switching regulator
US10224820B2 (en) 2005-08-26 2019-03-05 Power Integrations, Inc. Method and apparatus for digital control of a switching regulator
US8018694B1 (en) 2007-02-16 2011-09-13 Fairchild Semiconductor Corporation Over-current protection for a power converter
US7719243B1 (en) 2007-11-21 2010-05-18 Fairchild Semiconductor Corporation Soft-start system and method for power converter
US7872883B1 (en) 2008-01-29 2011-01-18 Fairchild Semiconductor Corporation Synchronous buck power converter with free-running oscillator
US7723972B1 (en) 2008-03-19 2010-05-25 Fairchild Semiconductor Corporation Reducing soft start delay and providing soft recovery in power system controllers

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GB2116787A (en) 1983-09-28
GB8312199D0 (en) 1983-06-08
FR2513045A1 (en) 1983-03-18
JPS58501492A (en) 1983-09-01
IT8249128A0 (en) 1982-09-16

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