US8363517B1

US8363517B1 - Optical storage system having differential phase detector

Info

Publication number: US8363517B1
Application number: US13/356,954
Authority: US
Inventors: Christopher Painter; Jin Xie
Original assignee: Marvell International Ltd
Current assignee: Cavium International; Marvell Asia Pte Ltd
Priority date: 2007-07-05
Filing date: 2012-01-24
Publication date: 2013-01-29
Anticipated expiration: 2028-07-07
Also published as: US8107329B1; US8089836B1; US8879372B1; US8634284B1

Abstract

A differential phase detector for an optical storage system is set forth. The differential phase detector includes a photodetector circuit arranged to detect light deviations associated with radial errors in the optical storage system. A non-linear equalizer is in communication with the photodetector circuit. The output of the non-linear equalizer is in communication with signal processing circuitry. The signal processing circuitry uses the equalized signals to generate one or more radial error signals.

Description

RELATED APPLICATION

This application is a continuation of U.S. application Ser. No. 12/168,594, filed Jul. 7, 2008 (now U.S. Pat. No. 8,107,329), which claims the benefit of U.S. Provisional Application No. 60/948,068, filed Jul. 5, 2007. The contents of U.S. application Ser. No. 12/168,594 (now U.S. Pat. No. 8,107,329), and U.S. Provisional Application No. 60/948,068 are each incorporated by reference in their entirety.

BACKGROUND

Optical storage systems are used in a wide range of applications. Such optical storage systems employ optical media (i.e., CDs, DVD discs, Blue Ray discs, HD discs, and other media) that stores data along tracks disposed radially about the disc-shaped media. The data may be in the form of reflective pits disposed in the media. The reflective pits are detected/written by optical read/write components that are moved radially to different tracks of the storage media.

Differential phase detection may be used during the reading of data from optical storage media. The differential phase detection generates a radial error signal that drives a radial control loop to properly align the optical read components with the track that is to be read. The differential phase detection signal may also be used when moving to a new radial location on the storage media (i.e., seeking). During such an operation, the differential phase detection signal may be post processed, along with other information, by a circuit known as a track counter. The track counter facilitates monitoring of the gross radial position of the optical read components.

A differential phase detector may use a photodetector array to determine whether the optical read components are aligned with the desired track. The photodetector array is used to measure a time-varying diffraction pattern. The physical dimensions of the features on the disc are comparable to the wavelength of the light used to read the data, and the photodetector array that processes the light reflected from the disc is illuminated by a diffraction pattern. The characteristics of this diffraction pattern are influenced by the radial position, and by the particular pattern of recorded data. When an objective lens of the optical read components is in alignment with a track centerline, the electrical signals generated by the elements of the photodetector are in phase. When there is a radial position error, however, the diffraction pattern rotates about the optical axis as a particular data bit is scanned.

The electrical signals provided by the photodetector array are communicated to a signal processing circuit. The signal processing circuit may include a phase detector that compares the phase relationship of the electrical signals. The phase relationship is used to generate the track error signal.

As the data density of optical media storage increases, detection of electrical signals from the photodetectors becomes more difficult. As a result, detection circuits for such high-density media may become more complicated and costly to implement. Further, phase detectors used in the signal processing circuitry may be subject to noise signals that inhibit the proper detection of the phase relationship of the electrical signals. This may result in corresponding errors in the radial error signal.

SUMMARY

The present invention is defined by the following claims, and nothing in this section should be taken as a limitation on those claims.

By way of introduction, the preferred embodiments described below provide a differential phase detector for use in an optical storage system. In one preferred embodiment, the differential phase detector includes a photodetector circuit arranged to detect light deviations associated with radial errors in the optical storage system. A non-linear equalizer is in communication with the photodetector circuit. The output of the non-linear equalizer is in communication with signal processing circuitry. The signal processing circuitry uses the equalized signals to generate one or more radial error signals. In another preferred embodiment, the non-linear equalizer includes a non-linear circuit in cascade with one or more linear filters. In another preferred embodiment, the non-linear circuit is a slicer and/or a non-linear circuit having a sigmoid transfer function. Other preferred embodiments are provided, and each of the preferred embodiments described herein can be used alone or in combination with one another.

The preferred embodiments will now be described with reference to the attached drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of a system used to read an optical storage disk.

FIG. 2 is a block diagram of one embodiment of a signal processing circuit that may be used in the system shown in FIG. 1.

FIG. 3 shows one embodiment of an equalizer that may be used in the signal processing circuit shown in FIG. 2.

FIG. 4 shows an exemplary transfer characteristic that may be associated with a non-linear circuit.

FIG. 5 shows another exemplary transfer characteristic that may be associated with a non-linear circuit.

FIG. 6 is a graph that compares an input signal of the equalizer shown in FIG. 3 with its corresponding output signal.

FIG. 7 is a block diagram of a first embodiment of a phase detector.

FIG. 8 is a timing diagram showing the signals used in the phase detector of FIG. 7.

FIG. 9 is a further timing diagram showing the signals used in the phase detector of FIG. 7.

FIG. 10 is a block diagram of a second embodiment of a phase detector.

FIG. 11 is a timing diagram showing the signals used in the phase detector of FIG. 10.

FIG. 12 is a block diagram of a third embodiment of a phase detector.

FIG. 13 is a timing diagram showing the signals used in the phase detector of FIG. 12.

FIG. 14 is a block diagram of a fourth embodiment of a phase detector.

FIG. 15 is a timing diagram showing signals used in the phase detector of FIG. 14.

FIG. 16 is a further timing diagram showing signals used in the phase detector of FIG. 14.

FIG. 17 is a functional block diagram of a digital versatile disk (DVD);

FIG. 18 is a functional block diagram of a high definition television;

FIG. 19 is a functional block diagram of a vehicle control system;

FIG. 20 is a functional block diagram of a cellular phone;

FIG. 21 is a functional block diagram of a set top box; and

FIG. 22 is a functional block diagram of a media player.

DETAILED DESCRIPTION

FIG. 1 is a block diagram of a system 100 used to read an optical storage disk. System 100 includes signal processing circuitry 105 in communication with electrical signals provided by a photodetector array 110. The signal processing circuitry 105 includes equalization circuitry that increases the sensitivity of the system 100 to low-amplitude, high-frequency electrical signals that are characteristic of high data density optical discs. Further, the signal processing circuitry 105 includes phase detection circuits that have substantial immunity to noise signals.

In system 100, optics and tracking circuits 115 are used to read data from a track of the optical disc as the optics and tracking circuits 115 move in alignment with a track in the direction shown by arrow 120. Optical signals 125 indicative of any radial error are communicated for detection by the photodetector array 110. Although other photodetector arrangements may be used, system 100 uses a four-quadrant photodetector array having a photodetector A, B, C, and D, arranged in each quadrant of a plane.

Optical signals

125 indicate when optics and tracking circuits 115 are out of alignment with track 120. When out of alignment, the optical signals introduce a delay between the electrical signals produced by elements A and B, and a similar delay between the electrical signals from elements C and D. Since these delays correspond to the amount of radial displacement, they may be used to generate a radial error signal. In FIG. 1, radial errors in the direction of arrow 130 cause the signals generated by photodetectors A and D to lead the signals generated by photodetectors B and C. Similarly, radial errors in the direction of arrow 135 cause signals generated by photodetectors B and C to lead the signals generated by photodetectors A and D.

Although other differential phase detection methods may be used, system 100 uses DTD4 (Differential Time Detection, Type 4). In a DTD4 structure, the radial error signal RE is calculated by passing the sum of the measured delays τ(A,B) and τ(C,D) through a lowpass filter. That is, the radial error is
RE=LPF[τ(A,B)+τ(C,D)]
where τ(x,y) represents the normalized delay between the signals x and y, and LPF represents a lowpass filtering operation. The measured delays may be normalized to the data rate of the readback signal, and therefore to the rotational rate of the disc. Consequently, the scaling of the radial error signal is invariant with respect to the rotational rate of the disc.

FIG. 2 is a block diagram of one embodiment of the signal processing circuitry 105 shown in FIG. 1. As shown, the electrical signals from photodetectors A through D are communicated to respective equalizers 205 through 220. Equalizers 205 through 220 may each be constructed in the same fashion. Consequently, the phase relationships of output signals of through df are maintained so that they correspond to the phase relationships of the input signals A through D.

The output signals af through df and their inverted counterparts, are provided as input signals to a plurality of phase detectors 225 through 240. Phase detector 225 is in communication with signals af and bf of

equalizer

205 and 210, respectively. Phase detector 230 is in communication with signals af and bf of

equalizer

205 and 210 through

inverters

245 and 250, respectively. Phase detector 235 is in communication with signals cf and df of

equalizer

215 and 220, respectively. Phase detector 240 is in communication with signals cf and df of

equalizer

215 and 220 through

inverters

255 and 260, respectively.

In this configuration, the phase detectors 225 through 240 measure different phase relationships between output signals af through df. Phase detector 225 measures the phase relationship between the rising edge of signal af and the rising edge of signal bf. Phase detector 230 measures the phase relationship between the falling edge of signal af and the falling edge of signal bf. Phase detector 235 measures the phase relationship between the rising edge of signal cf and the rising edge of signal df. Phase detector 240 measures the phase relationship between the falling edge of signal cf and the falling edge of signal df.

The output of the phase detectors 225 through 240 are provided to the input of a summing circuit 265. The resulting summation signal is provided to the input of post-processing circuitry 270 to generate a radial error signal 275. The post-processing circuitry 270 may include one or more filters to low-pass filter the summation signal provided by summation circuitry 265.

FIG. 3 illustrates one embodiment of an equalizer 305 that may be used to implement the equalizers 205 through 220 shown in FIG. 2. In order to increase the sensitivity of the signal processing circuitry 105 to the high data signal rates and/or high data densities associated with high density optical storage disks, equalizer 305 includes a non-linear circuit that, for example, may be memoryless.

In the embodiment shown in FIG. 3, equalizer 305 includes a linear filter 310 that is in communication with one of the electrical signals from photo detectors A through D. The output of linear filter 310 is provided to the input of a non-linear circuit 315. The output of non-linear circuit 315, in turn, is provided to the input of another linear filter 320. The output of linear filter 320 corresponds to one of the output signals of through df shown in FIG. 2.

Linear filter

310 includes a cascade arrangement of a high pass filter 325 followed by a low pass filter 330. High pass filter 325 assists in eliminating residual DC offset or low-frequency excursions of the electrical signals provided by photodetectors A through D thereby ensuring correct detection of zero crossings for the signals in subsequent processing. High pass filter 325 may have double real poles at −1/(k₁T) Hz, where T corresponds to the bit interval of a readback signal for the optical storage medium that is read, and where k₁corresponds to a constant that is chosen depending on the disk format of the optical storage medium. The value for k₁may be approximately 8 for a standard definition DVD format, and about 5 for a high-definition disk format. More particularly, the high pass filter 325 may have a transfer function corresponding to the following:
H(s)=s ²/(s+2π/k ₁ T))².

Low pass filter

330 may have two real poles at −1/(k₂T) Hz, where T corresponds to the bit interval of the readback signal for the optical storage medium that is read, and where k₂corresponds to a constant that is chosen depending on the disk format of the optical storage medium. The value for k₂may be approximately 4 for a standard definition DVD format, and about 3 for a high-definition disk format. More particularly, the low pass filter 330 may have a transfer function corresponding to the following:

H (s) = \frac{1}{{[s \frac{k_{2} T}{2 π} + 1]}^{2}} .

Filter

330 assists in attenuating high-frequency noise that might otherwise decrease the reliability of delay measurements.

The non-linear circuit 315 that follows the low pass filter 330 may be memoryless and implemented as an analog comparator (slicer). Such a circuit is characterized by a very high gain near its switching threshold, and sharply defined positive and negative saturation levels. When the signal provided by lowpass filter 330 to the non-linear circuit 315 is slightly greater than a predefined threshold voltage, the output signal of non-linear circuit 315 saturates at a voltage that is positive with respect to the datum (zero level) of the overall nonlinear equalizer circuit. When the input signal provided by lowpass filter 330 to the non-linear circuit 315 is slightly negative with respect to this threshold voltage, the output signal of the non-linear circuit 315 saturates at a level that is negative with the respect to the datum.

FIGS. 4 and 5 show exemplary transfer characteristics that may be associated with the non-linear circuit 315. FIG. 4 shows an ideal slicer characteristic where the output voltage Vout of the slicer transitions to a high voltage saturation level Vsh when the voltage input Vin goes above zero. Similarly, the output voltage Vout of the slicer transitions to a low voltage saturation level Vsl when the voltage input Vin falls below zero.

FIG. 5 shows a further transfer characteristic that may be associated with non-linear circuit 315. In FIG. 5, the transfer characteristic has a large slope, dvout/dvin, proximate the origin and reaches a high saturation voltage Vsh when the input voltage Vin is proximate a first threshold. The output voltage Vout reaches a low saturation voltage Vsl when the input voltage Vin is proximate a second threshold. The slope, dvout/dvin, may be about 4. Overall, the transfer characteristic shown in FIG. 5 has a sigmoid shape with soft knee breaks at the output saturation voltages.

Linear filter

320 includes a low pass filter 335 that is in communication with the output signal from the non-linear circuit 315. The low pass filter may have a single real pole at −1/(k₃T) where T corresponds to the bit interval of the readback signal for the optical storage medium that is read, and where k₃corresponds to a constant that is chosen depending on the disk format of the optical storage medium. The value for k₃may be approximately 4 for a standard definition DVD format, and about 3 for a high-definition disk format. More particularly, the low pass filter 335 may have a transfer function corresponding to the following:

H (s) = \frac{1}{s \frac{k_{3} T}{2 π} + 1}

The non-linear circuit 350 and low pass filter 335 act together to boost the amplitude of short duration pulses associated with certain optical disk formats and readers. This operation is shown in FIG. 6 which compares input signal A and output signal af of equalizer 305 to one another.

As noted, the output signals from the equalizers 205 through 220 are provided to the inputs of phase detectors 225 through 240 to provide signals corresponding to the phase differences between various combinations of signals af through df. FIG. 7 shows one embodiment of a phase detector 700 that may be used in phase detectors 225 through 240. Although the phase detector 700 shown in FIG. 7 is used to compare the phase difference between the rising edge of signal af and the rising edge of signal bf, the same phase detection circuitry may be used to compare the phase difference between any two signals.

As shown, signal af is provided to slicer 705 to generate output signal afl while signal bf is provided to slicer 710 to generate output signal bfl. Signals afl and bfl are provided to the clock inputs of flip-

flops

715 and 720, respectively. The output signals QA and QB provided by 715 and 720 correspond to the phase difference between signals afl and bfl and are in communication with the inputs of AND

gates

725 and 730 to generate phase signals Aout and Bout. Phase signals Aout and Bout are provided to a summation circuit 735 to provide a difference signal at output 740.

Phase detector 700 also includes other components that remove the “memory” capability associated with typical phase/frequency detectors. By removing this “memory” capability, the frequency detection associated with typical phase/frequency detectors is effectively removed. This allows the phase detector 700 to provide a correct indication of phase difference even when an aberration is present in one of the signals af, bf. The circuitry used to remove the “memory” capability may include NOR gate 745, AND gate 750, OR gate 755, and the

corresponding delay lines

760 and 765.

An aberration in a signal corresponds to a deviation from an expected shape of the signal. For example, an aberration in a signal can be a high signal during a time in which a low signal is expected or vice versa. An aberration in a signal can come from various sources, such as, for example, a noise glitch (i.e., a false or spurious signal caused by a brief unwanted surge of power), coupling between clock or pulse signals of the phase detection circuit, and/or ground bouncing.

A timing diagram showing the various signals of the phase detector 700 is shown in FIG. 8. Multiple phase detection cycles are shown in which a phase comparison between the rising edges (or falling edges depending on the desired phase detection operation) of two signals is executed. Signal Aout corresponds to the amount by which the leading edge of signal afl leads the leading edge of signal bfl. The spikes shown in signal Bout are of relatively short duration and do not have a substantial influence on the ultimate measurement of the phase difference between afl and bfl.

Although the phase detector 700 provides substantial immunity to aberrations in the phase detector signals, it may be sensitive to aberrations occurring during a falling edge of one or more of the input signals. The response of the phase detector 700 to an aberration 905 at a falling edge of signal bfl is shown in FIG. 9. In FIG. 9, there should be a positive going pulse in Aout during the time interval from 1.445 μs to 1.545 μs. However, the pulse is missing. This is due to the following:

- At about 1.33 μs, signal bfl has a falling edge that is followed by a noise spike. The rising edge of the noise spike triggers flip-flop 720 so that signal QB goes to a positive state.
- Signal QA is not asserted at this time, and afl and bfl are low. Consequently, the reset signals rsta and rstb are high thereby allowing signal QB to remain in a positive state.
- At about 1.44 μs, afl has a rising edge. This edge triggers flip-flop 715 so that QA is driven positive state.
- When both QA and QB are in a positive state, the reset signals rsta and rstb are driven to a low state, and both QA and QB are cleared.
- Since QA is not asserted, no pulse appears at Aout.

FIG. 10 shows a first embodiment of a phase detector 1000 that substantially eliminates lost pulses caused by aberrations at the falling edge of the input signals to the phase detector logic. In FIG. 10, signals of and bf are in communication with the inputs of

slicers

1005 and 1010 to generate signals afl and bfl. Signals afl and bfl are in communication with the inputs of AND gate 1015, OR gate 1020, and XOR gate 1025. The output of AND gate 1015 is in communication with delay line 1030 that, in turn, provides a clock signal clk to a flip-flop 1035. The output of OR gate 1020 provides a reset signal rst that is in communication with the reset input of flip-flop 1035. The inverted output of flip-flop 1035 is in communication with a delay line 1040 to generate signal Qbar. Signal Qbar, in turn, is in communication with an input of AND gate 1045. The output signal XORout of XOR gate is in communication with another input of AND gate 1045, the output signal of which is designated ANDout. Signal ANDout is in communication with an input of AND gate 1050 as well as AND gate 1055. Signal afl is provided to another input of AND gate 1050 to facilitate generation of signal Aout. Similarly, signal bfl is in communication with another input of AND gate 1055 to facilitate generation of signal Bout. Signals Aout and Bout are in communication with the inputs of a summation circuit 1060 to generate a difference signal at line 1065.

The operation of phase detector 1000 may be understood with reference to the timing diagram shown in FIG. 11. The operation of this logic can be summarized as follows:

- The exclusive OR of the equalized and sliced signals afl and bfl, denoted XORout has the desired pulse width corresponding to the phase difference between the rising edge of signal afl and the rising edge of signal bfl. However, an additional pulse having a width corresponding to the phase difference between the falling edge of afl and bfl is also present. This additional pulse is unwanted.
- The flip-flop 1035 is used to generate a window that may be used to gate the XORout signal between the rising edge of afl and the rising edge of bfl. The window may be generated with a single flip-flop.
  As shown in FIG. 11, the desired positive going pulse 1105 in Aout from 1.445 μs to 1.545 μs is present despite the presence of the aberration 1110 at the falling edge of signal bfl that occurs during the immediately preceding phase detection cycle.

FIG. 12 shows a second embodiment of a phase detector 1200 that substantially eliminates lost pulses caused by aberrations at the falling edge of the input signals to the phase detector logic. Phase detector 1200 operates to clear signal QB after it is triggered by an aberration at the falling edge of signal bfl.

Phase detector

1200 is similar to phase detector 700 with several exceptions. Signal afl is in communication with an input of a delay line 1205 to generate a clock signal aclk for flip-flop 1210. Similarly, signal bfl is in communication with an input of a delay line 1215 to generate a clock signal bclk for flip-flop 1220. The output of NOR gate 1225 is in communication with an input of AND gate 1230. Signal afl is in communication with another input of AND gate 1230. The output signal rsta of AND gate 1230 is in communication the reset input of flip-flop 1210. The output of NOR gate 1225 is also in communication with an input of AND gate 1235. Signal bfl is in communication with another input of AND gate 1235. The output signal rstb of AND gate 1235 is in communication the reset input of flip-flop 1220.

The operation of phase detector 1200 may be summarized as follows:

- From analysis of phase detector 700, the pulse omission is exacerbated when signal QB stays high for a long time. Therefore, it is desirable to clear signal QB after it is triggered by an aberration.
- Signal QB may be cleared whenever bfl is low. It is assumed that aberrations have a short duration. As such, signal QB is cleared substantially immediately after the occurrence of the aberration. Useful pulses in signal bfl are long, so when signal QB is triggered by a useful pulse, signal bfl stays high and QB is not cleared.
- This analysis also applies for signal QA and leads to the design shown in FIG. 12. Delay line 1205 for signal afl is to assure that signal rsta is steadily high at a rising edge of signal aclk.
  Waveforms of the same input stimuli shown in FIG. 9 are shown in FIG. 13. An aberration 1305 occurs proximate a falling edge of signal bfl. Nevertheless, the desired positive going pulse 1310 in Aout from 1.445 μs to 1.545 μs is present.

FIG. 14 shows a third embodiment of a phase detector 140 that substantially eliminates lost pulses caused by aberrations at the falling edge of the input signals to the phase detector logic. In FIG. 14, signal of is in communication with slicer 1405 to generate signal afl while signal bf is in communication with slicer 1410 to generate signal bfl. Signal afl is in communication with the data input of flip-flop 1415 and an input of XOR gate 1420. Signal afl is also in communication with another input of XOR gate 1420 through delay line 1425. Together, signal afl and the delayed version of signal afl are used to generate a signal aclk at the output of XOR gate 1420. Signal aclk is inverted by NOT gate 1430 and used to generate the clock signal to flip-flop 1415.

Signal bfl is in communication with the data input of flip-flop 1435 and an input of XOR gate 1440. Signal bfl is also in communication with another input of XOR gate 1440 through delay line 1445. Together, signal bfl and the delayed version of signal bfl are used to generate a signal bclk at the output of XOR gate 1440. Signal bclk is inverted by NOT gate 1450 and used to generate the clock signal to flip-flop 1415.

Elimination of the “memory” effect in phase detector 1400 is facilitated by the communications between AND

Gates

1455 and 1460, and NOR gate 1465. The output signals QA and QB of flip-

flops

1415 and 1435, respectively, are in communication with the inputs of a summing circuit 1465 that provides a difference signal at line 1470.

The operation of phase detector 1400 may be described with reference to the timing diagram of FIGS. 15 and 16 as follows:

- Signal bclk is generated as the exclusive OR of signal bfl and the delayed version of signal bfl. FIG. 15 shows the waveforms of the same input stimuli as FIG. 8 having aberrations 1501 proximate a falling edge of signal bfl. FIG. 16 is an exploded version of FIG. 15. In FIGS. 15 and 16, signal bclk has rising edges after the aberrations 1501.
- At the last rising edge of signal bclk, signal bfl is already stable at a low level. Consequently, signal QB is triggered by the last rising edge of bclk to ensure that signal QB only has a short duration. Signal QB is therefore in an asserted state for a short time and does not inhibit generation of signal QA during the subsequent phase detection cycle.
  The same analysis also applies to signal aft.

Referring now to FIGS. 17 through 21, various exemplary uses of the present invention are shown. As shown in FIG. 17, the present invention can be implemented in a digital versatile disk drive 1710. More particularly, the optical storage medium 1716 may include a differential phase detector and/or phase detector circuit such as those described above. The output of the storage medium 1716 may be in communication with the input of a DVD signal processing and/or control circuit 1712 having a processed output signal at line 1717. DVD signal processing and/or control circuit 1712 may also be in communication with mass data storage 1718 and memory 1719. The digital versatile disk system 1710 may be powered by power supply 1713. The signal processing and/or control circuit 1712 and/or other circuits (not shown) in the DVD 1710 may process data, perform coding and/or encryption, perform calculations, and/or format data that is read from and/or data written to the optical storage medium 1716. In some implementations, the signal processing and/or control circuit 1712 and/or other circuits (not shown) in the DVD 1710 can also perform other functions such as encoding and/or decoding and/or any other signal processing functions associated with a DVD drive.

The DVD drive 1710 may communicate with an output device (not shown) such as a computer, television or other device via one or more wired or wireless communication links 1717. The DVD 1710 may communicate with mass data storage 1718 that stores data in a nonvolatile manner. The mass data storage 1718 may include a hard disk drive (HDD). The HDD may be a mini HDD that includes one or more platters having a diameter that is smaller than approximately 1.8″. The DVD 1710 may be connected to memory 1719 such as RAM, ROM, low latency nonvolatile memory such as flash memory and/or other suitable electronic data storage.

Referring now to FIG. 18, the present invention can be implemented in a high definition television (HDTV) 1820. The differential phase detectors and/or phase detectors described above may be used in an optical storage system that is used to implement the mass data storage 1827. The HDTV 1820 receives HDTV input signals in either a wired or wireless format and generates HDTV output signals for a display 1826. In some implementations, signal processing circuit and/or control circuit 1822 and/or other circuits (not shown) of the HDTV 1820 may process data, perform coding and/or encryption, perform calculations, format data and/or perform any other type of HDTV processing that may be required.

The HDTV 1820 may communicate with mass data storage 1827 that stores data in a nonvolatile manner, such as an optical storage device. At least one DVD may have the configuration shown in FIG. 17. The HDTV 1820 may be connected to memory 1828 such as RAM, ROM, low latency nonvolatile memory such as flash memory and/or other suitable electronic data storage. The HDTV 1820 also may support connections with a WLAN via a WLAN network interface 1829 and be powered by power supply 1823.

Referring now to FIG. 19, the present invention may be implemented in a mass data storage 1946 of a vehicle control system 1930. Control system 1930 may include a powertrain control system 1932 in communication with the mass data storage 1946, memory 1947, and a WLAN interface 1948. Powertrain control system 1932 may interface with the vehicle through sensors 1936 and outputs 1938. In some implementations, the control system 1932 receives inputs from sensors 1936 such as temperature sensors, pressure sensors, rotational sensors, airflow sensors and/or any other suitable sensors and/or generate output control signals 1938 such as engine operating parameters, transmission operating parameters, and/or other control signals. Control system 1930 may also include other vehicle control systems 1940 that interface with the vehicle through sensors 1942 and outputs 1944. In some implementations, the control system 1940 may be part of an anti-lock braking system (ABS), a navigation system, a telematics system, a vehicle telematics system, a lane departure system, an adaptive cruise control system, a vehicle entertainment system such as a stereo, DVD, compact disk and the like. Power for the control system 1930 may be provided by power supply 1933. Still other implementations are contemplated.

Referring now to FIG. 20, the present invention can be implemented in the mass data storage 2064 of a cellular phone 2050 that may include a cellular antenna 2051. In some implementations, the cellular phone 2050 includes a microphone 2056, an audio output 2058 such as a speaker and/or audio output jack, a display 2060 and/or an input device 2062 such as a keypad, pointing device, voice actuation and/or other input device. The signal processing and/or control circuits 2052 and/or other circuits (not shown) in the cellular phone 2050 may process data, perform coding and/or encryption, perform calculations, format data and/or perform other cellular phone functions.

The cellular phone 2050 may communicate with mass data storage 464 that stores data in a nonvolatile manner, such as in optical storage device. The cellular phone 2050 may be connected to memory 2066 such as RAM, ROM, low latency nonvolatile memory such as flash memory and/or other suitable electronic data storage. The cellular phone 2050 also may support connections with a WLAN via a WLAN network interface 2068. Power to the cellular phone 2050 may be provided by power supply 2053.

Referring now to FIG. 21, the present invention can be implemented in the mass data storage 2190 of a set top box 2180. The set top box 2180 receives signals from a source such as a broadband source and outputs standard and/or high definition audio/video signals suitable for a display 2188 such as a television and/or monitor and/or other video and/or audio output devices. The signal processing and/or control circuits 2184 and/or other circuits (not shown) of the set top box 2180 may process data, perform coding and/or encryption, perform calculations, format data and/or perform any other set top box function.

The set top box 2180 may communicate with mass data storage to 190 that stores data in a nonvolatile manner. The mass data storage to 190 may include optical storage devices that include differential phase detectors and/or phase detectors such as those described above. The set top box 1480 may be connected to memory 1494 such as RAM, ROM, low latency nonvolatile memory such as flash memory and/or other suitable electronic data storage. The set top box 1480 also may support connections with a WLAN via a WLAN network interface 1496. Power for the set top box 2180 may be provided by power supply 2183.

Referring now to FIG. 22, the present invention can be implemented in a mass data storage 2210 of a media player 2200. In some implementations, the media player 2200 includes a display 2207 and/or a user input 2208 such as a keypad, touchpad and the like. In some implementations, the media player 2200 may employ a graphical user interface (GUI) that employs menus, drop down menus, icons and/or a point-and-click interface via the display 2207 and/or user input 2208. The media player 2200 further includes an audio output 2209 such as a speaker and/or audio output jack. The signal processing and/or control circuits 2204 and/or other circuits (not shown) of the media player 2200 may process data, perform coding and/or encryption, perform calculations, format data and/or perform any other media player function.

The media player 2200 may communicate with mass data storage 2210 that stores data such as compressed audio and/or video content in a nonvolatile manner. In some implementations, the compressed audio files include files that are compliant with MP3 format or other suitable compressed audio and/or video formats. The mass data storage 2210 may include optical storage devices that include differential phase detectors and/or phase detection circuits of the type described above. The media player 2200 also may support connections with a WLAN via a WLAN network interface 2216. Still other implementations in addition to those described above are contemplated.

It is intended that the foregoing detailed description be understood as an illustration of selected forms that the invention can take and not as a definition of the invention. It is only the following claims, including all equivalents, that are intended to define the scope of this invention.

Claims

1. Signal processing circuitry for an optical storage system comprising:

non-linear equalizer circuitry configured to receive a signal from a photodetector circuit, the signal indicative of one or more errors in the optical storage system, the non-linear equalizer circuitry being configured to non-linearly equalize the signal from the photodetector circuit; and

post-processing circuitry in communication with the non-linear equalizer circuitry to generate one or more error signals based on the non-linearly equalized signal,

wherein the non-linear equalizer circuitry comprises a non-linear circuit in cascade with at least one of a first linear filter or a second linear filter.

2. The signal processing circuitry of claim 1, where the non-linear circuit comprises a memory-less non-linear circuit in cascade with the first linear filter and the second linear filter.

3. The signal processing circuitry of claim 1, where the non-linear circuit has an approximately sigmoid transfer characteristic.

4. The signal processing circuitry of claim 1, where the first linear filter comprises:

a high pass filter configured to receive the signal from the photodetector circuit; and

a first low pass filter in communication with the high pass filter;

where the nonlinear circuit comprises a memory-less non-linear circuit in communication with the low pass filter; and

where the second linear filter comprises a second low pass filter in communication with the memory-less non-linear circuit.

5. The signal processing circuitry of claim 4, where the high pass filter has double real poles at −1/(k₁T) Hz, where T corresponds to the bit interval of a readback signal for an optical storage medium that is read, and where k₁corresponds to a constant that is chosen depending on a disk format of the optical storage medium.

6. The signal processing circuitry of claim 5, wherein the high pass filter has a transfer function comprising:

H(s)=s ²/(s+(2π/k ₁ T))².

7. The signal processing circuitry of claim 4, wherein the low pass filter has a transfer function comprising:

H (s) = \frac{1}{{[s \frac{k_{2} T}{2 π} + 1]}^{2}}

wherein T corresponds to the bit interval of a readback signal for an optical storage medium that is read, and wherein k₂corresponds to a constant that is chosen depending on a disk format of the optical storage medium.

8. The signal processing circuitry of claim 4, where the second low pass filter has a transfer function comprising:

H (s) = \frac{1}{s \frac{k_{3} T}{2 π} + 1}

wherein T corresponds to the bit interval of a readback signal for an optical storage medium that is read, and wherein k₃corresponds to a constant that is chosen depending on a disk format of the optical storage medium.

9. The signal processing circuitry of claim 1, wherein the signal is a differential signal comprising a first signal and second signal, the non-linear equalizer circuitry being configured to non-linearly equalize the first and second signals, the signal processing circuitry further comprising:

a differential phase detector that is configured to receive the non-linearly equalized first and second signals from the non-linear equalizer circuitry and generate an output signal indicative a phase relationship between the non-linearly equalized first and second signals,

wherein the post-processing circuitry is configured to receive the output signal from the differential phase detector and generate the one or more error signals based on the output signal.

10. The signal processing circuitry of claim 9, wherein the differential phase detector is configured to measure a phase relationship between a rising edge of the non-linearly equalized first signal and a rising edge of the non-linearly equalized second signal, or a phase relationship between a falling edge of the non-linearly equalized first signal and a falling edge of the non-linearly equalized second signal to generate the output signal.

11. The signal processing circuitry of claim 1, wherein the one or more errors in the optical storage system are indicative of a misalignment between tracking circuitry of the optical storage system and a track of an optical disc.

12. The signal processing circuitry of claim 1, wherein the signal is indicative of light deviations associated with the one or more errors in the optical storage system.

13. A method for executing differential phase detection in an optical storage system comprising:

receiving a first signal and a second signal from a photodetector circuit, the first and second signals indicative of one or more errors in the optical storage system;

non-linearly equalizing the first and second signals to generate non-linearly equalized first and second signals;

linearly filtering the first signal and the second signal at least one of:

before the first and second signals are non-linearly equalized; or

after the first and second signals are non-linearly equalized; and

generating one or more error signals based on the non-linearly equalized first and second signals.

14. The method of claim 13, wherein non-linearly equalizing the first and second signals comprises processing the first and second signals with a non-linear circuit having an approximately sigmoid transfer characteristic.

15. The method of claim 13, where linearly filtering the first and second signals before the first and second signals are non-linearly equalized comprises:

high pass filtering the generated electrical signals; and

low pass filtering signals that have been subject to the high pass filtering;

where non-linearly equalizing the first and second signals comprises processing the low pass filtering signals through a memory-less non-linear circuit; and

where linearly filtering the first and second signals after the first and second signals are non-linearly equalized comprises low pass filtering the non-linearly equalized first and second signals processed by the memory-less nonlinear circuit.

16. The method of claim 15, where the high pass filtering has double real poles at −1/(k₁T) Hz, where T corresponds to the bit interval of a readback signal for an optical storage medium that is read, and where k₁corresponds to a constant that is chosen depending on a disk format of the optical storage medium.

17. The method of claim 13, further comprising:

measuring a phase relationship between the non-linearly equalized first and second signals,

wherein generating the one or more error signals comprises generating the one or more or more error signals based on the measured phase relationship.

18. The method of claim 17, wherein measuring the phase relationship between the non-linearly equalized first and second signals comprises:

measuring a phase relationship between a rising edge of the non-linearly equalized first signal and a rising edge of the non-linearly equalized second signal, or a phase relationship between a falling edge of the non-linearly equalized first signal and a falling edge of the non-linearly equalized second signal to generate the output signal.

19. The method of claim 13, wherein the one or more errors in the optical storage system are indicative of a misalignment between tracking circuitry of the optical storage system and a track of an optical disc.

20. The method of claim 13, wherein the first and second signals are indicative of light deviations associated with the one or more errors in the optical storage system.