US3588598A - Lighting-control systems - Google Patents

Abstract

A LIGHTING CONTROL SYSTEM HAS ELECTRONICALLY TRIGGERED SWITCHING DEVICES CONTROLLING THE CURRENT SUPPLY TO A LIGHTING LOAD AND A FEEDBACK CIRCUIT FOR MODIFYING THE CONTROL CHARACTERISTICS OF THE SWITCHING DEVICES IN DEPENDENCE ON THE MEAN VALUE IN ANY HALF CYCLE OF THE VOLTAGE BETWEEN THE LOAD TERMINALS.

Description

O United States Patent 11113,588,598

' l [72] lnventor Anthony L. Issues [50] Field of Search 3l5/l94, London, England 199, 100 (D), 31 l; 323/22 (CR); 307/252 [2!] Appl. No. 695,202 [22] Filed Jan. 2, 1968 [56] References Clted [45] Patented June 28, 1971 UNITED STATES PATENTS 1 Assisnee Thom Electriwl Industries Limited, 3,193,728' 7/1965 Skirpan 315/194 I London, England 3,249,799 5 1966 Powell 315 311 1 Prwmy y 15,1967 3,414,766 12/1968 Miller.... 315/194 [33] Great Britain l 3 l] 2'2535/67 Primary Exammer.Jen'y D. Craig Att rneys-Norman J. O'Malley and Laurence Burns 4 ABSTRACT: A lighting control system has electronically trig- LIGHTING'CON'FROI, SYSTEMS ge red switching devices controlling the current supply to a 4 Claims 10 Drawmg lighting load and a feedback circuit for modifying the control [52] U.S. Cl .t 315/199, Characteristics of the switching devices in dependence on the 307/252, 315/31 1 mean value in any half cycle of the voltage between the load [51 1 Int. Cl H05b 37/02 terminals.

R3 R5 R5 CS Y J T S a/1% 5 l L I B v w c3? m E T Y X P]- J VT? 1,

1 C T R7 A mi 9T F 5cm L5 ,SCRZ

g1 La (5) A. c. l; R2 SUPPLY T T I L PATENTED JUN28 I97| SHEET 1 0T 7 PRIOR ART SCRZ Fig.1.

'EziL A L SUPPLY //v TRIGGER GLN'L'R G C T CONTROL SIGNAL IN PRIOR ART TRIGGER PULSES PATENTED JUN28 19m SHEET 2 BF 7 6 7 m w m w m w .r O O O 0 O W 0 5 NF 5 M31 m W m HF?! O F w 0 E 5 2 m la O A 6 O O w M w M m l. O O O O 8 R .J O w M R F P A NORMAL/ZED CONTROL VOLTS U MK Rg RDRNDO QwMQQTSQQQ PRIOR ART ANTHONY LEONARD ISAACS QQ M BY ATTORNEY PATENTEUJUN28IBZI 3,588,588

swam? ID i 3 W 0' 0-30: V 2 y E 7 0-10 h g vb Q u 0-032 PRIOR g ART =2 NORMAL/ZED CONTROL VOLTS U' I SCRI I ST\ ISCAZ ACSUPPL) w I A? L INVENTORV ANTHONY LEONARD ISAACS BY MONEY PATENTEU JUN28 1971 SHEET '4 OF 7 PRIOR ART INVENTOR ANTHONY LEONARD ISAACS ATTORNEY PATENTED JUN28 m1 SHEET 5 BF 7 3 5GB IE8 8358 PUEJ (.0 INVENTOR ANTHONY LEONARD ISAACS ATTORNEY PATENTEU JUN28 |97| SHEET 7 [IF 7 Fig.9.

a 8 q APSE 3225? NORMAL/ZED CONTROL VOLTAGE V.

INVE NTOR ANTHONY LEONARD ISAACS BY W W/ ATTORNEY LIGHTING-CONTROL SYSTEMS The present invention relates to lighting control systems.

In the specifications of British Pat. applications Nos. 47341/66 and 47344/66 there are disclosed novel means for controlling the lighting of stages in the theatre and TV studio, such means including dimmers controlling the light outputs of groups of lamps in response to analogue voltage signals. The present invention is especially concerned with such voltagecontrolled dimmers.

It is generally agreed by those skilled in the art of stage lighting that the light output of lamps for such use should vary as the square of the dimmer control setting, i.e. if the dimmer control is a linear voltage control producing a normalized output signal Actual control voltage Actual light output Maxmium light output then L should vary as v. The required range of stage lighting levels is large, particularly in producing such dramatic effects as fades between full light output and blackout; the required light output range may be 500: l for TV studio purposes and 10,000: I for the theatre stage. We find in fact that the generally preferred square law relationship between control signal and light output is applicable only in the upper two decades of normalized light output, i.e. from L=l.O to L=0.00l, v=l.0 to v=O.l; below v=O.I the preferred square law characteristic is too coarse for satisfactory control either manually or by a control signal varying linearly with time, as does the preferred analogue signal in the above-mentioned Pat. Specification No. 47344/66. We find that for a control signal v having an effective range of 1000: l, i.e. 3 decades, the power of v with which L varies must be reduced drastically in all but the upper decade of v; thus for TV purposes requiring a 500:1 range of normalized light output L we may have L v for l.() 1 0.l. but L u for 0.l v ().0l and L v for 0.01 v 0.00 l, and for theatre purposes requiring a l0,000:l range of light output we may have L v for:l.0 v 0.l, L v for O.l v 0.0l and L v for 0.00l v 0.000l. Thus voltage-controlled dimniers such as those of the aforesaid British Pat. Specifications No. 47341/66 and No. 43744/66 may operate to produce a normalized light output proportional to the square of the normalized control voltage v in the topmost decade of the control voltage range, l.O v 0.l, but in lower decades of v the law should be suited to the total range of normalized light output required. I

The lamps used for stage lighting are commonly incandescent lamps having tungsten filaments. In a kw. lainp for such use the normalized light output Actual light output Maximum rated light output may vary as R for l.O R BO. l ,where Actual RMS lamp volts Maximum rated RMS lamp volts trigger-pulse timing in the circuit of FIG. I and the output waveform,

FIGS. 3, 4a and 4b are further graphs illustrating the operation of the known circuit of FIG. 1,

FIG. 5 is a circuit diagram of another known lighting control system,

FIG. 6 contains graphs illustrating the operation of the known circuit of FIG. 5,

FIG. 7 shows a circuit diagram of asimple embodiment of the present invention,

FIG. 8 shows a circuit diagram of the, at present, preferred embodiment of the invention, and

FIG. 9 contains curves illustrating the performance obtainable with the circuit of FIG. 8.

In a known method of controlling AC power, thyratrons or thyristors (silicon controlled rectifiers, SCRs) or bidirectional thyristors (triacs) are used to control the angle of flow of current and hence to control the power in the lead. Thus in FIG. I thyristors SCRl and SCR2 are connected in parallel with one another in inverse sense and the parallel pair is connected in series between the supply terminals ST and the load L, in this case assumed to be a tungsten filament lamp. No power is delivered to the load L until suitable trigger pulses are applied to the control electrodes or gates of the thyristors. A trigger pulse is generated at a controlled point in each supply half cycle by a trigger generator G and is applied between gate and cathode of each thyristor by means of separate secondary windings S1, S2 on a pulse transformer PT. Resistors R1, R2 are included to swamp changes or differences in the gate input impedances of the thyristors. The anode of one or other thyristor is forward biased in each supply half cycle and this thyristoris triggered into conduction by the trigger pulse at its gate; its conduction ceases when the supply polarity reverses at the end of that supply half cycle. The angle of flow of current in the load is determined by the timing of the trigger pulses within the half cycles of the supply. This timing may be varied, for example, by a control signal applied at terminals CT operating upon suitable circuits in the trigger generator G.

The relationship between the trigger pulse timing and the output voltage or current waveform is shown in FIG. 2 at (a) and (b). If this the phase delay in radians between the start of a supply half cycle and the start of the trigger pulse at (b) in that half cycle the angle of current flow in a resistive loadis 9=1r radians. It can be shown that the normalized mean value M of the load voltage in any supply half cycle is given by Actual mean load volts Maximum mean load volts Sin 20 A tual RMS load volts Maximum RMS lpad volts 1r If the normalized angle of current flow 91ris made equal in any supply half cycle to the normalized control voltage v so that 9=7r when v=l, then M and R vary with v as shown in FIG. 3. Comparison of the curves for R and v in FIG. 3 shows that for control of tungsten filament lamps the 25 percent of the control signal range is ineffective; also it can be shown that for v 0.5, R v, i.e. if L R, L v; the midrange control is too coarse. Hence linear control of angle of current flow is unsuited to our purpose.

Better control is afforded if the normalized mean output voltage M is made equal to the normalized control voltage v; the normalized RMS output voltage R then varies as shown in FIG. 4(a), which shows N also for comparison; M, R, andW are replotted in FIG. 4(b) using logarithmic scales.

R is asymptotic to 11 for v 0.3

so that ifL R, L v for v 0.3.

Open-loop control means are known whereby the normalized mean output voltage M in any half cycle from a thyristorcontrolled supply may be made linearly proportional to the normalized control voltage v. Such open-loop means are FIGS. 6(a), (b) and (c) occurring at correspondingly referenced parts of the circuit of FIG. 5. in H0. 5 a transformer T and rcctifiers B produce an output having waveform V of FIG. 6(a). This is clipped by a resistor R3 and a loner diode ZD to provide the interrupted signal W of FIG. 6(a). The signal W is applied to the base 2 terminal of a unijunction transistor UJT through a current-limiting resistor Rd and the base ll terminal of the unijunction transistor is connected to the V rail through the primary winding of the pulse transformer PT. The emitter of the unijunction transistor is conneeted to a capacitor C which is charged rapidly through a diode D at the start of a supply half cycle by the signal X of FIG. Mb), appearing at the slider of a potential divider VlRh. A high value resistor R causes the signal V to charge the capacitor C, adding a cosine component, namely the waveform Y of HG. Mb), to the potential on it. The unijunction transistor UJT has the property that when the base 2 is biased to the potential W a fraction 1,W of this potential appears in the base region opposite the emitter. If the emitter voltage is less than 1 W, the emitter-base junction is reversebiased and no emitter current flows. If the emitter potential exceeds 1 W, emitter current flows and the unijunction transistor conducts heavily between the emitter and the base 1, discharging the capacitor C through the primary of the pulse transformer PT and producing pulses Z of FIG. Me) at its secondary terminals. If the current in the resistor Rd is less than the sustaining current (the "valley" current) of the unijunction transistor, this ceases to conduct when the capacitor C is discharged, and C recharges through the diode D and resistor R5. A sequence of trigger pulses may be produced in each half cycle of the supply, but only the first trigger pulse in each half cycle is needed to fire a thyristor; the thyristor then remains conducting until the supply polarity inverts at the end of the supply half cycle. Output current flow through the lamp L is as indicated in MG. 6(d).

it can be shown that if the peak value of V is V, if the supply period is To=llf0 and if T CR,,, where C and R represent the values of the capacitor and resistor respectively, then M varies linearly from 0 to l as X varies from "r nlV t0 nlV r. .15

If such thyristor dimmer systems as are shown in H6. it or FIG. 5 are used to control stage lighting in TV studios, the power controlled may be of the order of KM) kw. When triggered, thyristors develop full conduction in about ins. The fast on-going edges of such large switched powers can give rise to large induced and radiated fields which can interfere severely with studio microphone circuits. To minimize such effects it is necessary to include series inductance in the control system to limit rates of change of current.

In a system such as that of FIG. 11, a series inductor may be connected between the thyristors and the load or between the thyristors and the supply. Because of its interlayer capacitances a thyristor may be triggered accidentally by capacitive coupling of fast-rising anode voltage transients to its gate electrode. In general the user has more control over load transients than over supply transients, so that the inductor may be connected advantageously between the thyristors and the supply, where it protects the thyristors from false triggering by supply transients.

The junction areas and thermal capacitances of thyristors are small; thyristors may be destroyed within milliseconds of switching into a short-circuited load. Mechanical circuit breakers are in general too slow in operation to give effective protection; fast-blowing fuses may give adequate protection with constant-resistance loads but are impracticable where it is required to switch tungsten filament lamps from cold. The cold resistance ofa tungsten filament lamp is well below its resistance at full power; if switched from cold to full power, a Skw. or l0kw. tungsten lamp may draw an initial current l0 times its normal current at full power, and this may take l00ms. or 250ms. to fall to 30 percent of its initial value. Also, for reasons of safety, floor-mounted TV studio lamps or theatre stage lamps may operate at I I0 v. where overhead or other permanent fixtures operate at 240 v. such v. lamps or groups of lamps may operate from 240 v. dimmers through stcpdown transformers which draw a significant magnetizing current surge on switch-0n. Special characteristics are thus required of a protective fuse; it must fuse before the thyristors are destroyed by a load short circuit but must carry the coldstart current surge of a tungsten lamp fully loading the dimmer, and/or the transient magnetizing current surge of a transformer for such a lamp, without blowing. So-called crowbar" protective systems short-circuiting the supply between the fuse or circuit-breaker and the power-control thyristors in response to electronic detection of a fault condition at the output are uneconomic; the "crowbar thyristors must have power ratings as great as or greater than the control thyristors if they are to protect these and themselves survive.

A series inductor opposes the action of the fuse or circuit breaker; the energy stored in the inductor by the fault current sustains an arc across the gap in the fuse or breaker, lengthening the duration of the fault condition. We find that an inductance of the value necessary to give adequate interference suppression and adequate protection against false triggering hy supply transients so prolongs a short circuit fault condition by sustaining an arc in the protective fuse or circuit breaker that the control thyristors are destroyed before the fault condition is cleared.

Similar problems may arise with other electronically triggered switching devices which, for the purpose of this specification are defined as devices which can be triggered into their conducting state by a voltage applied to a control electrode and which will remain in that state until a voltage of opposite sense is applied to a suitable electrode In the specification of copending British Pat. application No. 22536/66 there is described and claimed means for substantially improving the performance of lighting control cir cuits by the inclusion ofa series-connected saturable inductor. This other specification contains a description and drawings of embodiments that are also included herein.

According to the present invention there is provided a lighting control system including, connected effectively in series between terminals for connection to a source of electric supply and terminals for connection to a load, one or more electronically triggered switching devices as hereinbefore defined, the system also comprising a feedback circuit arranged to feed to a circuit arranged to control the voltages applied to the control electrode or electrodes of the one or more switching devices a voltage dependent on the mean value in any half cycle of the voltage, or a simulation of the voltage, between said load terminals. This feedback can be arranged to produce desirable modifications in the control characteristics, as will hereinafter appear, particularly in respect of improved accuracy and reliability such as is obtainable with a closedloop control system.

Referring to FIG. 7, part of this circuit closely resembles that already described with reference to FlG. 5 and this part which has the same references as in FIG. 5 will not be further described in detail. Between the supply terminals ST and the thyristors SCRll and SCRZ is connected a fuse F in series with a saturable inductor L The degree to which the core of this inductor saturates in dependence upon the load current is controlled in design.

Feedback may be added to the system of FIG. 5 according to the present invention by the use of a transformer T, and a rectifier system B,. Control signals applied to terminals CS and filtered by a resistor R8 and capacitor C are fed to the base of a transistor VTZ while the signal from the transformer T, and rectifier system B,, the mean value of which is propor tional to the mean output voltage across the load L in any supply half cycle, is fed through a potential decoder R110, RM to the base of a second transistor VTll. A capacitor C2 smooths the feedback to provide a mean-level signal. The

transistors VT] andVT2 with a common emitter resistor R9 constitute a differential amplifier, producing across a resistor R7 a voltage proportional to the excess of the control signal over the feedback signal. The unijunetion pulse generator system functions as in FIG. 5 and FIG. 6, waveforms V, W, X and Y of FIG. 6 appearing on the leads so marked in FIG. 7. It should, however, he noted that these waveforms of FIG. 6 are to be read as being with reference to the voltage on the lead A in FIG. 7, Moreover the waveform X is now proportional to the difference between the control signal and the mean value of the output voltage in any supply halfcycle. The differential amplifier VI], VT}. contributes to the loop gain of the feedback control system, the fuse F and saturable inductor L, protect the power control thyristors SCR], SCR2 against load short circuits, and a resistor R6 limits the current available from an emitter follower transistor VT3 as the unijunetion transistor UJT discharges capacitor C.

Such feedback produces the same reduction in control nonlinearity as in control sensitivity so that if the control sensitivity reduction factor is made large by making the open-loop gain large the mean value M of the output voltage follows the control voltage accurately despite errors in the open-loop control law. Hence the normalized mean output voltage in any supply half cycle may be made accurately proportional to the control signal. However, since control may be exercised only once in each supply half cycle the usable loop gain is limited. as in a sampling system, by stability considerations.

The system of FIG. 7 adjusts itself so that the feedback voltage at VT] base is substantially equal to the control signal at the base of VT2; the closed-loop control sensitivity is thus determined primarily by the total voltage division ratio of the transformer T and the resistors R10 and R I If the saturable inductor L. of FIG. 7 is designed to give a ramp duration of about lms. when the system is switching a 50 c./s. supply at 6,=90, M and R are substantially equal for all values between 0 and 1. Hence the normalized RMS voltage output R of FIG. 7 in this case is substantially proportional to the normalized control voltage v, instead of the relation being R v for a system without the saturable inductor. If the normalized lamp light output L varies as R, then L v in this case as against L v for no inductor.

Feedback such as that provided by T,, B in FIG. 7 may be derived from the output of transformer T and rectifier system B of FIG. 5 provided that a further thyristor is used to simulate the action of the power control thyristors. Thus in the circuit of FIG. 8, a transformer T, and rectifier system B, produce the waveform V of FIG. 6(a) with respect to the lead A. This is clipped by Zener diodes ZD], ZDZ to provide the interrupted HT supplies, namely the waveform W of FIG. 6(b), for the control system. The capacitor C connected to the emitter of the unijunetion transistor UJT is charged initially by an emitter follower VTS, base drive for which is derived through an emitter follower VT4 from the voltage drop produced across a resistor R,, and a diode D] by the collector current of the transistor VT,. A cosine term is added to the voltage on the capacitor C, namely the waveform Y in FIG. 6(b), by the current flow in a resistor R and an adjustable resistor VR4 due to the signal V. When the voltage on the capacitor C exceeds the emitter breakdown voltage 1 W of the unijunetion transistor UJT, this transistor discharges the capacitor C through the primary winding P of the pulse transformer PT. Current in the base 2 of UJT is limited during discharge of C by the resistor R current in VTS and VT4 is limited by resistors R and R, respectively. The pulses produced on secondary windings S1, S2 of the pulse transformer PT by the discharge of the capacitor C trigger one or other of two relay thyristors SCR4 and SCRS through potential divider resistors R R and R R dependent on which of these thyristors is forward biased in the supply half cycle considered. The

thyristor SCR4 when triggered and forward biased connects trol thyristor SCR]: the thyristor SCRS when triggered and forward biased similarly triggers the power control thyristor SCR2 from the winding W2 through a resistor R Zener diode ZD4 and a divider R R A selenium surge suppressor SS limits, by reverse breakdown action, excessive voltage transients which might otherwise appear across the thyristors SCRI and SCR2; the saturable inductor L, limits the rate of rise of load current and protects the thyristors SCRI and SCR2 from false triggering by fast supply transients; the fuse F protects the thyristors S('R l, SCR2 against load short circuits. Diodes I), to I) catch" the overswings ol' the hack edges of the pulses on the relevant windings of the pulse transformer PT and capacitors (.4, C5 in conjunction with the leakage inductances of the transformer T protect the thyristors SCR4, SCRS against false triggering by supply transients.

When the capacitor C discharges, the resultant pulse voltage on the secondary S3 of the pulse transformer PT triggers a thyristor SCR3 through a potential divider R R The thyristor SCR3 simulates the power control thyristors SCR] and SCR2, gating the signal V from the winding W3 of the transformer T and rectificrs B to produce at a point PS 21 DC simulation of the controlled output voltage. A saturable inductor L,- may be included to limit the rate of rise of voltage at PS to equal proportionately that across the nominal lamp load L, making the simulation exact for one specific value of load resistance, but the inductor L,- tends to sustain conduction in the thyristor SCR3 at the end of each half cycle; if SCR3 fails to cease conduction at the end of each supply halfcycle the output at R loses proportionality to the controlled output of SCR] and SCR2 and control is lost. The inductor L,- is not necessary fundamentally and may be replaced by a short circuit. The simulator output at PS is smoothed by R and capacitor C4 and is applied as feedback for the purpose already described through a diode D6 and a potential divider R R to the base of the transistor VT]. DC control signals, applied at C8, smoothed by the resistor R8 and capacitor C3, are applied to the base of the transistor VT2, the emitter of which feeds the emitter of the transistor VTl through the resistor element ofa divider VR24. The voltage divider made up of the elements R4], D8, R42 defines the voltage appearing at the emitter of a transistor VT3. The voltage across a resistor R43 in the emitter circuit of the transistor VT3 hence defines the emitter and collector currents of VT3, the diode D8 compensating for thermal changes in the base-emitter potential difference of VT3. The divider VR24 acts as a universal shunt," partitioning the constant collector current of VT3 between transistors VT] and VT2 according to its setting.

In operation, the system adjusts itself in such a manner that simulator feedback voltage at the base of VT] is substantially equal to the control voltage at the base of VTZ. The voltage comparator VT], VT2 and the unijunetion transistor trigger circuit provide high open-loop gain despite the loss in R38, R39 and R40. A gain reduction factor of 10 or more is readily achieved for a control voltage range of 2.5 v. and an RMS out put voltage range of 250 v. The control voltage range is selected by choice of the division ratio of the divider R38, R39, R40. The minimum output level is set, as will be explained subsequently, at zero control voltage by means of VR4l, which controls the amplitude of the cosine term on the capacitor C, and the output level, say 200 v. RMS, corresponding to a control voltage near maximum, say 2.0 v. DC, is set by the divider VR24. A resistor R44 biases the system to zero output if the control signal input terminals CS are opencircuited. Diode D] compensates for variations in VTS baseemitter potential with temperature; further diodes may be added in series with D] as necessary to give minimum overall variation of system output voltage with temperature.

Where the inductor L is included and proportioned to make the simulator voltage rise time equal to the load voltage rise time the feedback is a true simulation of the output voltage in any half cycle and the control law is the same as for the feedback of FIG. 7, i.e. R v, I. v. Where no feedback inductor Ly is used in FIG. ii and the feedback gain reduction factor vider R24, R25 presents a low or a high output impedance respectively to the divider RE, R 30.

The circuits described with reference to'FIGS. 7, b and 0 have been developed specifically for the control of tungsten filament lamps of high power. Obviously, these circuits may be adapted to control any AC-operable lamp or other device havsimulator feedback of FIG. 8 without feedback choke L; and

L v for the full overall feedback of FIG. 7, against the L v law generally regarded by stage lighting engineers as desirable. We find that the systems of FIG. 7 or FIG. 8 may be used to provide an entirely satisfactory relationship of normalized.

light output L from tungsten sources against a normalized control voltage v over three decades of v by setting these circuits to give an appropriate residual output voltage at zero control volts. Thus in setting up the circuit of FIG. 8, VR41 is set at zero control volts to give a normalized residual output voltage R which may be in the range 0.1 to 0.25, dependent on the use to which the lighting system is to be put; VRZA is set to give R =l at v=l. Where, as in the circuit of FIG. 8, we have a response R =v and L=( v ")*-=v"- for a residual output R =O we now have a response of the form R,,=(a+( la)v) and L=(a+(1a)v) with'R a and Lu=a at l=0. and R=L=l at v=l. The responses of the system of FIG. 8 are shown in FIG. 9 for residual normalized RMS output voltages R,,=O.l2 5, 0.187 and 0.25; the resulting light output responses for L R are also shown, the corresponding residual normalized output levels being L -0.00025, 0.001 and 0.004. Similar results may be obtained with the circuit of FIG. 7 if R5 is made adjustable to provide a controlled residual output voltage at zero control volts.

We find, using tungsten filament lamps in the theatre, that if all lamps are faded together there is a fairly well defined maximum normalized lamp voltage at which the lamps may be switched on and off without producing noticeable effect; the corresponding value of R is about 0.125. Hence for theatre use the dimmers are preset to give R,,=0. l at v=0.

The characteristics of TV cameras and TV broadcast receivers are such that present studio lighting practice avoids the use of black-outs and maintains a higher minimum lighting level than the live theatre. For TV studio use the curves of FIG. 9 corresponding to R,,=0.25 at v=0 are more appropriate, and dimmers for such use are set to give this order of residual output. FIG. 9 shows that the system of FIG. 8, set to give a residual output R in the range 0.125 to 0.25 and used with tungsten filament lamps, results in an overall characteristic which lies between L v and L v in the upper decade of control voltage range and in which the power law of the relationship falls substantially in lower control voltage decades.

It is convenient in the circuit of FIG. 8 to catch" the potential on the resistor R39 at a voltage a little below that which appears by feedback action at zero control voltage. A small negative control voltage then results in a diode D7 catching the potential at R39, so removing the feedback by way of T SCR3 and R38; the system now operates at full open-loop gain for any increase in negative control voltage input, and cuts off rapidly. Thus the system may be switched from its residual output voltage R to cut off by means of a small negative control signal. The corner produced in this way in the control characteristic may be made sharp or rounded according as diing an output dependent on the angle of flow of supply current.

Simulator feedback may be used where true output feedback is inconvenient, difficult or impossible. It uses a restricted feedback loop which does not include the true output branch and its driving transducer but uses instead a model of these. Because the simulator does not model the effects of load changes the feedback does not reduce the output impedance of the system. Thus the feedback of FIG. 8 does not depend on the output voltages; it simulates the output voltage and the simulation is incomplete.

It is preferred to use the simulator feedback of FIG. 8 rather than the true output feedback of FIG. 7 because the lower output impedance of FIG. 7 could only aggravate difficulties in thyristor protection by aggravating the cold-start current surge in a tungsten lamp load.

I claim:

1. A lighting control system comprising terminals for connection to a source of electric supply, terminals for connection to a load, at least one electronically triggered switching device connected effectively in series between said source terminals and said load terminals, said device being one having a control electrode and capable of being triggered into a conducting state by a control voltage applied to said control electrode and remaining in this state until a voltage of opposite polarity is applied to the device, a control circuit for producing said control voltage, means for deriving a feedback voltage dependent on the mean value of any half cycle of a simulation of the voltage between said load terminals, said means including a switching device connected to be triggered into conduction with said electronically-triggered switching device, and means for applying said feedback voltage to said control circuit.

2. A control system according to claim 1 comprising means for interrupting the action of said feedback circuit at a point in the control range near the minimum output end thereof.

3. A lighting control system including first and second electric supply terminals, first and second lighting load terminals,

a conductive connection between the second electric supply 1 terminal and the second lighting load terminal, an output branch connecting the first electric supply terminal and the first lighting load terminal and including one or more first electronically operated current control devices, and a simulator branch energized effectively from the electric supply terminals and including one or more further electronically operated current control devices of like kind to the first such devices in series with a predetermined simulator load, the lighting control system further including a control circuit providingcorresponding control signals to the first and further electronically operated current control devices and a feedback circuit arraNged to feed to the control circuit a signal dependent on the mean value in any supply half cycle of the voltage across the predetermined simulator load.

41. A control system according to claim 3 in which the electronically operated current control devices are electronically triggered switching devices as hereinbefore defined.

Images