US2788496A - Active transducer - Google Patents

Active transducer Download PDF

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US2788496A
US2788496A US360301A US36030153A US2788496A US 2788496 A US2788496 A US 2788496A US 360301 A US360301 A US 360301A US 36030153 A US36030153 A US 36030153A US 2788496 A US2788496 A US 2788496A
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impedance
transducer
networks
poles
accordance
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US360301A
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John G Linvill
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AT&T Corp
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Bell Telephone Laboratories Inc
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Priority to NLAANVRAGE7905855,A priority Critical patent/NL187239B/xx
Priority to BE529361D priority patent/BE529361A/xx
Priority to NL97559D priority patent/NL97559C/xx
Application filed by Bell Telephone Laboratories Inc filed Critical Bell Telephone Laboratories Inc
Priority to US360301A priority patent/US2788496A/en
Priority to DEW13683A priority patent/DE1127401B/de
Priority to FR1102666D priority patent/FR1102666A/fr
Priority to GB16596/54A priority patent/GB753085A/en
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Publication of US2788496A publication Critical patent/US2788496A/en
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    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H11/00—Networks using active elements
    • H03H11/02—Multiple-port networks
    • H03H11/04—Frequency selective two-port networks
    • H03H11/10—Frequency selective two-port networks using negative impedance converters
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B3/00—Line transmission systems
    • H04B3/02—Details
    • H04B3/04—Control of transmission; Equalising
    • H04B3/16—Control of transmission; Equalising characterised by the negative-impedance network used

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  • FIG. 1 A first figure.
  • CONVERTER CONVERTER L J 1 0 Y J a 6 is 27 26 lNl ENTOR J G. LIN V/LL ATTORNEY nited States Patent "ice 2,788,496 it ACTIVE TRANSDUCER John G. Linvill, Whippany, N; 1., assignor to Bell Telephone Laboratories, Incorporated, New York, N. Y., a corporation of New York Application June 8, 1953, Serial No. 360,301
  • an R-C network usually called anR-C network, .is attractive.
  • an R-C filter introducesex cessive loss in the transmission band and requires many more elements than does a filter which includes both inductors and capacitors. These defects may be overcome by including an active element.
  • One active transducer of this type is disclosed in United States Patent 2,549,065, issued April 17, 1951, to R. L. Dietzold. .
  • the active element is a stabilized feedback amplifier.
  • passive elements are resistors and only one type of reactor, either capacitive or inductive.
  • the present invention is directed to another type of active transducer with unrestricted transmission characteristic.
  • the circuit comprises two passive networks and a negative impedance converter connected in tandem between them.
  • the negative impedance converter hereafter called simply a converter, has an impedance conversion ratio, designated M, which is negative. Thus, it presents at one pair of its terminals an impedance which is M times the impedance connected to its other terminal pair.
  • Each of the passivenetw orks is made up of one or more resistors and one or more reactors.
  • the reactors may include both inductors and capacitors, or they may be all of the same type. Usually, it is preferred to use only capacitors for the reactors.
  • Resistors and capacitors are, in general, cheaper, smaller, and more nearly ideal elements than are inductors.
  • the passive networks may be simple structures, either balanced or unbalanced. They may, for example, be lattice, ladder, bridged-T, or twin-T networks. 7 reactors required in the networks is no greater than that required in apassive transducer having a comparable transmission characteristic. 3
  • the passive networks At their ends facing the converter, the passive networks have driving-point impedances oneof which is equal to -M times the other at oneor more preselected active transducer in accordance with the invention.
  • Fig. l is a block diagram of an active transducer in accordance with the invention.
  • Fig. 2 represents the complex frequency plane it on which a re plotted the poles of the transfer impedance of a typical low-pass filter in accordance with the invention
  • Fig. 3 shows, on the complex frequency plane, the distribution of the zeroes and the poles of the difierence between the driving-point impedances of the passive networks at their ends facing the converter, for the low-pass filter example; e n
  • Fig 4 shows the circuitof the low-pass filter when the passive networks are unbalanced ladder structures made up of resistors and shunt capacitors; r i
  • Fig. 5 is atypical relativeresponseversus frequency characteristic obtainable with the low-pass filter of'Fig. 4;
  • Fig. 6 shows the circuit of a high-pass filter in accordance with the invention, in which the passive networks are constituted by resistors and shunt inductors;
  • Figs. 7, 8, and 9 show, respectively, the pole-zero dis tribution, the network configuration, and a typical characteristic of a second high-pass filter in accordance with the invention
  • Fig. 10 shows two active band-pass filters in accordance with the invention connected in tandem by an amplifier
  • Fig. 11 shows the relative response characteristic of the filter of Fig. 10
  • Fig. .12 shows a generalized lattice structure which may be used for the passive networks in Fig. 1;
  • Figs. 13 and 14 show, respectively, the network configuration and the characteristic of a low-pass filter in accordance with the invention in which a twin-T network provides a peak of attenuation at a finite frequency.
  • Fig. 1 shows in block diagram an embodiment of an The transducer comprises two four-terminal passive networks 1 and 2 and an interposed negative impedance converter 3 connected in tandem between a pair of input terminals 5, 6. and a pair of output terminals 7, 8.
  • a suitable source of signals may be connected to the The total number of input terminals and a suitable load, not shown, may be connected to the output terminals.
  • the converter 3 is an active four-pole network which presents at its input terminals 9, 10 an i impedance which, over the frequency range of interest, is equal to M times the impedance connected to its output terminals 11, 12.
  • the converter 3 may be of thevacuum-tube type, examples of which are disclosed in the paper by J. L. Merrill, Jr., entitled, Theory of the negative impedance converter, in the Bell System Technical Journal, vol. XXX, No. 1, January 1951, pages 88 to 10 9.
  • the converter 3 is of the transistor type, suitable circuits of which are disclosed in the copending United States patent application of R. L. Wallace, Jr.,
  • a converter using one or more transistors is preferred because it may be designed to have more nearly ideal characteristics. Therefore, a transducer employing a converter of this type may be designed to meet a prescribed transmission characteristic within closer limits, and the characteristic will be more stable with time.
  • the converter 3 has a current transfer ratio Mi, which is the ratio of the input current'lato the output In, and a voltage transfer ratio Me, given by the ratio of the input voltage Ea to the output voltage Eb.
  • a current transfer ratio Mi which is the ratio of the input current'lato the output In
  • a voltage transfer ratio Me given by the ratio of the input voltage Ea to the output voltage Eb.
  • One of these ratios is always negative.
  • the ratio M1 is substantially unity if junction transistors are used in the converter. Its value may be approximately doubled by using point contact transistors, but at a sacrifice in the stability of the converter. This ratio may, of course, be extended by associating a transformer with the converter. As explained in the above-mentioned patent application, the magnitude of Me may be selected within wide limits.
  • the impedance conversion ratio M of the converter which is the ratio of Me to Mt, may be given any negative value within a Wide range by choosing appropriate values of Me and Mi. A judicious choice of M may help in obtaining convenient values for the component impedance elements in the passive networks 1 and 2.
  • the transfer impedance Z'rof any four-terminal network may be expressed as the ratio of two polynomials in P giving The transfer impedance becomes infinite at the complex frequencies at which the denominator D(p) is zero. Therefore, these complex frequencies are the natural frequencies of the network.
  • passive R-C network-s, or R-L networks comprising only resistors and inductors
  • these zeroes are restricted to the negative real axis of the complex frequency plane. This constraint seriously limits the quality of approximation to an ideal filter characteristic obtainable if N(p) and D(p) are polynomials of a limited degree.
  • Active R-C or R-L networks can have natural frequencies anywhere in the left-half plane, the same as passive networks comprising resistors, inductors, and capacitors.
  • Equation 5 Z122. is the transfer impedance of the network 1, Z121; is the transfer impedance of the network 2, 2225 is the driving-point impedance of the network 1 at the terminals 9, 10, and Zllb is the driving-point impedance of the network 2- at the terminals 11, 12.
  • the derivation of Equation 5 involves, as an intermediate step, the evaluation'of the input current In. to the converter 3.
  • the driving-point impedance Z11 of the transducer at the input end is E Z 0 2 Z Z 11 I1 no 22a llb where E1 is the input voltage and Zlla. is the driving-point impedanceof the network 1 at the terminals 5, 6.
  • the driving-point impedance Z22 at the output end of the transducer is where Z221; is the driving-point impedance of the network 2 at the terminals 7, 8.
  • the zeroes of the numerator N(p) are associated with the structure of the network, not with its natural frequencies.
  • ladder networks have zeroes of transfer impedance at frequencies where shunt elements become short circuits or series elements become open circuits.
  • an R-C ladder-type network. made up of resistors and. three. shunt; capacitors will have three zeroes of transfer impedance at infinite frequency, where the capacitors are short circuits, irre spective of the values of the capacitors or the natural frequencies of the network.
  • Lattice, bridged-T, and twin-T networks will have zeroes of transmission at frequencies where a bridge-like balance occurs, regardless of the location of the natural frequencies of the complete network.
  • a suggested procedure for designing an active transducer in accordance with the invention comprises three steps. First, the designer prescribes the desired transfer impedance, within a constant multiplier, in the form of Equation 2. Next, he selects the zeroes of the denominator D(p) as the natural frequencies of an active network of the type shown in Fig. 1. Then, by a method to be described below, he obtains for the networks 1 and 2 a pair of driving-point impedances Z2221, and Zllb which are consistent with these natural frequencies. Finally, he synthesizes networks which have the driving-point impedances Z223. and Zllb. The networks chosen must also be of a form which will provide the desired zeroes of transmission at the zeroes of the numerator N(p).
  • the first example is a low-pass filter which has a Butterworth characteristic, a cut-off frequency-1 fc of 1000 cycles per second, and an attenuation rising at the rate of 18 decibels per octave of frequency.
  • the transfer impedance of such afilter. may be written as N (P) K 1m) r (if 21 21r +2 21 21rf where K is a numerical constant which determines the impedance level of the filter.
  • the transfer impedance will have three poles, corresponding to the zeroes of D(p), and three zeroes. The. zeroes all fall at infinite frequency. The poles are known to fall on a semicircle in the left half of the complex frequency plane.
  • Fig. 2
  • the natural frequencies of the filter must occur at these complex frequencies.
  • the circuit shown in Fig. 1 will have natural frequencies when the driving-point impedance Z2211 of the network 1 and the driving-point impedance Zllb of the network 2 are equal.
  • the sum Z of the impedance -Z1lb seen looking into the converter 3 at its input terminals 9, 10 and the impedance Z222. is zero, that is, when The zeroes of Z will, therefore, be at the complex fretmf where D(p) isithe denominator in Equation 14.
  • the driving point impedances and othercharacteristics of the filter are influenced by the selection.
  • Equation 19 the expansion of Equation 19 in partial fractions. It is found that the residues are always real but may be positive or negative. The terms are divided into a first group with positive residues and a second group with negative residues. It is known that any function with simple poles on the negative real axis and positive real residues in those poles is the driving-point impedance of an R-C network. Therefore, the first group of terms is associated with the impedance Zen. of the network 1. The second group of terms is associated with the impedance Zllb of the network 2.
  • the network 2 can only provide positive residues in the poles of Ziib, they will appear as negative residues when viewed from the input terminals 9, 10 of the converter 3. It is now assumed that the networks 1 and 2 will be ladder-type structures comprising resistors and shunt capacitors. The required values of the component elements are found by making a Cauer synthesis. I For the distributions of the critical frequencies Pa, Pa, Pb, 0,, (r and 0' shown in Fig. 3, the networks 1 and 2 will have the configurations shown in Fig. 4.
  • the network 1 comprises three series resistors R1, R2, and R3 and. two shunt capacitors C and C2.
  • the network 2 consists of the parallel combination of a resistor R4 and a capacitor C3.
  • the network 1 will include two capacitors and the network 2 only one capacitor.
  • the required values of the resistors R1, R2, R3, and R4, in ohms, are .600, 2040, 1200, and 1300, respectively, and the values of the capacitors C1, C2, and C3, in microfarads, are 0.384, 0.172, and 0.0820, respectively. It is assumed that the impedance conversion ratio M of the converter 3 is l.
  • Fig. 5 shows a typical relative response characteristic obtainable with the low-pass filter of Fig. 4. It is assumed that the signal source of voltage E1 connected to the terminals 5, 6 has zero internal impedance.
  • Fig. 6 shows the circuit of a second wavefilter'in accordance with the invention.
  • This is a high-pass filter having a Butterworth characteristic and an attenuation which rises at the rate of 18 decibels per octave.
  • the passive networks 1 and 2 are ladder-type RL structures, comprising resistors and shunt inductors. Their configurations are the same as those shown for the low-pass filter of Fig. 4 except that the three shunt capacitors C1, C2, and C3 are replaced, respectively, by the three shunt inductors L1, L2, and L3. Otherwise, the circuit of Fig. 6 is similar to that of Fig. 4.
  • the component resistors and inductors required in the filter of Fig. 6 may be evaluated by the same procedure described above in connection with the filter of Fig. 4.
  • Figs. 7, S, and 9 relate to another high-pass wave filter in accordance with the invention.
  • the filter has a Butterworth characteristic with an attenuation rising at the rate of 24 decibels .p'er octave.
  • the points 14, 15, 16, and 17 marked by xs are poles of the transfer impedance Z21 and zeroes of the impedance Z. These points all fall on a semicircle 23 in the left half of the plane and have a uniform spacing S from the origin. They also have a uniform angular spacing U of 45 degrees on the semicircle 23, and the points 14 and 17 have equal angular spacings of U/2 from the for axis.
  • the points 14 and 17 are conjugate, as are also the points and 16.
  • the points 19, 20, 21, and 22 marked by +s are poles of Z. These poles all fall on the negative real axis and have spacings of S1, S2, and S3, as shown.
  • the displacement of the nearest pole 22 from the origin is St.
  • the impedance Z21 has a fourth-order zero at the origin, as shown by the circle 24-.
  • Fig. 8 is a schematic circuit of a filter having the distribution of poles and zeroes shown in Fig. 7.
  • Each of the passive networks 1 and 2 is an RC ladder structure comprising two series capacitors and three shunt resistors. The procedure described in connection with Fig. 4 may be used to find the required values of these elements.
  • Fig. 9 shows a relative response characteristic obtainable with the filter of Fig. 8 when the source E1 and the load Rn each have a high impedance compared to the resistance of the end resistor connected in parallel therewith.
  • two or more active transducers may be connected in tandem and isolated from eachother by one or more amplifiers.
  • Fig. 10 shows two band-pass wave filters 25 and 26 connected in tandem between input terminals 5, 6 and output terminals 7, 8 and an interposed amplifier 27, preferably of the transistor type.
  • the filter 25 comprises two passive networks 29 and 30 connected through a converter 31.
  • the corresponding units are designated 33, 34, and 35.
  • Each of the networks 29 and 33 is constituted by the series combination of a resistor and a capacitor in a series branch.
  • Each of the networks 30 and 34 is made up of the parallel combination of a resistor and a capacitor in a shunt branch.
  • Fig. 11 shows a typical relative response characteristic obtainable with the filter of Fig. 10.
  • the midband frequency is located at 1000 cycles per second and the band width is approximately 200 cycles. Zeroes of tnan-smission occur at zero and infinite frequencies. By properly designing and adjusting the amplifier 27, considerable gain in the transmission band may be achieved, if desired.
  • Fig. 12 shows the generalized circuit of a lattice structure which may be used for either or both of the passive networks 1 and 2 in Fig. 1.
  • the lattice comprises two equal series impedances Za, Za and two equal diagonal impedances Zb, Zb connected between a pair of input terminals 37, 33 and a pair of output terminals 39, 40.
  • the synthesis of an active filter of the type shown in Fig. 1 using two such lattice networks to provide intermediate attenuation peaks proceeds as explained above in connection with Fig. 4 through the specification of the desired transfer impedanceZzi, the selection of the impedance Z, and the evaluation of the driving-point im pedances Z229. and Z1111. At this point, one has the driving-point impedances of the networks 1 and 2 and knows the frequencies at which these networks should introduce peak-s of attenuation.
  • the driving-point impedance Zn at either end of the lattice of Fig. 12 is D 2 B p) and the transfer impedance Zr in either direction is Z I) Z a T p Z 23
  • the driving-point impedance of the network 1 will be Z223.
  • the driving-point impedance of the network 2 will be 2111;.
  • the i-mpedances Za and Zb are to be R-C structures, one selects the largest permissible values for the constant multipliers. Finally, he determines the configurations of these branches and the required values of the component resistors and capacitors to provide the impedances Za and Zb.
  • the synthesis of an RC lattice network from a specified drivingpoint impedance and a transfer impedance specified within a constant multiplier is described in greater detail, for example, in the paper by J. L. Bower and P. F. Ordung entitled, The synthesis of resistor-capacitor networks, in the Proceedings of the I. R. B, vol. 38, No. 3, March 1950, pages 263 to 269.
  • the passive network 1 comprises a resistor R5 in series with a parallel twin-T structure.
  • One of the Ts is constituted by the two equal series resistors R6, R6 and an interposed shunt capacitor C4.
  • the other is made up of the equal series capacitors C5, C5 and the interposed shunt resistor R7.
  • the network 2 is an R-C ladder structure with a series resistor, a shunt resistor, and two shunt capacitors which will provide a 12-decibel per octave attenuation rate at the higher frequencies.
  • Fig. 14 gives the relative response characteristic of the filter. The band cuts off at 1000 cycles per secohd. The attenuation peak occurs at 2000 cycles.
  • the first step in designing the circuit of Fig. 13 is to select the poles and zeroes of the transfer impedance Z21 of the filter to provide an acceptable characteristic, that is, one having a reasonably fiat pass band from zero to 1000cycles, an attenuation peak at 2000 cycles, and a high-frequency attenuation rate of 12 decibels per octave.
  • the method described in my paper entitledyThe approximation with rational functions of prescribed magnitude and phase characteristics, in the Proceedings of the I. R. E., vol. 40, No. 6, June 1952, pages 711 to 721, may be followed in determining the locations of the poles and zeroes.
  • the initial resistor R5 is selected in accordance with the impedance of the voltage source E1.
  • the twin-T structure is designed to have reasonable values for the component elements Re, R6, R7, C4, C5, and C5 and to provide the attenuation peak at 2000 cycles.
  • a method of designing such a twin-T network is presented, for example, in United States Patent 2,106,785, to H. W. Augustadt, issued February 1,' 1938.
  • the driving-point impedance Z22. of the net- Work 1 at its end facing the converter 3 is expressed in the form of a partial fraction expansion.
  • the driving-point impedance Z1111 of the network 2 at its end facing the converter 3 is determined. This can be done through a consideration of the expression for Z, which, as given by Equation 18, is the difference between Z22a and Z111).
  • Z which, as given by Equation 18, is the difference between Z22a and Z111).
  • Its numerator has a constant multiplier K1 and includes the factors of D(p).
  • Its denominator includes as factors the denominator of Z229. and the denominator of Zub, as yet undetermined but known to be of the form (p-l-a) (p+b), Where a and b are as yet unknown.
  • the unknown constants K1, a, and b are selected consistent with the requirements that the residues in the poles of Z2211 must be positive and of the values already known and that the residues in the remaining poles, associated with Z11b, must be negative.
  • the constants K1, a, and b determined one may write an explicit expression for Z.
  • the impedance Zllb is found by substracting Z from Z22a- With Ziib thus determined, the final step is to synthesize an R-C ladder structure with shunt capacitors to obtain the network 2 shown in Fig. 13.
  • the input impedance Zr of the converter 3 depends upon the impedance conversion ratio M.
  • the ratio M changes with time, temperature, or other environmental conditions, causing a corresponding change in Z1.
  • - Equation 5 shows that a change in 'ZI from the value Z11b will not affect the zeroes of the transfer impedance Z21 of the filter but will change the poles. A change in the locations of these poles will-cause a change in the transmission characteristic of the filter.
  • the distance Q from the origin of the most remote pole 19 is greater than 8/2 but not more than 38, where S is the radius of the semicircle 23 on which all of the zeroes 14, 15, 16, and 17 of Z fall. Also, the pole spacings S1, S2, and S3 are: approximately equal.
  • An active transducer comprising two passive networks and a negative impedance converter connected in tandem therebetween, said networks having at their ends “facing said converter driving-point impedances which, at
  • a prescribed natural frequency of said transducer are related to each other by a numerical factor equal in magnitude to the impedance conversion ratio of said converter.
  • a transducer in accordance with claim 1 having a transfer impedance with unrestricted zeroes and poles, said networks comprisingresistors and only a single type of reactor.
  • a transducer in accordance with claim 1 said converter is of the vacuum-tube type.
  • a transducer in accordance with claim 1 having the transmission characteristic of a low-pass filter.
  • a transducer in accordance with claim 1 having the transmission characteristic of a band-pass filter.
  • a transducer in accordance with claim 1 having the transmission characteristic of a high-pass filter.
  • a transducer in accordance with claim 13 having the transmission characteristic of a wave filter with a peak of attenuation at a finite frequency other than zero.
  • twin-T structure comprises two T-networks connected in parallel, one of said T-networks including two series resistors and an interposed shunt capacitor and in which the other of said T-networks including two series capacitors and an interposed shunt resistor.
  • each of said passive networks is a ladder-type structure.
  • a transducer in accordance with claim 26 in which said zeroes fall approximately on a semicircle in the left half of said plane.
  • a transducer comprising two passive networks and an interposed negative impedance converter connected in tandem, said converter having an impedance conversion ratio approximately equal to -1, said networks comprising resistors and only a single type of reactor, and said networks having at their ends facing said converter driving-point impedances which are substantially equal at a preselected natural frequency of the transducer.
  • each of said networks is of the ladder type.
  • a transducer in accordance with claim 32 in which the difference between said driving-point impedances is an impedance having a plurality of zeroes and a plurality of poles, said poles being W in number, all of said poles being located on the negative real axis of the complex frequency plane, one of said poles being spaced from the origin by a distance Q which is between half and triple the average displacement of said zeroes from said origin, a second of said poles being located not more than Q/W from said origin, and the remaining poles being spaced approximately uniformly between said first and said second poles.

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US360301A 1953-06-08 1953-06-08 Active transducer Expired - Lifetime US2788496A (en)

Priority Applications (7)

Application Number Priority Date Filing Date Title
NLAANVRAGE7905855,A NL187239B (nl) 1953-06-08 N-gesubstitueerd 2-cyaanaziridinederivaat, werkwijze voor het bereiden daarvan en farmaceutisch preparaat dat een dergelijk preparaat bevat.
BE529361D BE529361A (ja) 1953-06-08
NL97559D NL97559C (ja) 1953-06-08
US360301A US2788496A (en) 1953-06-08 1953-06-08 Active transducer
DEW13683A DE1127401B (de) 1953-06-08 1954-04-08 Aktives UEbertragungssystem
FR1102666D FR1102666A (fr) 1953-06-08 1954-04-28 Transducteurs actifs
GB16596/54A GB753085A (en) 1953-06-08 1954-06-04 Improvements in or relating to electrical wave transmission networks

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DE (1) DE1127401B (ja)
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US2933703A (en) * 1958-05-09 1960-04-19 Bell Telephone Labor Inc Active impedance branch
US2936426A (en) * 1955-05-02 1960-05-10 Joseph F Mcclean Filter network
US3068329A (en) * 1959-04-28 1962-12-11 Bell Telephone Labor Inc Negative-impedance repeater
US3178650A (en) * 1960-12-05 1965-04-13 Hamasaki Joji Four-terminal, negative-resistance amplifying circuit
US3187266A (en) * 1960-09-12 1965-06-01 Rca Corp Impedance inverter coupled negative resistance amplifiers
US3202925A (en) * 1960-03-25 1965-08-24 Nippon Electric Co Filter amplifier
US3243740A (en) * 1960-10-20 1966-03-29 Westinghouse Electric Corp Reactance enhancing networks
US3243743A (en) * 1960-10-20 1966-03-29 Westinghouse Electric Corp Negative reactance tuned circuit
US3243739A (en) * 1960-10-20 1966-03-29 Westinghouse Electric Corp Negative reactive circuitry
US3255421A (en) * 1961-10-31 1966-06-07 United Aircraft Corp Negative resistance distributed amplifier
US3286206A (en) * 1963-04-24 1966-11-15 Kabushikikaisha Taiko Denki Se Active cr two-terminal circuit
US3289116A (en) * 1962-03-21 1966-11-29 Bell Telephone Labor Inc Prescriptive transformerless networks
US3408590A (en) * 1966-10-31 1968-10-29 Bell Telephone Labor Inc Active hybrid filter using frequency emphasizing and attenuating networks
US3594650A (en) * 1968-05-10 1971-07-20 Ericsson Telefon Ab L M Band selection filter with two active elements
US3731218A (en) * 1971-09-09 1973-05-01 United Aircraft Corp Active double tuned band pass filter
US3906350A (en) * 1973-03-12 1975-09-16 Comark Ind Inc Nyquist slope filter useful for monitoring video modulation at transmitting station
US4147997A (en) * 1976-06-23 1979-04-03 The Post Office Active filters utilizing networks of resistors and negative impedance converters
US4151493A (en) * 1977-10-03 1979-04-24 Northern Telecom Limited Negative impedance converters

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US4533951A (en) * 1982-09-27 1985-08-06 Rca Corporation System for generating and displaying a compatible high definition television signal by progressive scanning

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US2243440A (en) * 1936-09-30 1941-05-27 Rca Corp Wave transmission circuits
US2549065A (en) * 1948-11-02 1951-04-17 Bell Telephone Labor Inc Frequency discriminative electric transducer

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US2093665A (en) * 1933-01-30 1937-09-21 Rca Corp Star and delta connection of impedances
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US2197348A (en) * 1938-05-17 1940-04-16 Rca Corp Impedance inverter
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Cited By (18)

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US2936426A (en) * 1955-05-02 1960-05-10 Joseph F Mcclean Filter network
US2933703A (en) * 1958-05-09 1960-04-19 Bell Telephone Labor Inc Active impedance branch
US3068329A (en) * 1959-04-28 1962-12-11 Bell Telephone Labor Inc Negative-impedance repeater
US3202925A (en) * 1960-03-25 1965-08-24 Nippon Electric Co Filter amplifier
US3187266A (en) * 1960-09-12 1965-06-01 Rca Corp Impedance inverter coupled negative resistance amplifiers
US3243739A (en) * 1960-10-20 1966-03-29 Westinghouse Electric Corp Negative reactive circuitry
US3243740A (en) * 1960-10-20 1966-03-29 Westinghouse Electric Corp Reactance enhancing networks
US3243743A (en) * 1960-10-20 1966-03-29 Westinghouse Electric Corp Negative reactance tuned circuit
US3178650A (en) * 1960-12-05 1965-04-13 Hamasaki Joji Four-terminal, negative-resistance amplifying circuit
US3255421A (en) * 1961-10-31 1966-06-07 United Aircraft Corp Negative resistance distributed amplifier
US3289116A (en) * 1962-03-21 1966-11-29 Bell Telephone Labor Inc Prescriptive transformerless networks
US3286206A (en) * 1963-04-24 1966-11-15 Kabushikikaisha Taiko Denki Se Active cr two-terminal circuit
US3408590A (en) * 1966-10-31 1968-10-29 Bell Telephone Labor Inc Active hybrid filter using frequency emphasizing and attenuating networks
US3594650A (en) * 1968-05-10 1971-07-20 Ericsson Telefon Ab L M Band selection filter with two active elements
US3731218A (en) * 1971-09-09 1973-05-01 United Aircraft Corp Active double tuned band pass filter
US3906350A (en) * 1973-03-12 1975-09-16 Comark Ind Inc Nyquist slope filter useful for monitoring video modulation at transmitting station
US4147997A (en) * 1976-06-23 1979-04-03 The Post Office Active filters utilizing networks of resistors and negative impedance converters
US4151493A (en) * 1977-10-03 1979-04-24 Northern Telecom Limited Negative impedance converters

Also Published As

Publication number Publication date
NL97559C (ja)
GB753085A (en) 1956-07-18
FR1102666A (fr) 1955-10-25
BE529361A (ja)
NL187239B (nl)
DE1127401B (de) 1962-04-12

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