Classifications

• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04B—TRANSMISSION
  • H04B13/00—Transmission systems characterised by the medium used for transmission, not provided for in groups H04B3/00 - H04B11/00
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
  • H04L27/00—Modulated-carrier systems
  • H04L27/001—Modulated-carrier systems using chaotic signals

Definitions

• the present invention relates to a method and a device for wireless transmission of data.
• the method is suitable for transmitting digital data.
• the present invention is applicable in many fields of information transmission, for example, in telecommunications, measuring technology, sensors, and medical technology.
• deterministic methods based on progressive electromagnetic carrier waves is typical for wireless signal and data transmission. This means that a transmitter emits the modulated signal directly, or typically using a carrier wave.
• the useful information is modulated on the carrier wave using different modulation methods, such as amplitude, frequency, or phase modulation.
• the present invention is based on the object of specifying a method for wireless data transmission which combines the lowest possible power consumption at the transmitter and receiver with the highest possible range of the information transmission.
• This object is achieved by a method specified in Claim 1 and a device specified in Claim 6 for global scaling communication, or GSCOM in short, in which data transmission over large distances is implemented using global scaling (GS) modulation and demodulation of coupled random processes.
• GS global scaling
• GS is an introduced physical concept which illustrates that frequency distributions of physical variables such as masses, temperatures, weights, and frequencies of real systems are logarithmically scale-invariant, see H. Müller, Global Scaling, Special 1, Ehlers Verlag 2001.
• variable z represents the partial numerator in this case, whose value is fixed at the value 2 according to GS for subsequent frequency analyses.
• numeric values are a function of the mass units used as the basis.
• the variables to be analyzed are set in relation to physical constants y, the standard measures. These constants are only known within a predefined precision, however, because of which there are upper and lower limiting values for these constants.
• the integral partial denominators [n 0 ,n 1 ,n 2 . . . ] must always have an absolute value greater than the numerator because of the convergence condition for continued fractions, see O. Perron, Die Lehre von den Kettenbrüchen [the teaching of continued fractions], Teubner Verlag, Leipzig, 1950, and are always whole numbers divisible by 3.
• a predefined physical variable such as a frequency
• a GS continued fraction method may be decomposed according to the GS continued fraction method and converted into a continued fraction code. This is to be described for exemplary purposes by a GS continued fraction decomposition for a frequency f 0 .
• Equation (3) a continued fraction decomposition and the calculation of the partial denominators n 0 , n 1 , n 2 , n 3 , n 4 , etc., results.
• the frequency 2032 Hz corresponds to the GS continued fraction code [ ⁇ 48; 9086].
• GS node point frequencies according to equation (3) are, for example, 5 Hz, 101 Hz, 40804 Hz, 16461 kHz. The present invention will be described further base on these foundations of GS frequency analysis.
• a modulator and/or demodulator and a modulation method and/or demodulation method which allows cost-effective modulation and/or demodulation.
• the modulator is identified as a GS modulator and the demodulator is identified as a GS demodulator, since the modulation and demodulation are implemented based on GS.
• An additional object of the present invention is increasing the range and the security of information transmission.
• the transmission is performed through the manipulation of deterministic processes
• the transmission is performed through GS modulation and GS demodulation of coupled random processes.
• a device for wireless information transmission e.g., of data or signals, comprises a transmission unit having a modulator for global scaling modulation of information and having a coupler for coupling the information into a random process, a receiver unit having a demodulator for global scaling demodulation of information and a decoupler for decoupling the information from the random process.
• the device and the method use coupled random processes, in particular coupled noise processes, as information carriers.
• FIG. 10 shows the mean fluctuation of unfiltered binary numbers over the natural-logarithmic time axis for a period of time of 12 hours.
• the data was obtained using hardware according to the variation described in chapter 1.2 and using software according to equation (5).
• FIG. 1 shows tool GSC3000 for GS analysis of frequencies
• FIG. 2 device and method schematics of data transmission
• FIG. 3 shows detailed schematics of the method and the device
• FIG. 4 shows background noise of a semiconductor component
• FIG. 5 shows harmonic components of the background noise
• FIG. 6 shows a circuit diagram of an external noise generator for technical generation of white noise
• FIG. 7 shows variant a having an external noise module
• FIG. 8 shows an external noise module for variant a
• FIG. 9 shows variant c having an external modem
• FIG. 10 shows fluctuation of binary numbers over the natural-logarithmic time axis
• FIG. 11 shows an illustration of a noise spectrum of a bipolar transistor (BE section)
• a high degree of correspondence of the fine structure is recognized when the histograms of the random processes used as the basis are very similar even in their smaller instances, i.e., not only their statistical characteristics such as mean values, variances, etc., correspond, but rather also the frequencies of specific measured values in the particular histograms very frequently correspond.
• this correspondence is analyzed according to GS only in non-smoothed histograms.
• the identity and/or similarity of the fine structure of histograms is now defined as the degree of the actual synchronicity of random processes.
• random processes having a high degree of correspondence in the fine structure of the histograms are referred to as coupled random processes.
• Transmitter and receiver are implemented in this method through technical terminals which firstly contain a technical noise source or permit connection of a technical noise source and secondly may perform the following processing steps 1-8 in real time.
• the device contains a list according to FIGS. 2 and 3.
• a commercially-available computer such as a laptop having an integrated sound card, is used in each case for the transmitter device ( 3 , 4 , 6 , 7 ) and the receiver device ( 8 through 11 ).
• the generation ( 3 , 4 ), modulation ( 6 ), coupling ( 7 ), decoupling ( 8 ), and demodulation ( 9 ) of coupled random processes in a transmission link for coupled random processes ( 5 ) is shown based on the noise processes of the sound cards of two commercially available computers (transmitting unit 1 and receiving unit 2 ).
• the method is applicable for any technically generated random process which may be manipulated, e.g., based on external or internal noise generators, semiconductor components, processors, modems, etc.
• the terminals are commercially available computers, laptops, or even mobile telephones. However, the method is also applicable for other terminals, other sampling frequencies f 0 , other random processes, or other changes, also of other continued fraction code components, in the following example only n 2 .
• a transmitter and a receiver are tuned to a joint frequency band (e.g., from 5 Hz to 16.4 MHz) of a technical noise process.
• a joint frequency band e.g., from 5 Hz to 16.4 MHz
• the sound card of a commercially available computer or laptop may be used, for example.
• the frequency band of the noise is thus, for example, between 100 Hz and 15 kHz.
• Further technical noise sources would be semiconductor elements or computer processes, for example.
• the time curve of a typical noise signal of a technical noise source is shown in FIG. 4.
• the noise signals of the sound card are accessed using software, for example, using Windows commands, and the particular noise level is provided to downstream analysis software.
• the standing background waves influence all local wave, oscillation, and random processes, however, this is particularly visible and measurable if the local oscillation process oscillates in proximity to a natural oscillation of higher priority (this will be explained in the following section).
• the local process then comes into resonance with the background field, which may be empirically proven in that it no longer behaves statistically correctly, but rather prefers certain value instances and avoids others.
• Local oscillation processes which may be influenced very well by the background waves are all random processes, such as radioactive decomposition processes, noise processes, or weather processes.
• Suitable sources are technical noise processes, thermal noise, or shot noise.
• Thermal noise occurs in every electronic component subject to resistance and is caused by random velocity variations of the freely movable electrons and electron holes. As a function of the type of the component and the temperature, this noise is only a few ⁇ V and requires strong electronic amplification.
• Significantly stronger noise signals are provided by pn transitions of semiconductor components, either of Z diodes or of incorrectly polarized base emitter sections of bipolar silicon transistors. The noise is generated here at a pn boundary layer which is operated above the breakthrough voltage. The charge carriers break through the barrier layer because of the applied voltage and generate the shot noise (Verges, C. 1987, Handbook of Electrical Noise, TAB Books, Blue Ridge Summit, Pa.).
• the height of the achievable noise level is strongly dependent in this case on the height of the breakthrough voltage and the size of the flowing current.
• noise levels of >1 Vpp and linear noise spectra up into the MHz range may be achieved.
• breakthrough voltages of 7-12 V and currents of 10-200 ⁇ A generate noise levels of a few hundred mVpp on BE sections of selected bipolar transistors (see FIG. 11), so that additional amplification is often superfluous. Since the noise level grows proportionally to the square root of the current flowing at constant load, it may additionally be regulated within wide limits.
• Variant a requires an additional external module for generating technical noise in addition to the laptop.
• Variant b uses the noise generator implemented in the Pentium III processor and requires no additional hardware.
• Variant c implements all functions for coupling to the background wave in an external modem.
• Variant a coupling to the background wave using external noise generator
• FIG. 6 shows a circuit diagram of an external noise generator for generating the white noise
• FIG. 7 shows the construction, comprising the external module having the analog noise generator and the laptop having integrated sound cards 21 for analog/digital conversion 22 of the noise signal provided by the analog noise generator 20 and the computer system for digital filtering and the processing software.
• the analog noise generator 20 provides a pink to white noise signal, which is generated as described under 1.1.1. Noise signals which have a level drop of 3 dB per octave with rising frequency are referred to as pink. White noise signals, in contrast, display an approximately linear frequency response.
• Transistor T 1 generates the noise signal on its base emitter section, which is operated above the breakthrough voltage.
• Transistor T 2 is used as an impedance converter and amplifier and converts the noise current from T 1 into a noise voltage.
• the noise voltage is capacitively decoupled from T 2 at the collector and fed via a single-stage high-pass filter into the input of the sound card.
• the input-side channel of the sound cards comprises an amplifier, a band pass filter for frequencies from 100 Hz to 15 kHz, a 14-bit analog/digital converter and the interface to the PCI bus of the laptop.
• the sound card samples the low-frequency noise at a clock rate of 44.1 kHz, converts it into 14-bit width signed integer numbers and provides these via the driver software to the processing software.
• FIG. 8 shows the implementation of the noise generator 20 , which is connected to the laptop 21 .
• the processing software filters the numbers thus obtained and extracts the actual useful signal.
• Variant b coupling to the background wave using internal noise generator
• Variant b uses the internal random generator provided in the Pentium III as a noise source (The Intel® Random Generator, Techbrief 1999, Intel®). The additional external module from variant a is thus dispensed with.
• the processing software in variant b contains a driver function for the internal random generator instead of the driver function for activating and reading out the sound card.
• the further software-side processing of the noise signal is identical to variant a.
• Variant b has the disadvantage of restriction to computer systems having Pentium III or Pentium 4 processors.
• Variant c coupling to the background wave using external modem
• the modem 30 contains a broadband analog noise source (noise generator 32 ), an impedance converter 33 , a filter and amplifier 34 , an analog/digital converter 35 , and an interface component (controller 36 ) for the USB bus.
• the modem 30 may contain a microcontroller for digital filtering and preprocessing of the useful signal. These functions may, however, as in variants a and b, be assumed by the processing software on the laptop. The processing of the noise signal and useful signal is performed analogously to variants a and b. Significantly higher data rates are achievable than in variants a and b.
• the preprocessing and filtering of the data obtained by the method described in chapter 1.1.2 is performed through processing software installed on the laptop.
• This software contains, in addition to filters for the typical equalization, a special adaptive global scaling filter which temporarily stores the raw data obtained over a sufficiently long period of time and analyzes it in the time and value range according to the typical global scaling patterns.
• the analysis of the GS patterns is either performed on the basis of histograms over the entire value range of the raw data or on the basis of time in regard to the logarithmic-hyperbolic fluctuations of the individual data in the time range.
• the object of the software is to generate random numbers from the technical noise signals, electrical potentials, etc., which may be processed further later.
• Random numbers which are generated in this way are manipulated by the background wave, which may be determined empirically in that they do not behave statistically correctly when the sampling frequency f A is in proximity to a node point frequency.
• a number n does not occur arbitrarily randomly but rather at a logarithmic hyperbolic interval, similarly as it is calculated according to global scaling.
• a further method is the calculation of binary numbers from the slope of the noise signal in the sampling points.
• a positive slope results in a one and a uniform or negative slope results in a zero.
• the binary random numbers thus obtained may be linked to a progressive zero-one sequence logically via an exclusive-or function (EXOR), in order to obtain the best possible equipartition of zero and one.
• EXOR exclusive-or function
• FIG. 10 shows the mean fluctuation of unfiltered binary numbers over the natural-logarithmic time axis for a period of time of 12 hours.
• the data was obtained using hardware according to described variant a and using software according to equation (5).
• the data was additionally filtered using statistical software. For this purpose, the data was first differentiated by calculating the differential quotients. Subsequently, the differential quotients were added up in periods of time of 10 seconds and integrated using a sliding low pass function over 300 time periods.
• the typical periodic fluctuations expected according to global scaling are shown over the natural-logarithmic time axis in FIG. 5. 71 ⁇ 2 oscillations having constant period time and rising amplitude are shown. The maxima of the oscillation antinodes are at approximately ⁇ 3.6:1.6 minutes, ⁇ 2.7:4.0 minutes, ⁇ 1.8:9.9 minutes, ⁇ 0.9:24.4 minutes, 0.0:1.0 hours, 0.9:2.45 hours, 1.8:60 hours, (2.7:14.8 hours approximately). These oscillation antinodes identify the areas having the greatest fluctuations and are in the global scaling node points.
• random numbers are generated through sampling of the noise signal.
• the sampling of the noise processes in the transmitter and receiver is performed according to the present invention using a GS node point frequency f 0 and thus results in the generation of a GS time sequence of random numbers Z.
• Other node point frequencies may be ascertained using equation (3).
• the GS sampling signal is then converted into a normalized, dimensionless sequence of numeric values (Z), possibly of the value range N, for example, through residual class formation R modulo N (modulo operator) according to the formula Z ⁇ Z modulo N, N being a whole number.
• the random number sequence Z S thus arises at the transmitter S and the random number sequence Z E arises at the receiver E.
• the following sequence of random numbers has resulted through the sampling and is displayed on the monitors of the transmitter and receiver:
• Z S ⁇ . . . 10 23 2500 249 28 378 40456 . . . ⁇
• Z E ⁇ . . . 45 789 4581 45 3 6782 2360 . . . ⁇
• the two random number sequences Z S and Z E at the transmitter and receiver, respectively, are typically not chronologically synchronous without technical measures, however.
• the synchronous sampling may be implemented by the controller via an external radio clock on both terminals, for example.
• the precision of the synchronous clock is to be at least one order of magnitude more precise than the sampling frequency.
• the processing steps must be performed on the transmitter side even before the current random number is ascertained from the noise Z E (t n ) at the receiver at instant t n .
• These alteration speeds of random numbers may also be interpreted as a frequency f, the sampling period ⁇ t s for generating Z S or Z E determining the chronological scale.
• FIG. 5 represents a possible result f S ⁇ ⁇ of the derivation of the signal Z S from a noise process according to FIG. 4.
• a similar sequence of frequency values f E ⁇ ⁇ is calculated for the receiver within the same predefined frequency band.
• a global scaling frequency which may be represented by a GS continued fraction code of the structure [n 0 , n 1 , n 2 ].
• the frequency f R 1889.87 Hz is ascertained, for which a continued fraction code of the structure [n 0 , n 1 , n 2 ] exists.
• the partial denominator n 2 is ⁇ 3 in this example.
• the same frequency f R is found in this case within the frequency band at the transmitter and receiver, i.e., both original random number sequences Z S and Z E have precisely a shared GS alteration speed of their random numbers in the predefined frequency band.
• the GS modulation is performed, for example, through an alteration of the partial denominator n 2 , for example, by a sign change of n 2 .
• the following new continued fraction code [n 0 , n 1 , ⁇ n 2 ] thus results on the transmitter side and a new frequency f R ′ results through reversal of equation (3).
• This frequency f R ′ also mathematically represents an alteration speed of the random numbers and a new random number Z′ S (t n ) is calculated in the transmitter based thereon through the reversal of the derivation according to L. Euler from equation (4), which is coupled into the noise process in the following at the transmitter at instant t 0 .
• the manipulated random number Z′ S (t n ) was calculated on the transmitter side even before a new random number was generated at the transmitter or receiver via the noise process.
• equation (3) is also reversible.
• Z S ⁇ . . . 11( t i+0 )80( t i+1 )3421( t i+2 )345( t i+3 )245( t i+4 )4512( t i+5 )50712( t i+6 ) . . . 192( t n ) ⁇
• the newly calculated random number Z′ S (t n ) is converted into a noise level value having a dimension and coupled into the random process within the sampling period. This conversion is possible since the method of the conversion of the noise level value into random numbers is known and reversible from the preceding method steps.
• the noise on the transmitter side is thus modulated through this coupling of the noise level value belonging to Z′ S (t n ).
• the noise signal in the receiver is decoupled at instant t n by sampling using f 0 and converted into random numbers according to the same method as on the transmitter side.
• the receiver analyzes the frequency band previously tuned with the transmitter from [n 0 , n 1 ⁇ 1] through [n 0 , n 1 +1] and, based on the newly ascertained random number Z′ E (t n ), all existing frequencies within the frequency band through a GS analysis and determines the unique frequency f′ R for which the continued fraction code [n 0 , n 1 ⁇ n 2 ] exists.
• the partial denominator n 2 is determined for this frequency f′ R .
• the shared frequency f′ R 1882.969 Hz is found, for which a continued fraction code of the structure [n 0 , n 1 , n 2 ] exists.
• the partial denominator n 2 is thus +3.
• the receiver may now recognize whether the n 2 value was manipulated on the transmitter side.
• the expected sign of n 2 may be determined using a computer solely from the combination of sampling periods ⁇ t s , n 0 , and n 1 , because the frequency band is uniquely fixed by n 0 and n 1 in that the expected global scaling resonance frequency f R of the random process must be present.
• a frequency f R having the associated continued fraction code [ ⁇ 48, ⁇ 27, ⁇ n 2 ] is expected on the receiver side, which also applies for the non-modulated case in the transmitter on the receiver side.
• the partial denominator n 2 is thus +3.
• the receiver since a n 2 value of ⁇ 3 was expected on the receiver side, the receiver has recognized that the n 2 value of the resonance frequency f R was modulated on the transmitter side.
• the receiver thus recognizes the manipulation on the transmitter side when it is present.
• the technical transmission rate via the random process described here is determined and limited by the execution speed of steps 1 through 8 and by the sampling frequency f 0 . Therefore, transmission rates of 16 bits per second are thus currently implemented.
• An increase of the transmission rate is possible, for example, through the use of other sampling frequencies f 0 , faster computers, improved GS modulation of the continued fraction value n 2 (and/or higher elements of the continued fraction n 3 , n 4 , etc.) instead of only one sign reversal or the parallel use of multiple transmission channels.

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