US20070069705A1 - Circuits and methods for providing multiple phase switching regulators which employ the input capacitor voltage signal for current sensing - Google Patents

Circuits and methods for providing multiple phase switching regulators which employ the input capacitor voltage signal for current sensing Download PDF

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US20070069705A1
US20070069705A1 US11/604,912 US60491206A US2007069705A1 US 20070069705 A1 US20070069705 A1 US 20070069705A1 US 60491206 A US60491206 A US 60491206A US 2007069705 A1 US2007069705 A1 US 2007069705A1
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inductor
voltage
converter
current
converter circuit
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Christopher Umminger
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Analog Devices International ULC
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Linear Technology LLC
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H02M3/158—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load
    • H02M3/1584—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load with a plurality of power processing stages connected in parallel
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H02M3/158—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load
    • H02M3/1588—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load comprising at least one synchronous rectifier element
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H02M3/158—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load
    • H02M3/1584—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load with a plurality of power processing stages connected in parallel
    • H02M3/1586—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load with a plurality of power processing stages connected in parallel switched with a phase shift, i.e. interleaved
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • This invention relates to circuits and methods that enable multiphase or polyphase switching regulators to evenly divide load current between stages (phases) without the use of a current sense resistance or relying on the analysis of switches such as MOSFET devices. More specifically, this invention relates to systems and methods that monitor the time varying voltage change of the input capacitor in multiphase switching regulators and utilize this monitored value to control load sharing.
  • DC to DC regulator circuits are invaluable in situations where the need is present to regulate an input voltage, such as voltages from household power outlets, for a particular product or application. Power techniques present in such DC to DC regulators can solve the problems surrounding fluctuating and poorly-specified input signals. By producing a desired output voltage, either higher or lower than the regulator's input voltage, DC to DC regulators can control and vary an output voltage with respect to time.
  • multiple power converters can be placed in parrallel configurations by utilizing a common input and output node.
  • Parallel configurations can essentially increase the amount of current that is delivered to the common output node, thus increasing the regulator's potential output power.
  • the switching characteristics of multiple parallel power converters can also be shifted in time (phase) with respect to each other. If shifted properly, the power converters can ultimately divide and evenly share a common input current while maintaining a desired regulated output current. Advantages obtained by such multiple phase regulators include reduced input and output ripple currents. As a result, the size of the input and output capacitors, in addition to the size of the inductors utilized by the converter stages, are reduced.
  • the voltage-mode multiphase switching regulator method is one technique for controlling load current. This method utilizes the output voltage to control the amount of time, or duty cycle, that a converter stage's main switch is ON. By turning a converter's main switch ON, current is allowed to flow from the regulator's common input node, through the individual converter's inductor, to the regulator's common output node. By controlling the amount of time that a converter's main switch is ON, the amount of current delivered to the common output of the regulator is controlled. Examples of multiphase switching regulators that employ the voltage-mode control technique include Semtech's SC1144 of Camarillo, Calif. and National Semiconductor's LM2639 of Austin, Tex.
  • One alternate approach employs a sense resistor at the input of each individual converter stage.
  • the current across an individual sense resistor may be monitored and used to control the duty cycle of the respective converter stage.
  • the distribution of load current throughout the switching regulator is regulated.
  • a voltage-mode multiphase switching regulator that employs input current sense resistors to balance inductor currents is Linear Technology's LTC1702 of Milpitas, Calif.
  • Regulators that utilize sense resistors for balancing inductor current within the individual converter stages still have disadvantages.
  • current sense resistors cause additional dissipative losses in regulators which, in turn, reduce operating efficiency; the ratio of the power supplied to power provided to a regulator (where a ratio of unity, 1/1, is ideal).
  • sense resistors must have a very low resistive value with typical values residing in the 0.001 ⁇ to 0.1 ⁇ range. These sense resistors must have good matching characteristics. Resistors with low resistance values and good matching characteristics, however, are costly and difficult to manufacture.
  • Another known technique helps match average inductor current between stages while eliminating the use of sense resistors.
  • the technique uses the ON-resistance of each converter's power MOSPET switches to determine the amount of current flowing through the individual stages.
  • Examples of voltage-mode multiphase switching regulators that use on-board MOSFET switch resistance to control and balance current sharing throughout a regulator are Harris' HIP6301, HIP6302, and HIP6303.
  • the matching characteristics, however, of power MOSFET switches are poor, typically having a loose specification in the +/ ⁇ 30% range. Such matching characteristics are significantly worse than those found in sense resistors. Moreover, there is no guarantee that matching can be achieved between the parts of each individual MOSFET switch. Such disadvantages make MOSFET sense regulators problematic and deficient.
  • the present invention provides multiple phase switching regulators which utilize the voltage signal across the input capacitor in order to efficiently control load sharing between individual converter stages. Load balancing in a switching regulator of the present invention is realized through a correction based on the change in charge of the input capacitor.
  • the rate of change of the charge of the input capacitor is proportional to the current flowing through the inductors of those converter stages which have a main MOSFET switch that is ON.
  • a converter's main switch ON current is allowed to flow from the regulator's input node, through the individual converter's inductor, to the regulator's output node.
  • the amount of current delivered to the common output of the regulator is controlled. Therefore, by utilizing the charge of the input capacitor and manipulating the duty cycle of one or more converter stages, the distribution of load current throughout the regulator can also be determined and manipulated.
  • a sampling circuit is included in the present invention to measure the voltage across the input capacitor at specified intervals. Particularly, sampled voltage differences are measured when a converter's main MOSFET switch turns ON and OFF, thereby measuring the voltage difference of each cycle. In comparing these differences, a converter's inductor current may ultimately be equalized to inductor currents of the other converter stages. In this manner, the output load of the switching regulator is balanced evenly between each converter circuit when the changes in each separate stage's inductor currents are equivalent.
  • the sampling circuit of the present invention may also be used to measure any regulator component which has a voltage change proportional to the current in a converter stage's inductor.
  • the present invention preferably only needs to sample the voltage across the input capacitor. As a result, only one component is depended upon to take a current reading for every stage of the switching regulator. Thus, matching characteristics are less problematic.
  • the input capacitor is already present in most multiple phase switching regulator systems as a filtration and stabilizing device. This allows for regulators to be manufactured that are smaller, cheaper, less problematic; and more efficient.
  • FIG. 1 is a schematic diagram of a prior-art voltage-mode controlled two phase DC-DC switching regulator
  • FIG. 2 is a schematic diagram of a voltage-mode controlled two phase DC-DC switching regulator that employs current sensing of the input capacitor voltage for load sharing constructed in accordance with the principles of the present invention
  • FIGS. 3A-3D are illustrations of current and voltage waveforms when the duty cycle of each power stage of the two-phase voltage regulator constructed in accordance with principles of the present invention is less than 50%;
  • FIGS. 4A-4D are illustrations of current and voltage waveforms when the duty cycle of each power stage of the two-phase voltage regulator constructed in accordance with the principles of the present invention is greater than 50%.
  • FIG. 1 shows a prior art two-phase voltage-mode regulator 100 .
  • Voltage-mode regulator 100 includes converter stages 110 and 120 and utilizes the common output voltage to control the amount of time, or duty cycle, that the stages' respective main switches 111 and 121 are turned ON. In controlling the amount of time main switches 111 and 121 are turned ON, the amount of current at output node 192 is also controlled.
  • Converter circuit 110 operates as described below. At the beginning of each cycle, phase-one ramp output voltage V R1 from ramp generator 150 is reset to zero at node 193 and begins increasing. As a result, the negative input of pulse-width-modulator (PWM) comparator 114 turns LOW (e.g., zero volts). If the negative input voltage of PWM comparator 114 is less than control voltage V c effecting the positive input of PWM comparator 114 at node 195 , the output of PWM comparator 114 turns HIGH (e.g., 5 volts).
  • PWM pulse-width-modulator
  • phase-one voltage V R1 from ramp generator 150 becomes large enough to exceed control voltage V c at node 195 , the output of PWM comparator 114 turns LOW.
  • the output of driver 115 turns LOW and MOSFET switch 111 turns OFF while the output of inverting driver 116 turns HIGH and synchronous MOSFET switch 113 turns ON.
  • MOSFET switch 111 is OFF, current is not able to reach inductor 112 from node 191 and because switch 113 is ON current present in inductor 112 is able to flow from ground 170 .
  • the output of PWM comparator 114 turns LOW, a voltage of ⁇ V OUT is applied across inductor 112 , causing the current in inductor 112 to decrease.
  • phase-one voltage V R1 from ramp generator 150 resets to zero at the beginning of each cycle.
  • phase-one voltage V R1 resets, it is less than control voltage V C .
  • MOSFET switches 111 and 113 turn ON and OFF respectively, subsequently increasing the current in inductor 112 . Therefore, the control voltage V C determines the amount of time, or duty cycle, that main switch 111 is turned ON, which directly affects the amount of current in inductor 112 .
  • Converter circuit 120 is similar in operation to converter circuit 110 except that converter circuit 120 is governed by phase-two ramp voltage V R2 from ramp generator 150 at node 194 . At some instance during the cycle of phase-one ramp voltage V R1 , usually halfway, the delayed phase-two output voltage V R2 of ramp generator 150 is reset to zero and begins increasing.
  • main MOSFET switch 121 When phase-two output voltage V R2 is less than control voltage V C , main MOSFET switch 121 is ON and current from input node 191 may reach inductor 122 . Additionally, inverting driver 126 turns synchronous MOSFET switch 123 OFF so that current at the input of inductor 122 is restricted from flowing from ground 170 . Because MOSFET switch 121 is ON and synchronous MOSFET switch 123 is OFF, the current in inductor 122 increases.
  • phase-two output voltage V R2 is greater than control voltage V C , main MOSFET switch 121 is OFF. Additionally, inverting driver 126 turns synchronous MOSFET switch 123 ON and allows current at node 197 to flow from ground 170 . Because MOSFET switch 121 is OFF and synchronous MOSFET switch 123 is ON, the current in inductor 122 decreases.
  • main MOSFET switch 121 in converter circuit 120 is delayed relative to the switching of main MOSFET switch 111 in converter circuit 110 and may be, for example, a pre-determined time (e.g. one half period). Similar to converter circuit 110 , control voltage V c determines the amount of time, or duty cycle, that main MOSFET switch 121 is ON. As a result, control voltage V c directly affects the amount of current in inductor 122 as well as inductor 112 .
  • Control voltage V c at node 195 in regulator 100 is determined by resistive divider 182 , error amplifier 160 , compensation components 181 , and PWM comparators 114 and 124 . For this reason, these components also determine the state of main MOSFET switches 111 and 121 and control the duty cycles of converter circuits 110 and 120 . In controlling the duty cycles of the converter circuits in regulator 100 , the amount of current in each converter's inductors (e.g., inductor 112 and 122 ) is also controlled. The adjustment of the control voltage V c in order to distribute load current is a result of output voltage V OUT and is described below.
  • Control voltage V C is continuously adjusted in response to changes in output voltage V OUT such that output voltage V OUT is regulated and maintained ideally constant.
  • impedance 106 is included in regulator 100 . Impedance 106 serves to filter and isolate the switched input currents of regulator 100 through input capacitor 131 so that the current supplied from input sources is substantially DC.
  • FIG. 2 is a simplified schematic diagram of switching regulator 200 in accordance with the principles of the present invention.
  • Regulator 200 utilizes the voltage across input capacitor 231 for current sensing in order to control and balance current load sharing throughout the converter stages of regulator 200 .
  • Regulator 200 may contain impedance 206 , which may separate input voltage V IN at node 291 from input capacitor 231 and the rest of regulator 200 .
  • Impedance 206 serves to filter or isolate the switched input currents of the regulator through input capacitor 231 so that the current from input node 291 is substantially DC.
  • impedance 206 may consist of both a parasitic resistive element and an inductance that may be parasitic or explicitly added.
  • Switched, or AC, currents through input capacitor 231 will cause the voltage across input capacitor 231 to vary with time during the various switching cycles of the main MOSFET switches in the converter stages of regulator 200 .
  • the change in voltage across input capacitor 231 is proportional to the sum of the currents in the inductors of all converter stages which have a main MOSFET switch that is ON.
  • the variations of the voltage across capacitor 231 may be selectively sampled in order to measure the voltage change over specified time intervals or during particular events.
  • the voltage across capacitor 231 may be sampled at the instances when a converter's main MOSFET switch changes states. Accordingly, the difference in these samples may be used to measure the voltage increase or decrease across capacitor 231 when a particular converter's main MOSFET switch is ON.
  • the variations of the voltage across capacitor 231 are proportional to the current in the inductors of the regulator's stages.
  • load balancing between the converter stages of regulator 200 may be accomplished.
  • a correction may be made by changing the inductor current in one or more converter stages. Such a correction may, for example, be utilized when there is a need to equalize the currents in each stage's primary inductor.
  • converter stages 210 and 220 of regulator 200 operate in the same manner as converter stages 110 and 120 of regulator 100 from FIG. 1 .
  • Regulator 200 employs the voltage across input capacitor 231 for use in current sensing and load distribution. Measuring the voltage across input capacitor 231 is accomplished by sampling circuitry 285 , which, in turn, is controlled by sampling control circuitry 290 .
  • sampling control circuitry 290 is controlled by sampling control circuitry 290 .
  • the range and levels of HIGH and LOW voltage states for regulator 200 may vary with different design choices, performance goals, and applications requirements.
  • sampling control circuitry 290 momentarily closes switch 271 and samples the voltage across input capacitor 231 onto hold capacitor 235 . Accordingly, when main MOSFET switch 211 turns OFF, sampling control circuitry 290 momentarily closes switch 272 and samples the voltage across input capacitor 231 onto hold capacitor 236 .
  • Summing junction 283 takes the difference between the voltages on hold capacitors 235 and 236 .
  • This difference is the voltage change of capacitor 231 when phase-one voltage ramp voltage V R1 is less than control voltage V C1 . Accordingly, this time interval is when main MOSFET switch 211 is ON and allows load current to reach inductor 212 . As a result, the measurement of voltage change of capacitor 231 during this time is proportional to the current of inductor 212 .
  • sampling control circuitry 290 momentarily closes switch 273 and samples the voltage across input capacitor 231 onto hold capacitor 237 . Accordingly, when main MOSFET switch 221 turns OFF, sampling control circuitry 290 momentarily closes switch 274 and samples the voltage across input capacitor 231 onto hold capacitor 238 .
  • Summing junction 284 takes the difference between the voltages on hold capacitors 237 and 238 .
  • This difference is the voltage change of capacitor 231 when phase-two voltage ramp voltage V R2 is less than control voltage V C2 . Accordingly, this time interval is when main MOSFET switch 221 is ON and allows load current to reach inductor 222 . As a result, the measurement of voltage change of capacitor 231 during this time is proportional to the current of inductor 222 .
  • the voltage differences across capacitor 231 when main MOSFET switches 211 and 221 are ON are then utilized, if appropriate, to correct and balance the load distribution between converter circuits 210 and 220 . More particularly, these voltage differences, located at the outputs of summing junctions 283 and 284 , are utilized by error amplifier 263 so that corrections may performed to evenly distribute load current throughout regulator 200 .
  • load current can be distributed in different proportions between converter stages. For example, suppose that an application exists in which the load current to converter circuit 210 needs to be greater than the load current to converter circuit 220 . To accomplish this desired power distribution, the respective control voltages may be controlled by controlling the ON times of the main MOSFET switches.
  • the correction of load current between the converter stages of regulator 200 is accomplished as follows.
  • Current sharing amplifier 263 compares the voltage change of input capacitor 231 during phase-one MOSFET switch 211 ON time with the voltage change of input capacitor 231 during phase-two MOSFET switch 221 ON time. If differences exist in the voltage change of input capacitor 231 between phase-one and phase-two ON time, the output of current sharing amplifier 263 may accordingly adjust the input voltage to error amplifier 262 . In this manner, control voltage V C2 to converter 220 may be adjusted. Control voltage V C2 will be adjusted as a result of any differences in capacitor voltages until the phase-one and phase-two voltage changes of capacitor 231 are equal. When this occurs, the two inductor currents are also equal as will be described below. Essentially, the currents in inductor 212 and 222 will be measured and continuously adjusted so that the load current can be balanced between them.
  • Control voltage V C2 will then increase, thus increasing the amount of time that main MOSFET switch 221 is ON and the duty cycle of converter 220 .
  • the amount of current in inductor 222 will increase and the difference in voltage of input capacitor 231 between future phase-one MOSFET switch 211 ON and phase-two MOSFET switch 221 ON cycles will decrease.
  • phase-one MOSFET switch 211 ON time is less than the voltage decrease in phase-two MOSFET switch 221 ON time.
  • This example is reflective of the case when the current in inductor 212 is less than the current in inductor 222 .
  • current sharing amplifier 263 will increase the input to error amplifier 262 which will decrease control voltage V C2 .
  • control voltage V C2 is decreased, the amount of time that main MOSFET switch 221 is ON will decrease as well as the duty cycle of converter 220 .
  • the amount of current in inductor 222 will decrease and the magnitude of future changes in voltage of input capacitor 231 for future phase-two MOSFET switch 221 ON cycles decreases.
  • compensation components 282 , 244 , and 281 are chosen in such a way to help manipulate the effect of amplifier 263 , error control device 262 and control voltage V C2 .
  • the control voltage V C of converter circuits 210 and 220 is also manipulated by error dividers 289 and 282 , error control amplifiers 261 and 262 , and compensation components 286 and 281 .
  • regulator 200 tracks the current in inductor 212 and then corrects the current in inductor 222 to match that of inductor 212 . This is accomplished by sampling the voltage of input capacitor 231 and using this information to control the ON time of main MOSFET switch 221 of converter circuit 220 . Similarity, persons skilled in the art will appreciate that instead of tracking inductor 212 and manipulating inductor 222 , regulator 200 can be configured so that inductor 212 is tracked to inductor 222 . This may be accomplished by simply providing the sampled information to converter circuit 210 instead of converter circuit 220 . One example may simply involve connecting the output of amplifier 263 to the input of the negative terminal of amplifier 261 and exchanging the positive and negative inputs of amplifier 263 .
  • any one or more inductors (e.g., converter stages) of the switching regulator may be designed to track any one or more different inductors (e.g., converter stages).
  • a plurality of converter stages may also be used to track a plurality of different converter stages.
  • input capacitor 231 may be located elsewhere in regulator 200 and still be sampled for load sharing.
  • capacitor 231 may be an output capacitor similar to capacitor 233 .
  • principles of the present invention may be used to sample other elements that could be employed in a switching regulator. Particularly, principles of the present invention could allow for an easily tailored circuit that measures any component whose voltage can distinguish one or more converter stages' inductor currents.
  • FIGS. 3A-3D illustrate the case when the duty cycle of converter stages within a two-converter stage switching regulator is less than 50% such that the main MOSFET switches for the converter stages are never on simultaneously.
  • FIGS. 3A-3D have reference numerals of the form 3XX and denote the same structural elements throughout.
  • electrical events of electrical components that are described in the discussion of FIG. 2 are also illustrated in FIGS. 3A-3D .
  • current I 206 of FIG. 3A represents the current flowing through impedance 206 of FIG. 2 with respect to time.
  • reference numerals that contain subscript of the form 2XX included in FIGS. 3A-3D map to electrical events of corresponding 2XX electrical components of FIG. 2 .
  • FIG. 3A illustrates waveform 310 of current I 206 with respect to time.
  • FIG. 3B illustrates waveform 320 of current I 211 with respect to time.
  • I 211 is positive when MOSFET switch 211 is ON.
  • FIG. 3C illustrates waveform 330 of current I 221 with respect to time.
  • I 221 is positive when MOSFET switch 221 is ON.
  • FIG. 3D illustrates waveform 340 of the voltage V 231 across capacitor 231 . At the beginning of the first converter stage's duty cycle, switch 211 is turned ON while switch 221 is OFF. The voltage of capacitor 231 is represented by point 341 of FIG. 3D .
  • voltage V 231 across capacitor 231 begins to decrease. This decrease in voltage V 231 occurs because current I 211 through switch 211 and inductor 212 exceeds input current I 206 through impedance 206 .
  • the amount of decrease in voltage V 23 is proportional to the charge removed from capacitor 231 .
  • the amount of charge removed from capacitor 231 is proportional to the current flowing in inductor 212 and the duration of the on-time of switch 211 .
  • Point 342 occurs when switch 211 turns OFF and voltage V 231 begins to rise as input current I 206 recharges input capacitor 231 .
  • Point 343 occurs when switch 221 turns ON, the voltage will again decrease while the current I 203 through switch 221 and inductor 222 removes charge from capacitor 231 .
  • Point 344 occurs when switch 221 turns OFF, the voltage across input capacitor 231 will again rise until switch 211 turns on again at the beginning of the next cycle. If the current in inductor 212 is larger than the current in inductor 222 , the voltage decrease across the input capacitor 231 will be larger during the first phase (between points 341 and 342 ) while switch 211 is ON than during the second phase (between points 343 and 344 ) when switch 221 is ON. This asymmetry in the voltage waveform is utilized to correct the load sharing between the converters 210 and 220 .
  • FIGS. 4A-4D illustrate the case when the duty cycle of converter stages within a two-converter stage switching regulator is greater than 50% such that the main MOSFET switches for the converter stages spend a portion of the cycle conducting simultaneously.
  • FIGS. 4A-4D have reference numerals of the form 4 XX and denote the same structural elements throughout. However, electrical events of electrical components that are described in the discussion of FIG. 2 are also illustrated in. FIGS. 4A-4D .
  • current I 206 of FIG. 4A represents the current flowing through impedance 206 of FIG. 2 with respect to time.
  • reference numerals that contain subscript of the form 2XX included in FIGS. 4A-4D map to electrical events of corresponding 2XX electrical components of FIG. 2 .
  • FIG. 4A illustrates waveform 410 of current I 206 with respect to time.
  • FIG. 4B illustrates waveform 420 of current I 211 with respect to time.
  • I 233 is positive when MOSFET switch 211 is ON.
  • FIG. 4C illustrates waveform 430 of current I 221 with respect to time.
  • I 211 is positive when MOSFET switch 221 is ON.
  • FIG. 4D illustrates waveform 440 of the voltage V 232 across capacitor 231 .
  • the present invention is not limited to the embodiments described above that use voltage-mode switching regulators and sample the input capacitor. Instead, the present invention more generally involves implementing a load balancing procedure which utilizes sampling of a component whose voltage is proportional to the current through a regulator circuit's inductor. Moreover, principles of the present invention may be used in a regulator with a single converter stage.
  • Converter stage control can also be based on the sensing of an output capacitor voltage change (e.g., MAX 1710 “Quick-PWM” and Cherry Semiconductor's “V 2 ”).
  • the ramp generator that provides timing and control signals to the regulator of the present invention can be comprised of a variety of signals with different frequencies and amplitudes and does not necessarily have to be a generator, but a device used for the same purpose.
  • a wide variety of techniques can also be used to measure the input capacitor waveform and is not limited to a mere sampling circuit. Additional signal processing of the input capacitor waveform may occur, such as filtering or integration, that can be used for ease of implementation or to improve performance such as reducing the effect of switching noise on the signal.
  • This invention can also be used in other switching regulator topologies that use multiple phases such as boost or buck-boost topologies and with or without synchronous switches.
  • inverter 216 of FIG. 2 and MOSFET switch 213 of FIG. 2 may be replaced with a diode.
  • the present invention can also be used with output capacitor voltage sensing rather than, or in conjunction with, input capacitor voltage sensing. This is particularly advantageous with boost or buck-boost converters.
  • the apparatus of the present invention may be implemented using circuit configurations other than those shown and discussed above.
  • the present invention also may be implemented in single-phase and polyphase switching regulators implementing a variety of power converters such as DC-To-DC, and AC-To-DC. All such modifications are within the scope of the present invention, which is limited only by the claims that follow.

Abstract

Circuits and methods to correct the load sharing in multiphase switching regulators are provided. Using these systems and methods, the input capacitor voltage signal can be sampled and used for current sensing of the regulator's stages. Differences in the amount of output current for a converter stage can then be determined. Corrections needed to equalize the output current of the converter stages can then be determined and carried out.

Description

    BACKGROUND OF THE INVENTION
  • This invention relates to circuits and methods that enable multiphase or polyphase switching regulators to evenly divide load current between stages (phases) without the use of a current sense resistance or relying on the analysis of switches such as MOSFET devices. More specifically, this invention relates to systems and methods that monitor the time varying voltage change of the input capacitor in multiphase switching regulators and utilize this monitored value to control load sharing.
  • DC to DC regulator circuits (power converters) are invaluable in situations where the need is present to regulate an input voltage, such as voltages from household power outlets, for a particular product or application. Power techniques present in such DC to DC regulators can solve the problems surrounding fluctuating and poorly-specified input signals. By producing a desired output voltage, either higher or lower than the regulator's input voltage, DC to DC regulators can control and vary an output voltage with respect to time.
  • When particularly high power levels need to be delivered to an output, multiple power converters can be placed in parrallel configurations by utilizing a common input and output node. Parallel configurations can essentially increase the amount of current that is delivered to the common output node, thus increasing the regulator's potential output power. The switching characteristics of multiple parallel power converters can also be shifted in time (phase) with respect to each other. If shifted properly, the power converters can ultimately divide and evenly share a common input current while maintaining a desired regulated output current. Advantages obtained by such multiple phase regulators include reduced input and output ripple currents. As a result, the size of the input and output capacitors, in addition to the size of the inductors utilized by the converter stages, are reduced.
  • The voltage-mode multiphase switching regulator method is one technique for controlling load current. This method utilizes the output voltage to control the amount of time, or duty cycle, that a converter stage's main switch is ON. By turning a converter's main switch ON, current is allowed to flow from the regulator's common input node, through the individual converter's inductor, to the regulator's common output node. By controlling the amount of time that a converter's main switch is ON, the amount of current delivered to the common output of the regulator is controlled. Examples of multiphase switching regulators that employ the voltage-mode control technique include Semtech's SC1144 of Camarillo, Calif. and National Semiconductor's LM2639 of Austin, Tex.
  • One disadvantage of a voltage-mode multiphase switching regulator exists in the difficulty to control currents in each individual power stage. Slight differences in switch ON-times between phases can result in significant inequality in load sharing between stages. Additionally, as a result of the unequal parasitic resistances that exist between each individual stage's MOSFET switches and inductors, the average inductor currents in such a regulator are unequal. In order to fix such inequality errors, significantly larger MOSFET switches and inductors must be used to accommodate the unequal load sharing. However, this solution to the power distribution problem increases production costs and reduces power efficiency.
  • One alternate approach employs a sense resistor at the input of each individual converter stage. The current across an individual sense resistor may be monitored and used to control the duty cycle of the respective converter stage. In changing the duty cycle of the individual converter stage, the distribution of load current throughout the switching regulator is regulated. One example of a voltage-mode multiphase switching regulator that employs input current sense resistors to balance inductor currents is Linear Technology's LTC1702 of Milpitas, Calif.
  • Regulators that utilize sense resistors for balancing inductor current within the individual converter stages, however, still have disadvantages. For example, current sense resistors cause additional dissipative losses in regulators which, in turn, reduce operating efficiency; the ratio of the power supplied to power provided to a regulator (where a ratio of unity, 1/1, is ideal). Yet, in order to prevent dissipative losses from becoming excessive, sense resistors must have a very low resistive value with typical values residing in the 0.001Ω to 0.1Ω range. These sense resistors must have good matching characteristics. Resistors with low resistance values and good matching characteristics, however, are costly and difficult to manufacture. These regulators also must employ a capacitor, in addition to the sense resistor, for each converter circuit, thereby increasing production costs and design complexity. Finally, sense resistors are physically large and require valuable circuit board area. For at least portable product designs, circuit board area is limited, thus limiting the amount of portable applications in which switching regulators with sense resistors may be beneficial.
  • Another known technique helps match average inductor current between stages while eliminating the use of sense resistors. The technique uses the ON-resistance of each converter's power MOSPET switches to determine the amount of current flowing through the individual stages. Examples of voltage-mode multiphase switching regulators that use on-board MOSFET switch resistance to control and balance current sharing throughout a regulator are Harris' HIP6301, HIP6302, and HIP6303. The matching characteristics, however, of power MOSFET switches are poor, typically having a loose specification in the +/−30% range. Such matching characteristics are significantly worse than those found in sense resistors. Moreover, there is no guarantee that matching can be achieved between the parts of each individual MOSFET switch. Such disadvantages make MOSFET sense regulators problematic and deficient.
    • SUMMARY OF THE INVENTION
  • The present invention provides multiple phase switching regulators which utilize the voltage signal across the input capacitor in order to efficiently control load sharing between individual converter stages. Load balancing in a switching regulator of the present invention is realized through a correction based on the change in charge of the input capacitor.
  • At any one time the rate of change of the charge of the input capacitor is proportional to the current flowing through the inductors of those converter stages which have a main MOSFET switch that is ON. By turning a converter's main switch ON, current is allowed to flow from the regulator's input node, through the individual converter's inductor, to the regulator's output node. Through controlling the amount of time that a converter's main switch is ON and determining the current in a converter's inductor, the amount of current delivered to the common output of the regulator is controlled. Therefore, by utilizing the charge of the input capacitor and manipulating the duty cycle of one or more converter stages, the distribution of load current throughout the regulator can also be determined and manipulated.
  • A sampling circuit is included in the present invention to measure the voltage across the input capacitor at specified intervals. Particularly, sampled voltage differences are measured when a converter's main MOSFET switch turns ON and OFF, thereby measuring the voltage difference of each cycle. In comparing these differences, a converter's inductor current may ultimately be equalized to inductor currents of the other converter stages. In this manner, the output load of the switching regulator is balanced evenly between each converter circuit when the changes in each separate stage's inductor currents are equivalent. The sampling circuit of the present invention may also be used to measure any regulator component which has a voltage change proportional to the current in a converter stage's inductor.
  • The present invention preferably only needs to sample the voltage across the input capacitor. As a result, only one component is depended upon to take a current reading for every stage of the switching regulator. Thus, matching characteristics are less problematic.
  • Furthermore, the input capacitor is already present in most multiple phase switching regulator systems as a filtration and stabilizing device. This allows for regulators to be manufactured that are smaller, cheaper, less problematic; and more efficient.
    • BRIEF DESCRIPTION OF THE INVENTION
  • The above-mentioned objects and features of the present invention can be more clearly understood from the following detailed description considered in conjunction with the following drawings, in which the same reference numerals denote the same structural elements throughout, and in which:
  • FIG. 1 is a schematic diagram of a prior-art voltage-mode controlled two phase DC-DC switching regulator;
  • FIG. 2 is a schematic diagram of a voltage-mode controlled two phase DC-DC switching regulator that employs current sensing of the input capacitor voltage for load sharing constructed in accordance with the principles of the present invention;
  • FIGS. 3A-3D are illustrations of current and voltage waveforms when the duty cycle of each power stage of the two-phase voltage regulator constructed in accordance with principles of the present invention is less than 50%; and
  • FIGS. 4A-4D are illustrations of current and voltage waveforms when the duty cycle of each power stage of the two-phase voltage regulator constructed in accordance with the principles of the present invention is greater than 50%.
    • DETAILED DESCRIPTION OF THE INVENTION
  • FIG. 1 shows a prior art two-phase voltage-mode regulator 100. Voltage-mode regulator 100 includes converter stages 110 and 120 and utilizes the common output voltage to control the amount of time, or duty cycle, that the stages' respective main switches 111 and 121 are turned ON. In controlling the amount of time main switches 111 and 121 are turned ON, the amount of current at output node 192 is also controlled.
  • Converter circuit 110 operates as described below. At the beginning of each cycle, phase-one ramp output voltage VR1 from ramp generator 150 is reset to zero at node 193 and begins increasing. As a result, the negative input of pulse-width-modulator (PWM) comparator 114 turns LOW (e.g., zero volts). If the negative input voltage of PWM comparator 114 is less than control voltage Vc effecting the positive input of PWM comparator 114 at node 195, the output of PWM comparator 114 turns HIGH (e.g., 5 volts).
  • When the output of PWM comparator 114 turns HIGH, the output of driver 115 also becomes HIGH and MOSFET switch 111 will turn ON. This allows current to pass through MOSFET switch 111 and reach inductor 112, which ultimately affects the current at output node 192, as will be explained below. Additionally, the output of inverting driver 116 turns LOW, as a result of the HIGH output voltage from PWM comparator 114, and turns synchronous MOSFET switch 113 OFF.
  • For the above reasons, when the output of PWM comparator 114 turns HIGH, a voltage of approximately VIN−VOUT, where VIN is the voltage at node 191 and VOUT is the voltage at node 192, exists across inductor 112 which causes current in inductor 112 to increase. When MOSFET switch 111 is ON, synchronous MOSFET switch 113 is OFF. As a result, current at node 196 is not affected by MOSFET 113 since this current is not able to reach ground 170.
  • Conversely, when phase-one voltage VR1 from ramp generator 150 becomes large enough to exceed control voltage Vc at node 195, the output of PWM comparator 114 turns LOW. As a result, the output of driver 115 turns LOW and MOSFET switch 111 turns OFF while the output of inverting driver 116 turns HIGH and synchronous MOSFET switch 113 turns ON. Because MOSFET, switch 111 is OFF, current is not able to reach inductor 112 from node 191 and because switch 113 is ON current present in inductor 112 is able to flow from ground 170. When the output of PWM comparator 114 turns LOW, a voltage of −VOUT is applied across inductor 112, causing the current in inductor 112 to decrease.
  • The decrease in current in inductor 112 exists until phase-one voltage VR1 from ramp generator 150 resets to zero at the beginning of each cycle. As shown above, when phase-one voltage VR1 resets, it is less than control voltage VC. As a result, MOSFET switches 111 and 113 turn ON and OFF respectively, subsequently increasing the current in inductor 112. Therefore, the control voltage VC determines the amount of time, or duty cycle, that main switch 111 is turned ON, which directly affects the amount of current in inductor 112.
  • Converter circuit 120 is similar in operation to converter circuit 110 except that converter circuit 120 is governed by phase-two ramp voltage VR2 from ramp generator 150 at node 194. At some instance during the cycle of phase-one ramp voltage VR1, usually halfway, the delayed phase-two output voltage VR2 of ramp generator 150 is reset to zero and begins increasing.
  • When phase-two output voltage VR2 is less than control voltage VC, main MOSFET switch 121 is ON and current from input node 191 may reach inductor 122. Additionally, inverting driver 126 turns synchronous MOSFET switch 123 OFF so that current at the input of inductor 122 is restricted from flowing from ground 170. Because MOSFET switch 121 is ON and synchronous MOSFET switch 123 is OFF, the current in inductor 122 increases.
  • When phase-two output voltage VR2 is greater than control voltage VC, main MOSFET switch 121 is OFF. Additionally, inverting driver 126 turns synchronous MOSFET switch 123 ON and allows current at node 197 to flow from ground 170. Because MOSFET switch 121 is OFF and synchronous MOSFET switch 123 is ON, the current in inductor 122 decreases.
  • The switching action of main MOSFET switch 121 in converter circuit 120 is delayed relative to the switching of main MOSFET switch 111 in converter circuit 110 and may be, for example, a pre-determined time (e.g. one half period). Similar to converter circuit 110, control voltage Vc determines the amount of time, or duty cycle, that main MOSFET switch 121 is ON. As a result, control voltage Vc directly affects the amount of current in inductor 122 as well as inductor 112.
  • Control voltage Vc at node 195 in regulator 100 is determined by resistive divider 182, error amplifier 160, compensation components 181, and PWM comparators 114 and 124. For this reason, these components also determine the state of main MOSFET switches 111 and 121 and control the duty cycles of converter circuits 110 and 120. In controlling the duty cycles of the converter circuits in regulator 100, the amount of current in each converter's inductors (e.g., inductor 112 and 122) is also controlled. The adjustment of the control voltage Vc in order to distribute load current is a result of output voltage VOUT and is described below.
  • If the output voltage VOUT at node 192 decreases, the resulting voltage drop at the input of error control amplifier 160 will cause a temporary increase in control voltage Vc to appear across compensation components 181. For this reason, PWM comparators 114 and 124 increase the duty cycle of the main switches 111 and 121 and decrease the duty cycle of synchronous switches 113 and 123. As previously mentioned, the duty cycles of converter circuits 110 and 120 are the duty cycles of the main MOSFET switches ill and 121. Therefore, output voltage VOUT at node 192 increases until the negative input of error control amplifier 160 matches its reference voltage VREF. Generally, when control voltage VC increases, the amount of time that main MOSFET switches 111 and 121 are ON also increases.
  • Conversely, if output voltage VOUT, at node 192 increases, the resulting voltage increase at the input of error control amplifier 160 causes a temporary decrease in control voltage VC that appears across compensation components 181. As a result, PWM comparators 114 and 124 decrease the duty cycle of the main switches 111 and 121 and increase the duty cycle of the synchronous switches 113 and 123. Therefore, output voltage VOUT, decreases until the positive input to error amplifier 160 matches its reference VREF. Generally, when control voltage VC decreases, the amount of time that the main MOSFET switches 111 and 121 are OFF also increases.
  • Control voltage VC is continuously adjusted in response to changes in output voltage VOUT such that output voltage VOUT is regulated and maintained ideally constant. To create practical power supply designs, impedance 106 is included in regulator 100. Impedance 106 serves to filter and isolate the switched input currents of regulator 100 through input capacitor 131 so that the current supplied from input sources is substantially DC.
  • FIG. 2 is a simplified schematic diagram of switching regulator 200 in accordance with the principles of the present invention. Regulator 200 utilizes the voltage across input capacitor 231 for current sensing in order to control and balance current load sharing throughout the converter stages of regulator 200.
  • Regulator 200 may contain impedance 206, which may separate input voltage VIN at node 291 from input capacitor 231 and the rest of regulator 200. Impedance 206 serves to filter or isolate the switched input currents of the regulator through input capacitor 231 so that the current from input node 291 is substantially DC. Additionally, impedance 206 may consist of both a parasitic resistive element and an inductance that may be parasitic or explicitly added.
  • Switched, or AC, currents through input capacitor 231 will cause the voltage across input capacitor 231 to vary with time during the various switching cycles of the main MOSFET switches in the converter stages of regulator 200. Particularly, the change in voltage across input capacitor 231 is proportional to the sum of the currents in the inductors of all converter stages which have a main MOSFET switch that is ON.
  • The variations of the voltage across capacitor 231 may be selectively sampled in order to measure the voltage change over specified time intervals or during particular events. In preferred embodiments, the voltage across capacitor 231 may be sampled at the instances when a converter's main MOSFET switch changes states. Accordingly, the difference in these samples may be used to measure the voltage increase or decrease across capacitor 231 when a particular converter's main MOSFET switch is ON.
  • Generally, the variations of the voltage across capacitor 231 are proportional to the current in the inductors of the regulator's stages. By determining the current in each converter stage's inductor, load balancing between the converter stages of regulator 200 may be accomplished. If needed or desired, a correction may be made by changing the inductor current in one or more converter stages. Such a correction may, for example, be utilized when there is a need to equalize the currents in each stage's primary inductor.
  • Persons skilled in the art will appreciate that converter stages 210 and 220 of regulator 200 operate in the same manner as converter stages 110 and 120 of regulator 100 from FIG. 1. Regulator 200, however, employs the voltage across input capacitor 231 for use in current sensing and load distribution. Measuring the voltage across input capacitor 231 is accomplished by sampling circuitry 285, which, in turn, is controlled by sampling control circuitry 290. Persons skilled in the art will also appreciate that the range and levels of HIGH and LOW voltage states for regulator 200 may vary with different design choices, performance goals, and applications requirements.
  • For converter circuit 210, the sampling of current in inductor 212 for phase-one voltage VR1 occurs as follows. When main MOSFET switch 211 turns ON and allows input voltage VIN from node 291 to reach inductor 212, sampling control circuitry 290 momentarily closes switch 271 and samples the voltage across input capacitor 231 onto hold capacitor 235. Accordingly, when main MOSFET switch 211 turns OFF, sampling control circuitry 290 momentarily closes switch 272 and samples the voltage across input capacitor 231 onto hold capacitor 236.
  • Summing junction 283 takes the difference between the voltages on hold capacitors 235 and 236. This difference is the voltage change of capacitor 231 when phase-one voltage ramp voltage VR1 is less than control voltage VC1. Accordingly, this time interval is when main MOSFET switch 211 is ON and allows load current to reach inductor 212. As a result, the measurement of voltage change of capacitor 231 during this time is proportional to the current of inductor 212.
  • For converter circuit 220, the sampling of current in inductor 222 for phase-two voltage VR2 occurs as follows. When main MOSFET switch 221 turns ON and allows input voltage VIN from node 291 to reach inductor 222, sampling control circuitry 290 momentarily closes switch 273 and samples the voltage across input capacitor 231 onto hold capacitor 237. Accordingly, when main MOSFET switch 221 turns OFF, sampling control circuitry 290 momentarily closes switch 274 and samples the voltage across input capacitor 231 onto hold capacitor 238.
  • Summing junction 284 takes the difference between the voltages on hold capacitors 237 and 238. This difference is the voltage change of capacitor 231 when phase-two voltage ramp voltage VR2 is less than control voltage VC2. Accordingly, this time interval is when main MOSFET switch 221 is ON and allows load current to reach inductor 222. As a result, the measurement of voltage change of capacitor 231 during this time is proportional to the current of inductor 222.
  • The voltage differences across capacitor 231 when main MOSFET switches 211 and 221 are ON are then utilized, if appropriate, to correct and balance the load distribution between converter circuits 210 and 220. More particularly, these voltage differences, located at the outputs of summing junctions 283 and 284, are utilized by error amplifier 263 so that corrections may performed to evenly distribute load current throughout regulator 200.
  • Although it is advantageous to distribute load current evenly between converter stages, persons skilled in the art will appreciate that load current can be distributed in different proportions between converter stages. For example, suppose that an application exists in which the load current to converter circuit 210 needs to be greater than the load current to converter circuit 220. To accomplish this desired power distribution, the respective control voltages may be controlled by controlling the ON times of the main MOSFET switches.
  • The correction of load current between the converter stages of regulator 200 is accomplished as follows. Current sharing amplifier 263 compares the voltage change of input capacitor 231 during phase-one MOSFET switch 211 ON time with the voltage change of input capacitor 231 during phase-two MOSFET switch 221 ON time. If differences exist in the voltage change of input capacitor 231 between phase-one and phase-two ON time, the output of current sharing amplifier 263 may accordingly adjust the input voltage to error amplifier 262. In this manner, control voltage VC2 to converter 220 may be adjusted. Control voltage VC2 will be adjusted as a result of any differences in capacitor voltages until the phase-one and phase-two voltage changes of capacitor 231 are equal. When this occurs, the two inductor currents are also equal as will be described below. Essentially, the currents in inductor 212 and 222 will be measured and continuously adjusted so that the load current can be balanced between them.
  • For example, suppose the voltage of input capacitor 231 decreases more during phase-one MOSFET switch 211 ON time than during phase-two MOSFET switch 221 ON time. This is reflective of the case when the current in inductor 212 is greater than the current in inductor 222. When this occurs, the output voltage of summing junction 283 will be greater than the output of summing junction 284 because the voltage decrease of input capacitor 231 during the phase-one MOSFET switch 211 ON time is greater than the voltage decrease of input capacitor 231 during the phase-two MOSFET switch 221 ON time. Current sharing amplifier 263 will have a higher voltage at its negative input terminal than the voltage on its positive input terminal and will decrease the input of error control device 262. Control voltage VC2 will then increase, thus increasing the amount of time that main MOSFET switch 221 is ON and the duty cycle of converter 220. As a result, the amount of current in inductor 222 will increase and the difference in voltage of input capacitor 231 between future phase-one MOSFET switch 211 ON and phase-two MOSFET switch 221 ON cycles will decrease.
  • The magnitude of voltage change during phase-two will now be closer, if not equal, to the voltage change during phase-one. If the voltage change of phase-one is still greater than the voltage change of phase-two, the above process will repeat until the two changes are equal. If equal, it signifies an equal amount of current in inductors 212 and 222. In cycles when no difference between the two voltage changes exist, load current will be properly balanced and no adjustment will be needed.
  • As per another example, suppose the voltage decrease during phase-one MOSFET switch 211 ON time is less than the voltage decrease in phase-two MOSFET switch 221 ON time. This example is reflective of the case when the current in inductor 212 is less than the current in inductor 222. As a result, current sharing amplifier 263 will increase the input to error amplifier 262 which will decrease control voltage VC2. Because control voltage VC2 is decreased, the amount of time that main MOSFET switch 221 is ON will decrease as well as the duty cycle of converter 220. As a result, the amount of current in inductor 222 will decrease and the magnitude of future changes in voltage of input capacitor 231 for future phase-two MOSFET switch 221 ON cycles decreases.
  • The magnitude of voltage change during phase-two will now be closer to the voltage change during phase-one. If the voltage change of phase-one is still less than the voltage change of phase-two, the above process will repeat until the two changes are equal. If equal, it signifies an equal amount of current in inductors 212 and 222.
  • Persons skilled in the art will also appreciate that compensation components 282, 244, and 281 are chosen in such a way to help manipulate the effect of amplifier 263, error control device 262 and control voltage VC2. The control voltage VC of converter circuits 210 and 220 is also manipulated by error dividers 289 and 282, error control amplifiers 261 and 262, and compensation components 286 and 281.
  • For at least the above reasons, it may be interpreted that regulator 200 tracks the current in inductor 212 and then corrects the current in inductor 222 to match that of inductor 212. This is accomplished by sampling the voltage of input capacitor 231 and using this information to control the ON time of main MOSFET switch 221 of converter circuit 220. Similarity, persons skilled in the art will appreciate that instead of tracking inductor 212 and manipulating inductor 222, regulator 200 can be configured so that inductor 212 is tracked to inductor 222. This may be accomplished by simply providing the sampled information to converter circuit 210 instead of converter circuit 220. One example may simply involve connecting the output of amplifier 263 to the input of the negative terminal of amplifier 261 and exchanging the positive and negative inputs of amplifier 263.
  • Persons skilled in the art will also appreciate that principles of the present invention may be utilized in switching regulators with more than two converter stages. In such an embodiment, any one or more inductors (e.g., converter stages) of the switching regulator may be designed to track any one or more different inductors (e.g., converter stages). A plurality of converter stages may also be used to track a plurality of different converter stages.
  • Persons skilled in the art will also appreciate that input capacitor 231 may be located elsewhere in regulator 200 and still be sampled for load sharing. For example, capacitor 231 may be an output capacitor similar to capacitor 233. Furthermore, principles of the present invention may be used to sample other elements that could be employed in a switching regulator. Particularly, principles of the present invention could allow for an easily tailored circuit that measures any component whose voltage can distinguish one or more converter stages' inductor currents.
  • FIGS. 3A-3D illustrate the case when the duty cycle of converter stages within a two-converter stage switching regulator is less than 50% such that the main MOSFET switches for the converter stages are never on simultaneously.
  • The below descriptions of FIGS. 3A-3D have reference numerals of the form 3XX and denote the same structural elements throughout. However, electrical events of electrical components that are described in the discussion of FIG. 2 are also illustrated in FIGS. 3A-3D. For example, current I206 of FIG. 3A represents the current flowing through impedance 206 of FIG. 2 with respect to time. In this manner, reference numerals that contain subscript of the form 2XX included in FIGS. 3A-3D map to electrical events of corresponding 2XX electrical components of FIG. 2.
  • FIG. 3A illustrates waveform 310 of current I206 with respect to time. FIG. 3B illustrates waveform 320 of current I211 with respect to time. Persons skilled in the art will appreciate that I211 is positive when MOSFET switch 211 is ON. FIG. 3C illustrates waveform 330 of current I221 with respect to time. Persons skilled in the art will appreciate that I221 is positive when MOSFET switch 221 is ON. FIG. 3D illustrates waveform 340 of the voltage V231 across capacitor 231. At the beginning of the first converter stage's duty cycle, switch 211 is turned ON while switch 221 is OFF. The voltage of capacitor 231 is represented by point 341 of FIG. 3D. After point 341, voltage V231 across capacitor 231 begins to decrease. This decrease in voltage V231 occurs because current I211 through switch 211 and inductor 212 exceeds input current I206 through impedance 206. The amount of decrease in voltage V23, is proportional to the charge removed from capacitor 231. The amount of charge removed from capacitor 231, in turn, is proportional to the current flowing in inductor 212 and the duration of the on-time of switch 211.
  • Point 342 occurs when switch 211 turns OFF and voltage V231 begins to rise as input current I206 recharges input capacitor 231. Point 343 occurs when switch 221 turns ON, the voltage will again decrease while the current I203 through switch 221 and inductor 222 removes charge from capacitor 231. Point 344 occurs when switch 221 turns OFF, the voltage across input capacitor 231 will again rise until switch 211 turns on again at the beginning of the next cycle. If the current in inductor 212 is larger than the current in inductor 222, the voltage decrease across the input capacitor 231 will be larger during the first phase (between points 341 and 342) while switch 211 is ON than during the second phase (between points 343 and 344) when switch 221 is ON. This asymmetry in the voltage waveform is utilized to correct the load sharing between the converters 210 and 220.
  • Similarly, FIGS. 4A-4D illustrate the case when the duty cycle of converter stages within a two-converter stage switching regulator is greater than 50% such that the main MOSFET switches for the converter stages spend a portion of the cycle conducting simultaneously.
  • The below descriptions of FIGS. 4A-4D have reference numerals of the form 4XX and denote the same structural elements throughout. However, electrical events of electrical components that are described in the discussion of FIG. 2 are also illustrated in. FIGS. 4A-4D. For example, current I206 of FIG. 4A represents the current flowing through impedance 206 of FIG. 2 with respect to time. In this manner, reference numerals that contain subscript of the form 2XX included in FIGS. 4A-4D map to electrical events of corresponding 2XX electrical components of FIG. 2.
  • FIG. 4A illustrates waveform 410 of current I206 with respect to time. FIG. 4B illustrates waveform 420 of current I211 with respect to time. Persons skilled in the art will appreciate that I233 is positive when MOSFET switch 211 is ON. FIG. 4C illustrates waveform 430 of current I221 with respect to time. Persons skilled in the art will appreciate that I211 is positive when MOSFET switch 221 is ON. FIG. 4D illustrates waveform 440 of the voltage V232 across capacitor 231.
  • It is further assumed in FIG. 4D that the current in inductor 212 exceeds that in inductor 222. At the beginning of the cycle, point 441, both switches 211 and 221 are on. Because the sum of the currents I211 and I221 exceeds the input current I206 during this time, the voltage V231 across input capacitor 231 will decrease. When switch 221 turns OFF, point 442, the voltage may begin rising or falling, depending on whether input current I206 exceeds the switch current I211. When switch 221 turns on again, point 443, the voltage will again decrease. When switch 211 turns OFF, point 444, the voltage will increase because I206 is greater than the current I221 flowing in switch 221 and inductor 222. The voltage will increase until switch 211 turns ON again, point 445, to initiate the next cycle. Point 446 is attained when switch 221 turns OFF.
  • When the current in inductor 212 is larger than the current in inductor 222, the voltage decrease across the input capacitor 231 will be larger during the phase-one on-time (between points 441 and 444) while switch 211 is on than during the phase-two on-time (between points 443 and 446) when switch 221 is on. This asymmetry in the voltage waveform is used to correct the load sharing between the converters 210 and 220.
  • Persons skilled in the art will appreciate that the present invention is not limited to the embodiments described above that use voltage-mode switching regulators and sample the input capacitor. Instead, the present invention more generally involves implementing a load balancing procedure which utilizes sampling of a component whose voltage is proportional to the current through a regulator circuit's inductor. Moreover, principles of the present invention may be used in a regulator with a single converter stage.
  • Other control techniques may be used for the converter stages such as current mode control and its various embodiments, hysteretic-mode control, and hybrid techniques. Converter stage control can also be based on the sensing of an output capacitor voltage change (e.g., MAX 1710 “Quick-PWM” and Cherry Semiconductor's “V2”). Additionally, the ramp generator that provides timing and control signals to the regulator of the present invention can be comprised of a variety of signals with different frequencies and amplitudes and does not necessarily have to be a generator, but a device used for the same purpose.
  • A wide variety of techniques can also be used to measure the input capacitor waveform and is not limited to a mere sampling circuit. Additional signal processing of the input capacitor waveform may occur, such as filtering or integration, that can be used for ease of implementation or to improve performance such as reducing the effect of switching noise on the signal.
  • This invention can also be used in other switching regulator topologies that use multiple phases such as boost or buck-boost topologies and with or without synchronous switches. For example, if non-synchronous switching is desirable inverter 216 of FIG. 2 and MOSFET switch 213 of FIG. 2 may be replaced with a diode. As described above, the present invention can also be used with output capacitor voltage sensing rather than, or in conjunction with, input capacitor voltage sensing. This is particularly advantageous with boost or buck-boost converters.
  • Persons skilled in the art will recognize that the apparatus of the present invention may be implemented using circuit configurations other than those shown and discussed above. For example, the present invention also may be implemented in single-phase and polyphase switching regulators implementing a variety of power converters such as DC-To-DC, and AC-To-DC. All such modifications are within the scope of the present invention, which is limited only by the claims that follow.

Claims (26)

1-20. (canceled)
21. A system comprising:
a first converter circuit that operates at a first duty cycle;
a second converter circuit that operates at a second duty cycle, the first and second converter circuits having a common output and a common input; and
a capacitor coupled to the common input, wherein a control signal is provided to adjust at least one of the first and second duty cycles based on a change in voltage of the capacitor.
22. The system of claim 21, further comprising an error control device that is coupled to at least one of the first and second converter circuits, wherein an error input signal is provided to the error control device and the error control device provides a correction signal that controls the amount of current supplied by at least one of the first and second converter circuits to the common output.
23. The system of claim 21, further comprising a sampling circuit that samples the change in voltage of the capacitor.
24. The system of claim 21, further comprising a sampling circuit that samples the change in voltage of the capacitor, wherein the sampling circuit includes a control switching circuit.
25. The system of claim 21, wherein a signal generator is coupled to at least one of the first and second converter circuits and the signal generator provides a timing signal to at least one of the first and second converter circuits.
26. The system of claim 21, wherein a signal generator is coupled to at least one of the first and second converter circuits.
27. The system of claim 21, wherein a first terminal of the capacitor is coupled to the common input and a second terminal of the capacitor is coupled to ground.
28. The system of claim 21, wherein the first converter circuit includes a first inductor and the second converter circuit includes a second inductor.
29. The system of claim 21, wherein the first converter circuit includes a first inductor, the second converter circuit includes a second inductor, the control signal controls the amount of current flowing in the first inductor and equalizes the amount of current flowing through the first inductor to the amount of current flowing through the second inductor.
30. The system of claim 21, wherein the first converter circuit includes a first switch and a first inductor, the second converter circuit includes a second switch and a second inductor, the state of the first switch determines the amount of current flowing through the first inductor, and the state of the second switch determines the amount of current flowing through the second inductor.
31. A system comprising:
a first converter circuit that includes a first switch, a first inductor, and operates at a first duty cycle;
a second converter circuit that includes a second switch, a second inductor, and operates at a second duty cycle, the first and second converter circuits having a common output and a common input; and
an energy storage device coupled to the common input, wherein a control signal is provided to adjust at least one of the first and second duty cycles based on a change in voltage of the energy storage device.
32. The system of claim 31, further comprising an error control device that is coupled to at least one of the first and second converter circuits, wherein an error input signal is provided to the error control device and the error control device provides a correction signal that controls the amount of current supplied by at least one of the first and second converter circuits to the common output.
33. The system of claim 31, further comprising a sampling circuit that samples the change in voltage of the energy storage device.
34. The system of claim 31, further comprising a sampling circuit that samples the change in voltage of the energy storage device, wherein the sampling circuit includes a control switching circuit.
35. The system of claim 31, wherein a signal generator is coupled to at least one of the first and second converter circuits and the signal generator provides a timing signal to at least one of the first and second converter circuits.
36. The system of claim 31, wherein a signal generator is coupled to at least one of the first and second converter circuits.
37. The system of claim 31, wherein a first terminal of the energy storage device is coupled to the common input and a second terminal of the energy storage device is coupled to ground.
38. The system of claim 31, wherein the first converter circuit includes a first inductor and the second converter circuit includes a second inductor.
39. The system of claim 31, wherein the first converter circuit includes a first inductor, the second converter circuit includes a second inductor, the control signal controls the amount of current flowing in the first inductor and equalizes the amount of current flowing through the first inductor to the amount of current flowing through the second inductor.
40. The system of claim 41, wherein the first converter circuit includes a first switch and a first inductor, the second converter circuit includes a second switch and a second inductor, the state of the first switch determines the amount of current flowing through the first inductor, and the state of the second switch determines the amount of current flowing through the second inductor.
41. A system comprising:
first converter circuit means for providing a first current to an output, wherein the first converter circuit means operates at a first duty cycle;
second converter circuit means for providing a second current to the output, wherein the second converter circuit means operates at a second duty cycle and the first and second converter circuit means have a common input; and
an energy storage device coupled to the common input, wherein a control signal is provided to adjust at least one of the first and second duty cycles based on a change in voltage of the energy storage device.
42. The system of claim 41, further comprising sampling circuit means for sampling the change in voltage of the energy storage device.
43. The system of claim 41, wherein a signal generator is coupled to at least one of the first and second converter circuit means and the signal generator provides a timing signal to at least one of the first and second converter circuit means.
44. The system of claim 41, wherein a signal generator is coupled to at least one of the first and second converter circuit means.
45. The system of claim 41, wherein a first terminal of the energy storage device is coupled to the common input and a second terminal of the energy storage device is coupled to ground.
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