US20060132112A1 - High efficiency, high slew rate switching regulator/amplifier - Google Patents
High efficiency, high slew rate switching regulator/amplifier Download PDFInfo
- Publication number
- US20060132112A1 US20060132112A1 US11/281,878 US28187805A US2006132112A1 US 20060132112 A1 US20060132112 A1 US 20060132112A1 US 28187805 A US28187805 A US 28187805A US 2006132112 A1 US2006132112 A1 US 2006132112A1
- Authority
- US
- United States
- Prior art keywords
- linear amplifier
- voltage
- switch
- regulating apparatus
- coupled
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Granted
Links
Images
Classifications
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
- H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
- H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
- H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
- H02M3/158—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load
- H02M3/1584—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load with a plurality of power processing stages connected in parallel
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/0045—Converters combining the concepts of switch-mode regulation and linear regulation, e.g. linear pre-regulator to switching converter, linear and switching converter in parallel, same converter or same transistor operating either in linear or switching mode
Definitions
- the second switching regulator 18 comprises a first comparator 26 having a first input and a second input which are coupled across resistor R 12 , and a second comparator 28 having a first input and a second input which are coupled across resistor R 11 .
- the second switching regulator 18 further includes a first switch 27 , which is a pMOS device, a second switch 29 , which is an nMOS device, an inductor 31 and an active diode 32 .
- the output of the first comparator 26 is coupled to the first switch 27 , which has a source terminal coupled to the supply voltage, V SUPPLY .
- the output of the second comparator 28 is coupled to the gate of the second switch 29 .
Landscapes
- Engineering & Computer Science (AREA)
- Power Engineering (AREA)
- Dc-Dc Converters (AREA)
- Continuous-Control Power Sources That Use Transistors (AREA)
- Amplifiers (AREA)
Abstract
Description
- This patent application, and any patent(s) issuing therefrom, claims priority to U.S. provisional patent application No. 60/628,652, filed on Nov. 18, 2004, which is incorporated herein by reference in its entirety.
- The present invention relates to an improved switching regulator and/or amplifier, and more specifically, to a novel, cost effective design for a switching regulator and/or amplifier that provides for both high efficiency and high slew rate.
- It has been known in the prior art to utilize switching regulators/amplifiers in applications such as, but not limited to: (1) voltage regulators utilized for supplying a relatively fixed DC voltage to a load whose current demands change very quickly such as CMOS logic processors whose activity can go from negligible (such as in standby) to very high or vise-versa in a few nanoseconds, for example, at the change of state of a control signal; and (2) “digital” amplifiers or programmable regulators where the load is relatively fixed but its voltage is changed very rapidly in response to an external command, such as DSL line drivers and supplies or modulators for communication transmitters where the power level or information signal level is changed often and abruptly over a wide dynamic range.
- It is also noted that the foregoing applications are characterized by a step down operation where the supply voltage is relatively fixed or slowly varying (such as, for example, a battery), and the widely varying load current is sourced at a voltage that is either fixed or varying, but at a lower value than the supply voltage.
- Prior art designs for switching regulators/amplifier to be utilized in the foregoing applications have generally included buck topology switching regulators having low value inductors, high switching frequencies and hysteretic control algorithms without loop filters to achieve high load current slew rates. As is known:
However, the use of such low value inductors results in large values of ripple current and conduction losses, while high switching frequencies result in larger switching losses, both of which undesirably lower efficiency. - In an effort to satisfy performance requirements, it has been known in the prior art to add a cascaded linear amplifier/low drop out regulator immediately before the load, even though the losses due to the load current at the required voltage overhead of the linear stage can be large. Such prior art systems are described, for example, in U.S. Pat. Nos. 4,378,530 and 5,905,407.
FIG. 1 illustrates an exemplary block diagram of such a device. - Referring to
FIG. 1 , the device includes aprogrammable switching regulator 12 cascaded with alinear amplifier stage 14. In addition, the device includes overheadvoltage reference supply 16, and resistors R1 and R2, which are coupled in series to one another and to the output node, VO. The overheadvoltage reference supply 16 causes VR=VO+VB1, which is necessary for the linear amplifier to operate, as VR must be larger than VO by an “overhead voltage”. Resistors R1 and R2 form a voltage divider circuit, and provide a feedback signal to thelinear amplifier stage 14. The output of thelinear amplifier stage 14 operating in conjunction with the output of theprogrammable switching regulator 12 generate the output voltage, VO, of the device, which is coupled to the load (e.g., a power amplifier in a cell phone application). VSUPPLY corresponds to the voltage source for the device (e.g., a battery in a cell phone application), and VREF sets the output voltage needed to supply the power level required by the load. It is noted that in some applications, VREF will represent the instantaneous power requirement of the load and will include content data (e.g., voice or data information to be transmitted) which is superimposed on the VREF signal utilizing any suitable modulation technique. In operation, thelinear amplifier stage 14 essentially functions as the power supply regulator operative to generate a substantially clean signal, VO, which is representative of the instantaneous power required for the task currently at hand. - However, if the output voltage of the switching regulator cannot change rapidly enough to follow voltage changes in VREF, then VR must be set to the instantaneous peak value of VO plus enough additional voltage margin B so that the linear amplifier does not “clip” on signal peaks. If the supply voltage, VSUPPLY, is significantly greater than VR, use of the switching regulator saves most of the power equal to ILOAD*(VSUPPLY−VR), which would otherwise be dissipated in the linear amplifier.
- While these known prior art devices provide for an improvement in efficiency, for example, by allowing for a reduction in the switching frequency of the switching regulator, due to the requirements of today's applications and the continued demand for reducing power requirements so as to extend battery life, a further increase in the overall operating efficiency of switching regulators/amplifiers is necessary. It is an object of the present invention to satisfy these needs.
- In view of the foregoing, it is a primary objective of the present invention to provide a novel switching regulator/amplifier which exhibits improved efficiency and slew rate performance relative to known prior art devices. It is also an objective of the present invention to provide a cost effective design for the novel switching regulator/amplifier so that the device represents a practical solution to the aforementioned problems.
- Specifically, the present invention relates to a regulating apparatus having an output node and being operative for regulating the voltage level at the output node in response to a reference signal provided as an input to the regulating apparatus. The regulating apparatus includes a linear amplifier stage operative for receiving the reference signal and being capable of sourcing current to the output node when the reference signal indicates that the present voltage level at the output node is less than a desired voltage level at the output node. The regulating apparatus further includes a switching regulator, which is controlled by the linear amplifier stage, and which is operative for sourcing current to the output node when the amount of current being sourced to the output node by the linear amplifier stage exceeds a predetermined threshold.
- The switching regulator/amplifier of the present invention provides numerous advantages over the prior art. One advantage of the present invention is that it provides a highly efficient switching regulator/amplifier that minimizes the power requirements for operation. This is accomplished in-part by reducing the power dissipated by the linear amplifier contained in the device, by providing a separate current path that is capable of providing the steady state current requirements to the load (i.e., the linear amplifier is activated only during fast changing transient voltage swings in the load). As a result, as one example, the present invention advantageously allows for an extension of battery operation time of a cell phone between charges.
- In addition, the switching regulator/amplifier provides for an increased slew rate capability. As the result of the design of the present invention, which incorporates the use of a “free-wheeling” switch, it is possible to rapidly reduce the load current to substantially zero (i.e., on the order of a few nanoseconds). Moreover, when the load current is reduced in the foregoing manner, the design of the present invention does not immediately dissipate the current (i.e., as explained below the current is temporarily stored), and therefore if the load must be increased shortly after the reduction, the stored current is again coupled/provided to the load. The foregoing operation allows the switching regulator/amplifier of the present invention to exhibit both a high slew rate capability and increased operating efficiency.
- Yet another advantage of the present invention is that the design provides a “feed-forward” control system in which the switching regulator/amplifier reacts to changes in the desired voltage set point when adjusting the current delivered to the load. The control of the switching regulator/amplifier does not utilize the output voltage signal. As a result, the design of the present invention further improves both slew rate performance (as the load current is adjusted more rapidly in comparison to a device that modifies the current delivered to the load based on changes in the output voltage of the regulator) and efficiency performance (as there is no sense resistor coupled to the output of device, which would result in an increase in power dissipation).
- Additional objects, advantages, and novel features of the invention will become apparent to those skilled in the art upon examination of the following description, or may be learned by practice of the invention. While the novel features of the invention are set forth below, the invention, both as to organization and content, will be better understood and appreciated, along with other objects and features thereof, from the following detailed description taken in conjunction with the drawings.
- The accompanying drawings, which are incorporated into and form a part of the specification, illustrate several aspects and embodiments of the present invention and, together with the general description given above and detailed description given below, serve to explain the principles of the invention. Such description makes reference to the annexed drawings. The drawings are only for the purpose of illustrating preferred embodiments of the invention and are not to be treated as limiting the invention.
- In the drawings:
-
FIG. 1 illustrates a block diagram of a prior art implementation of a switching regulator/amplifier that utilizes a linear amplifier in the design. -
FIG. 2 illustrates an exemplary block diagram of a switching regulator/amplifier in accordance with the present invention. -
FIG. 3 illustrates a schematic diagram of an exemplary implementation of the switching regulator/amplifier of the present invention. -
FIG. 4 illustrates a first alternative embodiment of the output stage of the linear amplifier stage. -
FIG. 5 illustrates a second alternative embodiment of the output stage of the linear amplifier stage. - Throughout the above-mentioned drawings, identical reference numerals are used to designate the same or similar component parts.
- The present invention now will be described more fully hereinafter with reference to the accompanying drawings, in which preferred embodiments of the invention are shown. This invention may, however, be embodied in many different forms and should not be construed as limited to the embodiments set forth herein: rather, these embodiments are provided so that this disclosure will be thorough and complete, and will fully convey the scope of the invention to those skilled in the art, like numbers refer to like elements throughout.
- Referring to
FIG. 2 , similar to the prior art design illustrated inFIG. 1 , the switch regulator/amplifier of the present invention comprises aprogrammable switching regulator 12 cascaded with alinear amplifier stage 14, as well anoverhead voltage supply 16 and resistors R1 and R2. Thelinear amplifier stage 14 receives VREF as an input signal. The foregoing components are coupled together in the same manner as illustrated inFIG. 1 . However, the design also includes asecond switching regulator 18 coupled between the supply voltage VSUPPLY and the output, VO, as shown inFIG. 2 . VR is the minimum supply voltage for the linear amplifier that allows it to follow the signal peaks of VREF without clipping. Thelinear amplifier stage 14 providescontrol signals 17 to thesecond switching regulator 18, which govern the operation of thesecond switch regulator 18. - As explained in more detail below, the inclusion of the
second switching regulator 18 disposed between the power supply, VSUPPLY, and the load, VO, provides for a second current path so as to allow for the steady state current (or slowly changing current) required by the load to be delivered to the load via thesecond switching regulator 18 without accessing/utilizing thelinear amplifier stage 14. In the present invention, thelinear amplifier stage 14 is primarily utilized to deliver fast changing (i.e., transient) current requirements to the load. As a result of this design, the utilization of thelinear amplifier stage 14, which exhibits low efficiency and high power dissipation, is minimized thereby increasing the overall efficiency of the device. -
FIG. 3 illustrates a schematic diagram of an exemplary implementation of the switching regulator/amplifier illustrated inFIG. 2 . It is noted that the present invention is not limited to the specific embodiment disclosed inFIG. 3 , as variations to the particular design are clearly possible. - Referring to
FIG. 3 , theprogrammable switching regulator 12 receives VSUPPLY as an input voltage and generates an output voltage signal VR. In addition, overheadvoltage reference supply 16 is coupled to theprogrammable switching regulator 12. It is noted that theprogrammable switching regulator 12 furnishes an output voltage, VR, which is equal to the average value of the linear amplifier output voltage, VO, plus an additional voltage VB1, where VB1 equals the peak to average value of VO and a small additional voltage necessary to assure thatlinear amplifier 14 does not clip (i.e., voltage saturate) on peaks of the VREF signal. As such, theprogrammable switching regulator 12 need only have a response time fast enough to follow the average value of VO and not its instantaneous value or envelope. - More specifically, the output voltage, VR, of the programmable switching regulator follows the value of its input reference voltage, (VO+VB), within the capability of its control bandwidth or response time as set forth by its internal clock or switching frequency. Thus, the voltage VR follows the average value of VO plus the additional voltage of VB1, where the averaging period is set by the control bandwidth of the programmable switching regulator or may be adjusted to a specific value by adding an additional low pass filter in its control input line. The choice of averaging period and value of VR are selected to match the characteristics of the VREF signal and linear amplifier response such that the value of VB1 is equal to the maximum value of VO average to positive peak value during any sliding time averaging period as a window. The objective is to minimize the value of VB1 to the smallest value of the average to peak during the response time of the switching regulator so that most of the voltage difference between VSUPPLY and VO can be absorbed by the switching regulator at typically 90% efficiency rather than be wasted as voltage drop across the linear amplifier. Thus, by choosing an appropriate VB1 to match the AC signal characteristics of VO (and therefore VREF when not distorting) and the response time of the switching regulator, essentially any programmable switching regulator response time and signal characteristic of VO can be accommodated. However, it is noted that if the switching regulator response is too slow relative to the rate of change of the VREF signal, efficiency improvements from use of the programmable switching regulator may be small and overall system efficiency inadequate.
- Continuing, in the given embodiment the
linear amplifier stage 14 includes anerror amplifier 22, alinear amplifier 24 comprising an NPN transistor, resistor R11 coupled between the base and emitter terminals of thelinear amplifier 24, resistor R12 coupled to the collector of thelinear amplifier 24, and capacitor Cc and resistor Rc connected in series and coupled to the output of theerror amplifier 22. The emitter of thelinear amplifier 24 is coupled to the load, VO. As shown inFIG. 3 , the output signal, VR, of theprogrammable switching regulator 12 is coupled to both resistor R12 and theerror amplifier 22 and functions as the amplifier supply voltage. In operation, theerror amplifier 22 and thelinear amplifier 24 form a linear amplifier/regulator that has sufficient bandwidth to allow the output, VO, to follow the reference VREF in the presence of rapid time variations in VREF and/or the load current. As shown, a portion of the output signal, VO, is fed-back to the input of theerror amplifier 22 so as to allow the error amplifier to generate an output signal indicative of the difference between the desired output voltage level and the actual output voltage level, and cause VO to follow (VREF*(R1+R2)/R2). - Referring again to
FIG. 3 , in the given embodiment, thesecond switching regulator 18 comprises afirst comparator 26 having a first input and a second input which are coupled across resistor R12, and asecond comparator 28 having a first input and a second input which are coupled across resistor R11. Thesecond switching regulator 18 further includes afirst switch 27, which is a pMOS device, asecond switch 29, which is an nMOS device, aninductor 31 and anactive diode 32. As shown, the output of thefirst comparator 26 is coupled to thefirst switch 27, which has a source terminal coupled to the supply voltage, VSUPPLY. The output of thesecond comparator 28 is coupled to the gate of thesecond switch 29. It is noted that theinductor 31 is coupled between the source and drain terminals of thesecond switch 29, and the body of thesecond switch 29 is not connected to either its source or drain, but rather to ground as shown inFIG. 3 . Theinductor 31 is also coupled between the drain terminal of thefirst switch 27 and the load, VO. It is further noted that the drain terminal of thefirst switch 27 and the source terminal ofsecond switch 29 are coupled together and are also coupled todiode 32. In the preferred embodiment, thediode 32 is an “active” type diode comprising a comparator and NMOS transistor as described in pending application Ser. No. 11/094,369 filed Mar. 31, 2005, which is hereby incorporated by reference in its entirety. - Turning to the operation of the device as a system, it is noted that without the
second switching regulator 18 in the device, the entire load current would have to pass through thelinear amplifier stage 14, and as a result the power dissipation due to the load current times the overhead voltage B1 required for proper operation would greatly reduce the efficiency of the device. However, by including thesecond switching regulator 18, which has minimal switching and conduction losses, most of the load current passes through thesecond switching regulator 18 and therefore bypasses thelinear amplifier stage 14, thereby greatly improving overall efficiency. It is noted, however, that the linear path is always present and can supply the entire incremental load current during transients. - At initial turn of the power supply, VSUPPLY, with VREF already having a desired value and VB1 set appropriately as described earlier, VO is zero and the input to the programmable switching regulator is VB1. The programmable switching regulator output voltage VR rises toward (VO+VB1) at a rate set by its inherent response time, and the linear regulator now has a non-zero supply voltage, and so long as VO<(VREF*(R1+R2)/R2), it continues to increase VO toward VR, and therefore the output voltage VO thus ramps up at a rate set by the slew rate of the programmable switching regulator. When VO=(VREF*(R1+R2)/R2), VO has reached steady state and its stays at that voltage until the programmable switching regulator output, VR, has reached [(VREF*(R1+R2)/R2)+VB1), at which point it remains static unless or until VREF changes. Thus, power on requires no special function within the device design, and the operation of the second switching regulator will be the same as described in the following for all modes of operation including start up.
- Continuing, during operation, if the
second switching regulator 18 off, the load current flows through thelinear amplifier stage 14 including thelinear amplifier 24. This results in an increase in the voltage drop across R12, which if greater than the upper threshold of thefirst comparator 26, results in the turn on offirst switch 27 and therefore the supply of current to the load, VO, through theinductor 31. As the inductor current increases, the current in thelinear amplifier 24 decreases because their sum is the present load current. This reduces the voltage drop across R12 until such time that the reduction in voltage across R12 causes it to become less than the lower threshold of thefirst comparator 26 and turns off thefirst switch 27, thereby preventing further current from being supplied to the load from VSUPPLY. Thus, at steady state, thecomparator 26 switches on and off at some duty cycle, and most of the load current flows through theinductor 31, and consists of a DC component and an AC triangular component. The sum of the DC component and AC component of the inductor current and the linear amplifier current equals the load current. Thus, the linear amplifier AC current is 180 degrees out of phase with the AC component of the inductor current and there is no AC voltage ripple present at the load, VO. - The switching frequency of the
second switching regulator 18 is set by the relationship between VSUPPLY-VO, the value of theinductor 31, the value of hysteresis set by thefirst comparator 26 and the voltage drop from the current through resistor R12. It is noted that when thefirst switch 27 is off, inductor current flows through thediode 32, which as noted above is preferably of the “active” type, and therefore has a forward voltage drop that is negligible with respect to VO. The actual values utilized for the various components are typically based on the specific application for which the device will be utilized n conjunction with well known design relationships. - From the foregoing discussion, it is clear that the circuit of the present invention is capable of handling steady state and increasing load current exceedingly well. However, the circuit is also capable of handling rapidly decreasing load currents, and does so in a manner which provides for both high slew rates and improved efficiency. In operation, during transients when the inductor current is larger than the load current, the
linear amplifier stage 14 starts to turn off when the voltage across R11 becomes less than VBE oflinear amplifier 24, thereby turning offlinear amplifier 24. The value of R11 is part of the amplifier design and the threshold ofcomparator 28 should be about 0.8*VBE with a few millivolts of hysteresis to avoid noise effects. At this time, thesecond comparator 28 turns on thesecond switch 29, which allows the inductor current to recirculate and slowly decay in value without being passed into the load, VO. Specifically, the inductor current recirculates in an autonomous loop formed by theinductor 31 and the second switch 29 (which is referred to herein as a free-wheeling switch). Thus, the foregoing configuration allows the load current to be rapidly reduced to substantially zero on the order of a few nanoseconds. In other words, the device allows the total current sourced by the overall regulator/amplifier to go to nearly zero during transients even though thelinear amplifier stage 14 can only source current, and prevents voltage overshoots in most any dissipative load without degrading efficiency. Further, as VO is not used to control thesecond switching regulator 18, it has no ripple voltage and can precisely track VREF. - It is further noted that by utilizing the “free-wheeling”
switch 29 in the device of the present invention, when the load current is reduced in the foregoing manner, the device of the present invention does not immediately dissipate the current (i.e., the current is temporarily stored in the inductor and autonomous loop). As such, if the load current must be increased shortly after the reduction, the stored current is again coupled to the load. This would occur upon deactivation of thesecond switch 29, which occurs when thelinear amplifier stage 14 becomes active again (i.e., VREF indicates a desired increase in load voltage) and the voltage across R11 is greater than the trip point of thesecond comparator 28. This operation of not dissipating the inductor current and allowing for the reuse of the stored inductor current allows the switching regulator/amplifier of the present invention to exhibit high slew rates and increased efficiency. - It is also noted that by sensing the collector current of
linear amplifier 24 instead of the output current of thelinear amplifier stage 14, the output impedance of thelinear amplifier stage 14 is advantageously not increased by a sensing resistor. Furthermore, the value of R11 can be relatively large so that a small current threshold of thesecond comparator 28 can be achieved with minimal error due to the voltage offset of thesecond comparator 28. - While an exemplary embodiment of the present invention is set forth above in
FIG. 3 is it noted that the present invention is not intended to be limited to the disclosed embodiments as various implementations of the device are clearly possible. For example,FIGS. 4 and 5 illustrate alternative embodiments of the output stage of thelinear amplifier 24. - More specifically, in a first variation, the
linear amplifier 24 can comprise two matchedparallel transistors FIG. 4 , where the emitter area of 24 A, is K* (area of 24 B) and R2 is K*R2. Thus, with K large, R2 can be sized more conveniently but still maintain the threshold of thefirst comparator 26 the same with respect to the total collector current oflinear amplifier 24 ofFIG. 3 , and the total current gain of thelinear amplifier 24 will not change appreciably even iftransistor 24 B voltage saturates. In a second variation, an additional linear amplifier stage consisting oftransistor 25 andmirror 26 also could be added tolinear amplifier 24, as shown inFIG. 5 , to further lower the output impedance of the linear amplifier and make its frequency compensation easier without changing the operating voltages from those of the configuration shown inFIG. 3 or 4. It is noted that if utilizing the alternative embodiments for thelinear amplifier 24, in addition to the linear amplifier, the resistor R12 would be replaced by the circuit shown inFIGS. 4 and 5 , and the inputs of the first comparator would be coupled across R12′. InFIG. 5 , the inputs of the second comparator would be coupled across resistor R11′. - As noted above, the switching regulator/amplifier of the present invention provides numerous advantages over the prior art. One advantage of the present invention is that it provides a highly efficient switching regulator/amplifier that minimizes the power requirements for operation. This is accomplished in-part by reducing the power dissipated by the linear amplifier contained in the device, by providing a separate current path that is capable of providing the steady state current requirements to the load (i.e., the linear amplifier is activated only during fast changing transient voltage swings in the load). As a result, as one example, the present invention advantageously allows for an extension of battery operation time of a cell phone between charges.
- Another advantage is that the switching regulator/amplifier of the present invention provides for an increased slew rate capability. As the result of the present invention, which incorporates the use of a “free-wheeling” switch, it is possible to rapidly reduce the load current to substantially zero (i.e., on the order of a few nanoseconds). Moreover, when the load current is reduced in the foregoing manner, the design of the present invention does not immediately dissipate the current (i.e., as explained above the current is temporarily stored), and therefore if the load must be increased shortly after the reduction, the stored current is again coupled to the load. The foregoing operation allows the switching regulator/amplifier of the present invention to exhibit high slew rate capabilities and improved efficiency.
- Yet another advantage of the present invention is that the design provides a “feed-forward” control system in which the switching regulator/amplifier reacts to changes in the desired voltage set point when adjusting the current delivered to the load. The control of the switching regulator/amplifier does not utilize the output voltage signal. As a result, the design of the present invention further improves both slew rate performance (as the load current is adjusted more rapidly in comparison to a device that modifies the current delivered to the load based on changes in the output voltage of the regulator) and efficiency performance (as there is no sense resistor coupled to the output of device, which would result in an increase in power dissipation).
- While specific embodiments of, and examples for, the invention are described herein for illustrative purposes, various equivalent modifications are possible within the scope of the invention, as those skilled in the relevant art will recognize.
- For example, it is noted that the
programmable switching regulator 12 operates to maintain VR−VO greater than the linear stage drop out voltage even if the short term voltage slew of VO exceeds VR, by choosing voltage offset B1 appropriately. This is necessary to maintain efficiency if the VSUPPLY−VO voltage differential is much larger than the dropout voltage of the linear regulator. In the event that the VSUPPLY−VO voltage differential is not larger than the dropout voltage of the linear regulator, it is possible to omit the programmable switching regulator from the design. - The aforementioned variations are merely examples. Further, the terms used in the following claims should not be construed to limit the invention to the specific embodiments disclosed in the specification and the claims. Rather, the scope of the invention is to be determined entirely by the following claims, which are to be construed in accordance with established doctrines of claim interpretation.
Claims (19)
Priority Applications (1)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
US11/281,878 US7292015B2 (en) | 2004-11-18 | 2005-11-18 | High efficiency, high slew rate switching regulator/amplifier |
Applications Claiming Priority (2)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
US62865204P | 2004-11-18 | 2004-11-18 | |
US11/281,878 US7292015B2 (en) | 2004-11-18 | 2005-11-18 | High efficiency, high slew rate switching regulator/amplifier |
Publications (2)
Publication Number | Publication Date |
---|---|
US20060132112A1 true US20060132112A1 (en) | 2006-06-22 |
US7292015B2 US7292015B2 (en) | 2007-11-06 |
Family
ID=36635781
Family Applications (1)
Application Number | Title | Priority Date | Filing Date |
---|---|---|---|
US11/281,878 Expired - Fee Related US7292015B2 (en) | 2004-11-18 | 2005-11-18 | High efficiency, high slew rate switching regulator/amplifier |
Country Status (2)
Country | Link |
---|---|
US (1) | US7292015B2 (en) |
JP (1) | JP4832056B2 (en) |
Cited By (9)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US7271660B1 (en) * | 2005-05-11 | 2007-09-18 | National Semiconductor Corporation | Selectively adding auxiliary frequency compensation depending on the behaviour of an output transistor of a rail-to-rail operational amplifier |
US20130342185A1 (en) * | 2011-03-03 | 2013-12-26 | Nec Corporation | Power supplying apparatus and control method thereof |
US20140042999A1 (en) * | 2012-08-10 | 2014-02-13 | Texas Instruments Incorporated | Switched mode assisted linear regulator with ac coupling with capacitive charge control |
US8847564B2 (en) * | 2007-07-06 | 2014-09-30 | Advanced Analogic Technologies, Incorporated | DC/DC converter using synchronous freewheeling MOSFET |
US20150155783A1 (en) * | 2012-08-10 | 2015-06-04 | Texas Instruments Incorporated | Switched mode assisted linear regulator with dynamic buck turn-off using zcd-controlled tub switching |
US20150364991A1 (en) * | 2014-06-16 | 2015-12-17 | City University Of Hong Kong | Input filter for a power electronic system |
US9250694B1 (en) * | 2013-05-10 | 2016-02-02 | Sridhar Kotikalapoodi | Method and apparatus for fast, efficient, low noise power supply |
US9473023B2 (en) | 2012-08-10 | 2016-10-18 | Texas Instruments Incorporated | Switched mode assisted linear regulator with seamless transition between power tracking configurations |
US11194356B2 (en) * | 2019-06-28 | 2021-12-07 | Analog Devices International Unlimited Company | Linear stage efficiency techniques for H-bridge systems |
Families Citing this family (14)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US7421593B2 (en) * | 2004-11-19 | 2008-09-02 | Intel Corporation | Parallel-connected voltage regulators for supplying power to integrated circuit so that second regulator minimizes current output from first regulator |
US8035362B2 (en) * | 2005-04-20 | 2011-10-11 | Nxp B.V. | Amplifier system with DC-component control |
JP4751105B2 (en) * | 2005-05-26 | 2011-08-17 | ローム株式会社 | Power supply device control circuit, power supply device using the same, and electronic equipment |
US7957847B2 (en) * | 2005-09-30 | 2011-06-07 | Hitachi Global Storage Technologies Netherlands, B.V. | Voltage regulating systems responsive to feed-forward information from deterministic loads |
US7521907B2 (en) * | 2006-03-06 | 2009-04-21 | Enpirion, Inc. | Controller for a power converter and method of operating the same |
TWI310124B (en) * | 2006-04-24 | 2009-05-21 | Ind Tech Res Inst | Power supply apparatus |
US7705574B2 (en) * | 2008-06-13 | 2010-04-27 | Hamilton Sundstrand Corporation | Remote power controller with power sharing circuit |
US8587268B1 (en) * | 2008-06-18 | 2013-11-19 | National Semiconductor Corporation | System and method for providing an active current assist with analog bypass for a switcher circuit |
US8248044B2 (en) * | 2010-03-24 | 2012-08-21 | R2 Semiconductor, Inc. | Voltage regulator bypass resistance control |
US8917067B2 (en) | 2010-03-24 | 2014-12-23 | R2 Semiconductor, Inc. | Assisting an output current of a voltage converter |
US8797772B2 (en) | 2011-06-30 | 2014-08-05 | Texas Instruments Incorporated | Low noise voltage regulator |
US8994347B2 (en) | 2012-06-04 | 2015-03-31 | R2 Semiconductor, Inc. | Assisting a load current of a switching voltage regulator |
US8847688B1 (en) | 2012-08-03 | 2014-09-30 | Google Inc. | Over-voltage protection in a high-swing amplifier |
US9748845B1 (en) * | 2013-11-02 | 2017-08-29 | Sridhar Kotikalapoodi | Method and apparatus for wide bandwidth, efficient power supply |
Citations (45)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US4378530A (en) * | 1979-07-04 | 1983-03-29 | Unisearch Limited | High-efficiency low-distortion amplifier |
US4502152A (en) * | 1978-08-16 | 1985-02-26 | Lucas Industries Limited | Low current linear/high current chopper voltage regulator |
US4727308A (en) * | 1986-08-28 | 1988-02-23 | International Business Machines Corporation | FET power converter with reduced switching loss |
US4943902A (en) * | 1987-11-23 | 1990-07-24 | Viteq Corporation | AC to DC power converter and method with integrated line current control for improving power factor |
US4959606A (en) * | 1989-01-06 | 1990-09-25 | Uniphase Corporation | Current mode switching regulator with programmed offtime |
US5083078A (en) * | 1990-05-12 | 1992-01-21 | Daimler-Benz Ag | Device for supplying power to an electronic computer in a motor vehicle |
US5258701A (en) * | 1992-09-02 | 1993-11-02 | The United States Of America As Represented By The Secretary Of The Army | DC power supply |
US5305192A (en) * | 1991-11-01 | 1994-04-19 | Linear Technology Corporation | Switching regulator circuit using magnetic flux-sensing |
US5414341A (en) * | 1993-12-07 | 1995-05-09 | Benchmarq Microelectronics, Inc. | DC-DC converter operable in an asyncronous or syncronous or linear mode |
US5479090A (en) * | 1993-11-24 | 1995-12-26 | Raytheon Company | Power converter having optimal dynamic operation |
US5600234A (en) * | 1995-03-01 | 1997-02-04 | Texas Instruments Incorporated | Switch mode power converter and method |
US5903447A (en) * | 1997-07-23 | 1999-05-11 | Murata Manufacturing Co., Ltd. | Current-mode control device and switching power supply employing same |
US5905407A (en) * | 1997-07-30 | 1999-05-18 | Motorola, Inc. | High efficiency power amplifier using combined linear and switching techniques with novel feedback system |
US5929620A (en) * | 1996-11-07 | 1999-07-27 | Linear Technology Corporation | Switching regulators having a synchronizable oscillator frequency with constant ramp amplitude |
US5949229A (en) * | 1996-08-28 | 1999-09-07 | Samsung Electronics, Co., Ltd. | Power factor correction circuit having an error signal multiplied by a current signal |
US5982160A (en) * | 1998-12-24 | 1999-11-09 | Harris Corporation | DC-to-DC converter with inductor current sensing and related methods |
US6034517A (en) * | 1998-10-27 | 2000-03-07 | Linear Technology Corporation | High efficiency step-down switching regulators |
US6046516A (en) * | 1994-10-28 | 2000-04-04 | Siemens Aktiengesellschaft | Electronic switch for use with inductive loads |
US6066943A (en) * | 1998-10-08 | 2000-05-23 | Texas Instruments Incorporated | Capacitive-summing switch-mode power conversion control |
US6166528A (en) * | 1999-11-02 | 2000-12-26 | Fairchild Semiconductor Corporation | Lossless current sensing in buck converters working with low duty cycles and high clock frequencies |
US6222356B1 (en) * | 1998-04-01 | 2001-04-24 | Siemens Aktiengesellschaft | Current mode switching regulator configured such that a measuring resistor is not needed to measure the current at an inductor |
US6229289B1 (en) * | 2000-02-25 | 2001-05-08 | Cadence Design Systems, Inc. | Power converter mode transitioning method and apparatus |
US6249110B1 (en) * | 1999-04-16 | 2001-06-19 | Robert Bosch Gmbh | Circuit configuration for generating a stabilized power supply voltage |
US6268756B1 (en) * | 1999-03-31 | 2001-07-31 | Sony Corporation | Fast high side switch for hard disk drive preamplifiers |
US6307356B1 (en) * | 1998-06-18 | 2001-10-23 | Linear Technology Corporation | Voltage mode feedback burst mode circuit |
US6313610B1 (en) * | 1999-08-20 | 2001-11-06 | Texas Instruments Incorporated | Battery protection circuit employing active regulation of charge and discharge devices |
US6366070B1 (en) * | 2001-07-12 | 2002-04-02 | Analog Devices, Inc. | Switching voltage regulator with dual modulation control scheme |
US6404261B1 (en) * | 1999-03-27 | 2002-06-11 | Koninklijke Philips Electronics N.V. | Switch circuit and semiconductor switch, for battery-powered equipment |
US6476589B2 (en) * | 2001-04-06 | 2002-11-05 | Linear Technology Corporation | Circuits and methods for synchronizing non-constant frequency switching regulators with a phase locked loop |
US6486643B2 (en) * | 2000-11-30 | 2002-11-26 | Analog Technologies, Inc. | High-efficiency H-bridge circuit using switched and linear stages |
US6498466B1 (en) * | 2000-05-23 | 2002-12-24 | Linear Technology Corp. | Cancellation of slope compensation effect on current limit |
US6509721B1 (en) * | 2001-08-27 | 2003-01-21 | Koninklijke Philips Electronics N.V. | Buck regulator with ability to handle rapid reduction of load current |
US6522178B2 (en) * | 1999-04-22 | 2003-02-18 | International Rectifier Corporation | Controlling high side devices without using level shift switches |
US6541947B1 (en) * | 1998-09-10 | 2003-04-01 | Robert Bosch Gmbh | Step-down constant-current transformer |
US6636023B1 (en) * | 1999-10-14 | 2003-10-21 | Juniper Networks, Inc. | Combined linear and switching voltage regulator |
US6661211B1 (en) * | 2002-06-25 | 2003-12-09 | Alcatel Canada Inc. | Quick-start DC-DC converter circuit and method |
US6661210B2 (en) * | 2002-01-23 | 2003-12-09 | Telfonaktiebolaget L.M. Ericsson | Apparatus and method for DC-to-DC power conversion |
US6724174B1 (en) * | 2002-09-12 | 2004-04-20 | Linear Technology Corp. | Adjustable minimum peak inductor current level for burst mode in current-mode DC-DC regulators |
US6744241B2 (en) * | 2002-06-07 | 2004-06-01 | Infineon Technologies Ag | Method for driving a switch in a switch-mode converter, and a drive circuit for driving a switch |
US6828766B2 (en) * | 2002-05-30 | 2004-12-07 | Stmicroelectronics S.R.L. | Voltage regulator |
US6873140B2 (en) * | 2002-07-12 | 2005-03-29 | Stmicroelectronics S.R.L. | Digital contoller for DC-DC switching converters |
US6894471B2 (en) * | 2002-05-31 | 2005-05-17 | St Microelectronics S.R.L. | Method of regulating the supply voltage of a load and related voltage regulator |
US7030596B1 (en) * | 2003-12-03 | 2006-04-18 | Linear Technology Corporation | Methods and circuits for programmable automatic burst mode control using average output current |
US7129681B2 (en) * | 2002-08-23 | 2006-10-31 | Ricoh Company, Ltd. | Power supply apparatus having parallel connected switching and series regulators and method of operation |
US7148665B2 (en) * | 2002-07-25 | 2006-12-12 | Ricoh Company, Ltd. | Power supplying methods and apparatus that provide stable output voltage |
Family Cites Families (2)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
JPS57204928A (en) * | 1981-06-11 | 1982-12-15 | Sony Corp | Stabilized power supply circuit |
JPH0973332A (en) * | 1995-09-04 | 1997-03-18 | Canon Inc | Constant current supply device |
-
2005
- 2005-11-18 JP JP2005334446A patent/JP4832056B2/en not_active Expired - Fee Related
- 2005-11-18 US US11/281,878 patent/US7292015B2/en not_active Expired - Fee Related
Patent Citations (47)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US4502152A (en) * | 1978-08-16 | 1985-02-26 | Lucas Industries Limited | Low current linear/high current chopper voltage regulator |
US4378530A (en) * | 1979-07-04 | 1983-03-29 | Unisearch Limited | High-efficiency low-distortion amplifier |
US4727308A (en) * | 1986-08-28 | 1988-02-23 | International Business Machines Corporation | FET power converter with reduced switching loss |
US4943902A (en) * | 1987-11-23 | 1990-07-24 | Viteq Corporation | AC to DC power converter and method with integrated line current control for improving power factor |
US4959606A (en) * | 1989-01-06 | 1990-09-25 | Uniphase Corporation | Current mode switching regulator with programmed offtime |
US5083078A (en) * | 1990-05-12 | 1992-01-21 | Daimler-Benz Ag | Device for supplying power to an electronic computer in a motor vehicle |
US5305192A (en) * | 1991-11-01 | 1994-04-19 | Linear Technology Corporation | Switching regulator circuit using magnetic flux-sensing |
US5258701A (en) * | 1992-09-02 | 1993-11-02 | The United States Of America As Represented By The Secretary Of The Army | DC power supply |
US5479090A (en) * | 1993-11-24 | 1995-12-26 | Raytheon Company | Power converter having optimal dynamic operation |
US5414341A (en) * | 1993-12-07 | 1995-05-09 | Benchmarq Microelectronics, Inc. | DC-DC converter operable in an asyncronous or syncronous or linear mode |
US6046516A (en) * | 1994-10-28 | 2000-04-04 | Siemens Aktiengesellschaft | Electronic switch for use with inductive loads |
US5600234A (en) * | 1995-03-01 | 1997-02-04 | Texas Instruments Incorporated | Switch mode power converter and method |
US5949229A (en) * | 1996-08-28 | 1999-09-07 | Samsung Electronics, Co., Ltd. | Power factor correction circuit having an error signal multiplied by a current signal |
US5929620A (en) * | 1996-11-07 | 1999-07-27 | Linear Technology Corporation | Switching regulators having a synchronizable oscillator frequency with constant ramp amplitude |
US5903447A (en) * | 1997-07-23 | 1999-05-11 | Murata Manufacturing Co., Ltd. | Current-mode control device and switching power supply employing same |
US5905407A (en) * | 1997-07-30 | 1999-05-18 | Motorola, Inc. | High efficiency power amplifier using combined linear and switching techniques with novel feedback system |
US6222356B1 (en) * | 1998-04-01 | 2001-04-24 | Siemens Aktiengesellschaft | Current mode switching regulator configured such that a measuring resistor is not needed to measure the current at an inductor |
US6307356B1 (en) * | 1998-06-18 | 2001-10-23 | Linear Technology Corporation | Voltage mode feedback burst mode circuit |
US6541947B1 (en) * | 1998-09-10 | 2003-04-01 | Robert Bosch Gmbh | Step-down constant-current transformer |
US6066943A (en) * | 1998-10-08 | 2000-05-23 | Texas Instruments Incorporated | Capacitive-summing switch-mode power conversion control |
US6034517A (en) * | 1998-10-27 | 2000-03-07 | Linear Technology Corporation | High efficiency step-down switching regulators |
US5982160A (en) * | 1998-12-24 | 1999-11-09 | Harris Corporation | DC-to-DC converter with inductor current sensing and related methods |
US6404261B1 (en) * | 1999-03-27 | 2002-06-11 | Koninklijke Philips Electronics N.V. | Switch circuit and semiconductor switch, for battery-powered equipment |
US6268756B1 (en) * | 1999-03-31 | 2001-07-31 | Sony Corporation | Fast high side switch for hard disk drive preamplifiers |
US6249110B1 (en) * | 1999-04-16 | 2001-06-19 | Robert Bosch Gmbh | Circuit configuration for generating a stabilized power supply voltage |
US6522178B2 (en) * | 1999-04-22 | 2003-02-18 | International Rectifier Corporation | Controlling high side devices without using level shift switches |
US6313610B1 (en) * | 1999-08-20 | 2001-11-06 | Texas Instruments Incorporated | Battery protection circuit employing active regulation of charge and discharge devices |
US6636023B1 (en) * | 1999-10-14 | 2003-10-21 | Juniper Networks, Inc. | Combined linear and switching voltage regulator |
US6166528A (en) * | 1999-11-02 | 2000-12-26 | Fairchild Semiconductor Corporation | Lossless current sensing in buck converters working with low duty cycles and high clock frequencies |
US6229289B1 (en) * | 2000-02-25 | 2001-05-08 | Cadence Design Systems, Inc. | Power converter mode transitioning method and apparatus |
US6611131B2 (en) * | 2000-05-23 | 2003-08-26 | Linear Technology Corp. | Cancellation of slope compensation effect on current limit |
US20030025484A1 (en) * | 2000-05-23 | 2003-02-06 | Linear Technology Corporation | Cancellation of slope compensation effect on current limit |
US6498466B1 (en) * | 2000-05-23 | 2002-12-24 | Linear Technology Corp. | Cancellation of slope compensation effect on current limit |
US6486643B2 (en) * | 2000-11-30 | 2002-11-26 | Analog Technologies, Inc. | High-efficiency H-bridge circuit using switched and linear stages |
US6476589B2 (en) * | 2001-04-06 | 2002-11-05 | Linear Technology Corporation | Circuits and methods for synchronizing non-constant frequency switching regulators with a phase locked loop |
US6366070B1 (en) * | 2001-07-12 | 2002-04-02 | Analog Devices, Inc. | Switching voltage regulator with dual modulation control scheme |
US6509721B1 (en) * | 2001-08-27 | 2003-01-21 | Koninklijke Philips Electronics N.V. | Buck regulator with ability to handle rapid reduction of load current |
US6661210B2 (en) * | 2002-01-23 | 2003-12-09 | Telfonaktiebolaget L.M. Ericsson | Apparatus and method for DC-to-DC power conversion |
US6828766B2 (en) * | 2002-05-30 | 2004-12-07 | Stmicroelectronics S.R.L. | Voltage regulator |
US6894471B2 (en) * | 2002-05-31 | 2005-05-17 | St Microelectronics S.R.L. | Method of regulating the supply voltage of a load and related voltage regulator |
US6744241B2 (en) * | 2002-06-07 | 2004-06-01 | Infineon Technologies Ag | Method for driving a switch in a switch-mode converter, and a drive circuit for driving a switch |
US6661211B1 (en) * | 2002-06-25 | 2003-12-09 | Alcatel Canada Inc. | Quick-start DC-DC converter circuit and method |
US6873140B2 (en) * | 2002-07-12 | 2005-03-29 | Stmicroelectronics S.R.L. | Digital contoller for DC-DC switching converters |
US7148665B2 (en) * | 2002-07-25 | 2006-12-12 | Ricoh Company, Ltd. | Power supplying methods and apparatus that provide stable output voltage |
US7129681B2 (en) * | 2002-08-23 | 2006-10-31 | Ricoh Company, Ltd. | Power supply apparatus having parallel connected switching and series regulators and method of operation |
US6724174B1 (en) * | 2002-09-12 | 2004-04-20 | Linear Technology Corp. | Adjustable minimum peak inductor current level for burst mode in current-mode DC-DC regulators |
US7030596B1 (en) * | 2003-12-03 | 2006-04-18 | Linear Technology Corporation | Methods and circuits for programmable automatic burst mode control using average output current |
Cited By (12)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US7271660B1 (en) * | 2005-05-11 | 2007-09-18 | National Semiconductor Corporation | Selectively adding auxiliary frequency compensation depending on the behaviour of an output transistor of a rail-to-rail operational amplifier |
US8847564B2 (en) * | 2007-07-06 | 2014-09-30 | Advanced Analogic Technologies, Incorporated | DC/DC converter using synchronous freewheeling MOSFET |
US20130342185A1 (en) * | 2011-03-03 | 2013-12-26 | Nec Corporation | Power supplying apparatus and control method thereof |
US20140042999A1 (en) * | 2012-08-10 | 2014-02-13 | Texas Instruments Incorporated | Switched mode assisted linear regulator with ac coupling with capacitive charge control |
US20150155783A1 (en) * | 2012-08-10 | 2015-06-04 | Texas Instruments Incorporated | Switched mode assisted linear regulator with dynamic buck turn-off using zcd-controlled tub switching |
US9112413B2 (en) * | 2012-08-10 | 2015-08-18 | Texas Instruments Incorporated | Switched mode assisted linear regulator with AC coupling with capacitive charge control |
US9112409B2 (en) * | 2012-08-10 | 2015-08-18 | Texas Instruments Incorporated | Switched mode assisted linear regulator with dynamic buck turn-off using ZCD-controlled tub switching |
US9473023B2 (en) | 2012-08-10 | 2016-10-18 | Texas Instruments Incorporated | Switched mode assisted linear regulator with seamless transition between power tracking configurations |
US9250694B1 (en) * | 2013-05-10 | 2016-02-02 | Sridhar Kotikalapoodi | Method and apparatus for fast, efficient, low noise power supply |
US20150364991A1 (en) * | 2014-06-16 | 2015-12-17 | City University Of Hong Kong | Input filter for a power electronic system |
US9698672B2 (en) * | 2014-06-16 | 2017-07-04 | City University Of Hong Kong | Input filter for a power electronic system |
US11194356B2 (en) * | 2019-06-28 | 2021-12-07 | Analog Devices International Unlimited Company | Linear stage efficiency techniques for H-bridge systems |
Also Published As
Publication number | Publication date |
---|---|
JP4832056B2 (en) | 2011-12-07 |
US7292015B2 (en) | 2007-11-06 |
JP2006158193A (en) | 2006-06-15 |
Similar Documents
Publication | Publication Date | Title |
---|---|---|
US7292015B2 (en) | High efficiency, high slew rate switching regulator/amplifier | |
EP0846996B1 (en) | Power transistor control circuit for a voltage regulator | |
US7345459B1 (en) | Emulated inductor current automatic correction without knowledge of actual inductor current ramp for emulated peak control mode PWM | |
US7656139B2 (en) | Creating additional phase margin in the open loop gain of a negative feedback amplifier system using a boost zero compensating resistor | |
US7176668B2 (en) | Switching regulator with advanced slope compensation | |
US9882408B2 (en) | Battery charging system including current observer circuitry to avoid battery voltage overshoot based on battery current draw | |
US6593725B1 (en) | Feed-forward control for DC-DC converters | |
US8334681B2 (en) | Domino voltage regulator (DVR) | |
US5672959A (en) | Low drop-out voltage regulator having high ripple rejection and low power consumption | |
US8026708B2 (en) | Voltage regulator | |
US11543843B2 (en) | Providing low power charge pump for integrated circuit | |
EP0993104A2 (en) | Capacitive-summing switch-mode power conversion control and method | |
EP1514163A2 (en) | Multimode voltage regulator | |
CN109391147B (en) | Step-down voltage converter | |
US20230229182A1 (en) | Low-dropout regulator for low voltage applications | |
US9817427B2 (en) | Static offset reduction in a current conveyor | |
US6522114B1 (en) | Noise reduction architecture for low dropout voltage regulators | |
CN112306130A (en) | Low Dropout (LDO) voltage regulator circuit | |
US6437638B1 (en) | Linear two quadrant voltage regulator | |
US9774251B2 (en) | Boost converter with improved stability | |
CN117318484A (en) | DC-DC converter control circuit | |
US12062980B2 (en) | DC-DC converter circuit | |
US7298121B2 (en) | Circuit and method for increasing the stability of switch-mode power supplies | |
CN117280294A (en) | Auxiliary circuit, chip system and device for LDO | |
Rao et al. | ON Chip LDO voltage regulator with improved transient response in 180nm |
Legal Events
Date | Code | Title | Description |
---|---|---|---|
AS | Assignment |
Owner name: MATSUSHITA ELECTRIC INDUSTRIAL CO., LTD., JAPAN Free format text: ASSIGNMENT OF ASSIGNORS INTEREST;ASSIGNORS:OSWALD, RICHARD;YAMAMOTO, TAMOTSU;ISHI, TAKUYA;REEL/FRAME:017631/0941 Effective date: 20060223 |
|
STCF | Information on status: patent grant |
Free format text: PATENTED CASE |
|
FEPP | Fee payment procedure |
Free format text: PAYOR NUMBER ASSIGNED (ORIGINAL EVENT CODE: ASPN); ENTITY STATUS OF PATENT OWNER: LARGE ENTITY |
|
FPAY | Fee payment |
Year of fee payment: 4 |
|
AS | Assignment |
Owner name: PANASONIC CORPORATION, JAPAN Free format text: CHANGE OF NAME;ASSIGNOR:MATSUSHITA ELECTRIC INDUSTRIAL CO., LTD.;REEL/FRAME:031947/0358 Effective date: 20081001 |
|
AS | Assignment |
Owner name: PANASONIC CORPORATION, JAPAN Free format text: LIEN;ASSIGNOR:COLLABO INNOVATIONS, INC.;REEL/FRAME:031997/0445 Effective date: 20131213 |
|
AS | Assignment |
Owner name: COLLABO INNOVATIONS, INC., CANADA Free format text: ASSIGNMENT OF ASSIGNORS INTEREST;ASSIGNOR:PANASONIC CORPORATION;REEL/FRAME:033021/0806 Effective date: 20131212 |
|
FPAY | Fee payment |
Year of fee payment: 8 |
|
FEPP | Fee payment procedure |
Free format text: MAINTENANCE FEE REMINDER MAILED (ORIGINAL EVENT CODE: REM.); ENTITY STATUS OF PATENT OWNER: LARGE ENTITY |
|
LAPS | Lapse for failure to pay maintenance fees |
Free format text: PATENT EXPIRED FOR FAILURE TO PAY MAINTENANCE FEES (ORIGINAL EVENT CODE: EXP.); ENTITY STATUS OF PATENT OWNER: LARGE ENTITY |
|
STCH | Information on status: patent discontinuation |
Free format text: PATENT EXPIRED DUE TO NONPAYMENT OF MAINTENANCE FEES UNDER 37 CFR 1.362 |
|
FP | Lapsed due to failure to pay maintenance fee |
Effective date: 20191106 |