TWI774315B - Motor control device and motor control method - Google Patents

Abstract

The present disclosure relates to a motor control method, including the following steps: receiving a frequency command and an excitation current setting value as a motor speed command; performing a magnetic flux calculation program to generate a magnetic flux voltage command; obtaining a synchronous coordinate voltage command, and generating a three-phase current to a sensorless motor; calculating a synchronous coordinate feedback current, and calculating a current effective value of the three-phase current according to the three-phase current; calculating a virtual power feedback value according to the synchronous coordinate voltage command and the synchronous coordinate feedback current; performing a steady state calculation program to calculate a virtual power command according to the frequency command and the current effective value; calculating a virtual power error value between the virtual power command and the virtual power feedback value; adding the magnetic flux voltage command and the virtual work error value to adjust the synchronous coordinate voltage command and change the three-phase current.

Description

Motor control device and motor control method

The present disclosure relates to a motor control device and a motor control method for driving a sensorless motor to operate.

With the changes in energy policies in countries around the world, the requirements for the operating efficiency of motors are also increasing. Synchronous reluctance motor (SynRM) has key advantages such as high operating efficiency, low manufacturing cost, robust mechanical structure, etc., and synchronous reluctance motor also has the advantages of permanent magnet motor and induction motor. Therefore, in order to meet the needs of energy policy, the research of synchronous reluctance motor is also paid more and more attention.

At present, the control of the synchronous reluctance motor mostly needs to be equipped with a position sensor or an encoder to facilitate the motor control. However, the use of position sensors or encoders tends to significantly increase manufacturing costs. In addition, in miniaturized motor products, the limited space cannot easily accommodate position sensors or encoders.

In addition, many studies on motors without position sensors require the use of estimators, such as flux observers and position and speed estimators, to perform algorithms. to obtain information such as the position and speed of the motor. However, the application of the estimator requires a large number of correct motor parameters to establish a motor model, so as to accurately calculate the information such as the position and rotation speed of the motor.

The present disclosure relates to a motor control method for a sensorless motor, comprising the following steps: receiving a frequency command and an excitation current setting value as a motor speed command; running a magnetic flux calculation program according to the motor speed command to Generate a magnetic flux voltage command; convert the magnetic flux voltage command into a synchronous coordinate voltage command, and then generate a three-phase current to the motor; calculate the synchronous coordinate feedback current according to the three-phase current, and then calculate the current effective value of the three-phase current; According to the synchronous coordinate voltage command and the synchronous coordinate feedback current, calculate the virtual power feedback value; according to the frequency command and current RMS, run the steady-state operation program to calculate the virtual power command; calculate the difference between the virtual power command and the virtual power feedback value The virtual power error value of ; and superimpose the magnetic flux voltage command and virtual power error value, adjust the synchronous coordinate voltage command, and then change the three-phase current.

The present disclosure also relates to a motor control method for a sensorless motor, comprising the following steps: receiving a frequency command and an excitation current setting value as a motor speed command; and running a magnetic flux calculation program according to the motor speed command to Generate magnetic flux voltage command; convert the magnetic flux voltage command into synchronous coordinate voltage command, and then generate three-phase current to the motor without sensor; calculate the synchronous coordinate feedback current according to the three-phase current, and then calculate the three-phase current According to the synchronous coordinate voltage command and the synchronous coordinate feedback current, calculate the real power feedback value; according to the real power feedback value and the current effective value, calculate the air gap power variation; Adjust the synchronous coordinate voltage command, and thereby change the three-phase current to the sensorless motor.

The present disclosure also relates to a motor control device for a sensorless motor. The motor control device includes a magnetic flux calculation unit, a drive calculation unit, a feedback calculation unit and a compensation calculation unit. The magnetic flux computing unit is used for receiving the frequency command and the setting value of the excitation current to calculate the magnetic flux voltage command. The driving arithmetic unit is used to convert the magnetic flux voltage command into the synchronous coordinate voltage command, and then generate three-phase current to the motor. The feedback operation unit is used to calculate the synchronous coordinate feedback current and obtain the current effective value of the three-phase current according to the three-phase current. The feedback computing unit calculates the virtual power feedback value according to the synchronous coordinate voltage command and the synchronous coordinate feedback current. The compensation operation unit is used to calculate the virtual power command according to the frequency command and the current RMS. The compensation operation unit is used to calculate the virtual power error value between the virtual power command and the virtual power feedback value. The driving operation unit is also used to superimpose the magnetic flux voltage command and the virtual power error value to adjust the synchronous coordinate voltage command, thereby changing the three-phase current.

The present disclosure mainly proposes a driving technology of a synchronous reluctance motor without a position sensor. The present disclosure uses only basic motor parameters, and does not need to consider nonlinear parameters, and is paired with compensation methods that improve efficiency and stability. Therefore, the present disclosure has the operational performance of high efficiency and good stability, and has the advantages of low motor parameter quantity and simpler calculation method compared with the conventional method.

Several embodiments of the present invention will be disclosed in the drawings below, and for the sake of clarity, many practical details will be described together in the following description. It should be understood, however, that these practical details should not be used to limit the invention. That is, in some embodiments of the invention, these practical details are unnecessary. In addition, for the purpose of simplifying the drawings, some well-known structures and elements will be shown in a simple and schematic manner in the drawings. It should be noted that the term "sensorless motor" hereinafter generally refers to a motor without a position sensor (eg, an encoder).

Please refer to FIG. 1A , FIG. 1B , FIG. 1C , FIG. 1D , FIG. 1E , FIG. 2A , and FIG. 2B at the same time to describe the following embodiments. As shown in FIG. 1A , the motor control device 100 includes a magnetic flux computing unit 110 , a compensation computing unit 120 , a stability computing unit 130 , a driving computing unit 140 , a feedback computing unit 150 and a current sensing device 160 . The motor control device 100 is used to drive the sensorless motor 200 . The motor 200 may be a synchronous reluctance motor, but the invention is not limited thereto.

As shown in FIG. 2A and FIG. 2B , the present invention provides a motor control method 300 , and the motor control method 300 is suitable for the sensorless motor 200 . The motor control method 300 includes steps S201 to S211 , and the motor control method 300 is executed by the motor control device 100 (as shown in FIG. 1A ).

As shown in FIGS. 1A , 1B and 2A, in step S201 , the magnetic flux calculation unit 110 of the motor control device 100 receives the frequency command ω _e and the excitation current set value Io as the motor rotational speed command. In some embodiments, as shown in FIG. 1B , the magnetic flux computing unit 110 includes a predetermined magnetic flux curve 110 a and a magnetic flux computing program 110 b. The predetermined magnetic flux curve 110a is a curve obtained according to the excitation current setting value Io and the ratio of the frequency command ω _e (eg, Io/ω _e ), and is used to record the characteristics of the motor 200 . Through the predetermined magnetic flux curve 110a, the magnetic flux computing unit 110 can only receive the frequency command ω _e to obtain the excitation current set value Io, or only receive the excitation current set value Io to obtain the frequency command ω _e . Alternatively, in the case where the predetermined magnetic flux curve 110a is not set, the magnetic flux operation unit 110 simultaneously receives the excitation current setting value Io and the frequency command ω _e . In addition, the frequency command ω _e can generate the driving angle Θ _e through the 1/S operator. The 1/S operator is a Laplace transform or Fourier transform commonly used by those skilled in the art, because those in the art can understand its meaning and operation method, and the present disclosure is not limited thereto. , so it is not repeated here.

As shown in Fig. 1A, Fig. 1B and Fig. 2A, in step S202, the magnetic flux computing unit 110 runs the magnetic flux computing program 110b according to the motor rotational speed command (including the excitation current setting value Io and the frequency command ω _e ). to generate the magnetic flux voltage command V _vf . Please refer to FIG. 1A , FIG. 1B and FIG. 3 to describe the operation principle of the magnetic flux calculation program 110 b.

The magnetic flux calculation program 110b includes steps S301 to S304. In step S301, the magnetic flux computing unit 110 establishes the first straight-axis equation and the first quadrature-axis equation according to the motor rotational speed command. The first straight-axis equation is shown in equation (1), and the first quadrature-axis equation is shown in equation (2).

(1)

(1)

(2)

(2)

Among them, v _d : the direct-axis voltage of the motor stator, v _q : the quadrature-axis voltage of the motor stator, R _s : the resistance of the motor stator, L _d : the direct-axis self-inductance of the motor, L _q : the quadrature-axis self-inductance of the motor, i _d : the direct-axis current of the motor stator, i _q : the quadrature-axis current of the motor stator, ω _γ : the rotational speed of the motor rotor, p: the differential term factor. It is particularly noted that equation (1) and equation (2) represent the two-shaft voltage equations of the motor 200 . R _s , L _d , and L _q of equation (1) and equation (2) are linear parameters of the motor, and these linear parameters are preset programs of the motor control device 100 (or the magnetic flux operation unit 110 ). In addition, id and i _q are derived from the excitation current setting value Io, and ω _γ is derived from the frequency command ω _{e .}

)，以分別建立第二直軸方程式(如方程式(3)所示)及第二交軸方程式(如方程式(4)所示)。 In step S302, the magnetic flux operation unit 110 removes the differential term parameters of the first straight-axis equation and the first quadrature-axis equation (eg:

) to establish a second straight-axis equation (as shown in equation (3)) and a second quadrature-axis equation (as shown in equation (4)), respectively.

(3)

(3)

(4)

(4)

The reason for removing the differential term parameters of the first straight-axis equation and the first quadrature-axis equation is to assume that the motor 200 maintains a steady state (ie, the rotational speed of the motor 200 remains unchanged). Therefore, equation (3) is also regarded as a straight-axis steady state equation, and equation (4) is also regarded as a quadrature-axis steady state equation. At this time, _{id can also be regarded as the steady-state value of the direct-axis current, and i q can} also be regarded as the steady-state value of the quadrature-axis current.

In step S303, the magnetic flux computing unit 110 sets the quadrature axis parameters (eg: i _q ) of the second quadrature axis equation and the second quadrature axis equation to zero, respectively, to establish a third quadrature axis equation (eg, equation (5)) shown) and the third quadrature equation (as shown in equation (6)).

(5)

(5)

(6)

(6)

Setting the quadrature axis parameters of the second straight axis equation and the second quadrature axis equation to zero is to assume that the motor 200 is not connected to any load (ie, no load). It should be noted that, in order to improve the control efficiency of the motor 200, when calculating the magnetic flux voltage command V _vf , it is necessary to assume that the motor 200 maintains an ideal state, and the ideal state is that the motor 200 maintains a steady state and is no-load.

In step S304, the magnetic flux computing unit 110 calculates the magnetic flux voltage command V _vf according to the third straight-axis equation and the third quadrature-axis equation. Among them, the calculation method of the magnetic flux voltage command V _vf is shown in equation (7):

(7)

(7)

As shown in FIG. 1A and FIG. 2A, in step S203, the driving operation unit 140 receives the magnetic flux voltage command V _vf as the driving voltage signal Vs, and the driving operation unit 140 includes: a first coordinate conversion circuit 141, a second The coordinate conversion loop 142 and the modulation loop 143 are provided. The first coordinate conversion circuit 141 converts the magnetic flux voltage command V _vf into _a synchronous coordinate voltage command Va according to the voltage compensation angle _Θh , and the second coordinate conversion circuit ₁₄₂ generates three-phase according to the driving angle _Θe and the synchronous coordinate voltage command Va voltage V _abc , and the modulation circuit 143 switches the three-phase voltage V _abc to generate the three-phase current I _abc to the motor 200 . The generation method of the voltage compensation angle Θ _h is described in the following paragraphs. The synchronous coordinate voltage command Va includes: _a direct-axis voltage command V _δ and a quadrature-axis voltage command V _γ . It should be noted that when the motor control device 100 operates for the first time, since no feedback signal is generated, the magnetic flux voltage command V _vf is equal to the driving voltage signal V _s . In addition, since the operation principle of the driving operation unit 140 belongs to the conventional technology in the art, it is not repeated here. Alternatively, the motor control device 100 further includes a current sensing device 160 , and the current sensing device 160 is used to detect the three-phase current I _abc and transmit the measured value of the three-phase current I _abc to the feedback operation unit 150 . In other embodiments, the current sensing device 160 detects any two-phase current in the three-phase current I _abc to the feedback operation unit 150 . The feedback operation unit 150 calculates the unmeasured third-phase current according to the received two-phase currents.

As shown in FIGS. 1A and 2A, in step S204, the third coordinate conversion circuit 151 of the feedback operation unit 150 calculates the synchronous coordinate feedback current according to the measured value of the three-phase current I _abc and the driving angle Θ _e I _a , and the synchronous coordinate feedback current i _a includes the direct axis feedback current i _δ and the quadrature axis feedback current i _γ . Next, the current calculation loop 152 of the feedback calculation unit 150 calculates the effective current value Is of the three-phase current I _abc according to the direct-axis feedback current i _δ and the quadrature-axis feedback current i _γ , running equation (8). where Equation (8) is as follows:

(8)

(8)

As shown in FIGS. 1A and 2A, in step S205, the power calculation loop 153 of the feedback calculation unit 150 according to the synchronous coordinate voltage command Va (including the direct-axis voltage command V _δ and the quadrature-axis voltage command V _γ ) and the synchronization Coordinate feedback current ia (including direct axis feedback current i _δ and quadrature axis feedback current i _γ ), calculate the virtual work feedback value Qin. The power calculation loop 153 includes the equation (9), and calculates the virtual power feedback value Qin according to the equation (9). where equation (9), as follows:

(9)

(9)

As shown in FIGS. 1A and 2A, in step S206, the power calculation loop 153 of the feedback calculation unit 150 according to the synchronous coordinate voltage command Va (including the direct-axis voltage command V _δ and the quadrature-axis voltage command V _γ ) and the synchronization Coordinate feedback current ia (including direct axis feedback current i _δ and quadrature axis feedback current i _γ ), calculate the real power feedback value Pin. The power calculation loop 153 includes the equation (10), and calculates the real power feedback value Pin according to the equation (10). where equation (10), as follows:

(10)

(10)

As shown in FIG. 1A , FIG. 1C and FIG. 2B , in step S207 , the steady-state compensation loop 121 of the compensation operation unit 120 runs the steady-state operation program to calculate the virtual work according to the frequency command ω _e and the current effective value Is Command Qref. The steady-state compensation loop 121 includes a virtual power controller 121a and an error controller 121b, and the virtual power controller 121a is used to execute a steady-state operation program. Please refer to FIG. 1C, FIG. 2A, and FIG. 4 to illustrate the operation principle of the steady-state operation program 500:

The steady-state operation program 500 includes steps S501 to S503. In step S501, the virtual power controller 121a establishes the direct-axis steady state equation (as shown in equation (3)) and the quadrature-axis steady state equation (as shown in equation (4)) according to the frequency command ω _e and the current effective value Is ) to calculate the steady-state value of the direct-axis current id and the steady-state value of the _quadrature -axis current i _q .

In step S502, the virtual power controller 121a adjusts the difference between the steady-state value _{id of the direct-axis current and the steady-state value of the quadrature-axis current iq to} fall within the error (ideally: the steady-state value of the direct-axis current _id is equal to Steady-state value of quadrature axis current i _q ).

In step S503, when the virtual power controller 121a determines that the difference falls within the error, the direct-axis current steady-state value _id and the quadrature-axis current steady-state value _iq are substituted into the direct-axis steady state equation (equation (3)) and the quadrature-axis steady state equation (equation (4)) to calculate the virtual work command Qref. The calculation method of calculating the virtual work command Qref is shown in equation (11) and equation (12):

(11)

(11)

After arranging Equation (8) and Equation (11), Equation (12) can be obtained:

(12)

(12)

As shown in FIG. 1A , FIG. 1C and FIG. 2B , in step S208 , the steady-state compensation loop 121 of the compensation operation unit 120 calculates the error value between the virtual power command Qref and the virtual power feedback value Qin, and uses The error controller 121b outputs the error value as the virtual power error value V _com to the low-pass filter 123 to filter out noise. In some embodiments, the motor control device 100 directly superimposes the magnetic flux voltage command V _vf and the virtual power error value V _com as the driving voltage signal V _s , and adjusts the synchronous coordinate voltage command Va ₍ including The direct-axis voltage command V _δ and the quadrature-axis voltage command V _γ ), and then the three-phase current I _abc is changed to drive the motor 200 .

As shown in FIG. 1A , FIG. 1D and FIG. 2B , in step S209 , when the low-speed compensation circuit 122 of the compensation arithmetic unit 120 determines that the direct-axis feedback current i _δ of the motor 200 is smaller than the preset threshold value (representing the The motor 200 is running at a low speed), the low speed compensation circuit 122 calculates the error value i _err between the excitation current set value Io and the direct axis feedback current i _δ of the synchronous coordinate feedback current i _a , and uses the error controller 122a to calculate the error value i err An excitation error value _Vtor is generated. When the motor 200 operates at a low speed, the driving performance of the motor 200 will be significantly reduced. Therefore, the excitation error value _Vtor calculated by the low-speed compensation circuit 122 is used as compensation. The driving performance of the motor 200 that maintains a low speed can be effectively increased. In some embodiments, the signal obtained by superimposing the excitation error value V _tor and the virtual power error value V _com is filtered through the low-pass filter 123 to remove noise. The excitation error value V _tor is used to superimpose the magnetic flux voltage command V _vf and the virtual power error value V _com to adjust the synchronous coordinate voltage command _Va .

As shown in FIG. 1A , FIG. 1E and FIG. 2B , in step S210 , the stability controller 130 a of the stability calculation unit 130 calculates the air gap power value P according to the real power feedback value Pin and the current effective value Is _AG . The air-gap power (P _AG ) represents the power transferred from the air gap between the stator and the rotor to the rotor of the motor. The stability controller 130a includes equation (13) to calculate the air gap power value P _AG , and equation (13) is as follows:

(13)

(13)

Next, the high-pass filter 130b of the stability calculation unit 130 calculates the air-gap power variation ΔP _{AG according to the real-time air-gap power value P AG .} The air gap power variation ΔP _AG represents the vibration of the motor 200 , or the vibration of a load (not shown) connected to the motor 200 . Next, the error controller 130c of the stability computing unit 130 generates the voltage compensation angle Θ _h according to the air gap power variation ΔP _AG . Among them, the calculation method of the voltage compensation angle Θ _h is shown in equation (14):

Θ _h = -K _p ·ΔP _AG (14)

In equation (14), the voltage compensation angle Θ _h can be regarded as the oscillation of the motor 200 , -K _p is a negative proportional constant, wherein -K _p is a proportional constant inversely proportional to the output frequency. That is to say, the voltage compensation angle Θ _h represents the compensation amount that the motor 200 needs to generate a reaction torque to overcome the vibration of the motor 200 . Therefore, in order to reduce the aforementioned vibration of the motor 200, the stability calculation unit 130 provides the voltage compensation angle _Θh to the drive calculation unit 140 according to the air gap power variation ΔP _AG , so that the drive calculation unit 140 adjusts the synchronous coordinate voltage command _Va , and Further, the three-phase current I _abc is changed, so that the motor 200 generates a counter torque.

As shown in FIGS. 1A and 2B, in step S211, when the motor 200 operates at a low speed, the motor control device 100 mainly superimposes the excitation error value V _tor , the magnetic flux voltage command V _vf and the virtual power error value V _com is used as the driving voltage signal V _s , so that the driving operation unit 140 adjusts the synchronous coordinate voltage command Va and _{further changes the three-phase current I abc .} When the motor 200 is not running at a low speed, the motor control device 100 mainly superimposes the magnetic flux voltage command V _vf and the virtual power error value V _com as the driving voltage signal V _s , so that the driving arithmetic unit 140 adjusts the synchronous coordinate voltage command V _a , and thus change the three-phase current I _abc .

In the aforementioned steps S201 to S211, various compensations are performed for different conditions of the motor 200, but the present disclosure is not limited thereto. In one embodiment, if the motor 200 has _a low probability of low-speed torque, the motor control device 100 can adjust the synchronous coordinate voltage command Va only according to the magnetic flux voltage command V _vf and the virtual power error value V _com . Similarly, according to different control requirements, the motor control device 100 can also adjust the synchronous coordinate voltage command _Va only according to the air gap power variation ΔP _AG to change the three-phase current I _abc . In other words, the motor control apparatus 100 may selectively compensate for one or more conditions (ie, no-load condition, on-load condition, low speed torque condition, or air gap power variation) depending on driving conditions.

In the present disclosure, the motor 200 is driven first through steps S201 to S203, and then through steps S204 to _S206 , the synchronous coordinate feedback current i _a (direct axis feedback current i _δ and Feedback current i _γ ), current RMS value Is, virtual power feedback value Qin, real power feedback value and other data. Finally, through steps S207 to S211, the virtual power error value V _com , the excitation error value V _tor and the air gap power variation (ie, the corresponding voltage compensation angle Θ _h ) are generated to compensate the motor 200 under different conditions. error.

The "loop" and "controller" mentioned in the present disclosure may be digital logic circuits, hardware circuits or other programming languages, but the present disclosure is not limited thereto.

The various elements, method steps or technical features in the foregoing embodiments can be combined with each other, and are not limited by the order of description in the text or the order of presentation of the drawings in the present disclosure.

100: motor control device 110: magnetic flux calculation unit 110a: predetermined magnetic flux curve 110b magnetic flux calculation program 120: compensation calculation unit 121: steady state compensation circuit 121a: virtual power controller 121b: error controller 122: low speed compensation circuit 122a : error controller 123 : low-pass filter 130 : stability calculation unit 130 a : stability controller 130 b : high-pass filter 130 c : error controller 140 : drive calculation unit 141 : first coordinate conversion loop 142 : second coordinate conversion Loop 143: Modulation loop 150: Feedback calculation unit 151: Third coordinate conversion loop 152: Current calculation loop 153: Power calculation loop 160: Current sensing device 200: Motor ω _e : Frequency command V _a : Synchronous coordinate voltage command V _vf : magnetic flux voltage command V _s : driving voltage signal V _γ : quadrature axis voltage command V _δ : direct axis voltage command V _abc : three-phase voltage signal V _tor : excitation error value V _com : virtual power error value i _a : Synchronous coordinate feedback current I _abc : Three-phase current Io: Excitation current setting value i _γ : Quadrature axis feedback current i _δ : Direct axis feedback current i _err : Error value Is: Current effective value Pin: Real power feedback Value Qin: virtual power feedback value Qref: virtual power command Θ _h : voltage compensation angle Θ _e : driving angle P _AG : air gap power value ΔP _AG : air gap power variation 300: motor control method S201-S211: steps S301-S304: Step 500: Steady-state operation program S501-S503: Step 1/S: Operator

FIG. 1A is a schematic diagram of a motor control apparatus according to some embodiments of the present disclosure. FIG. 1B is a schematic diagram of a magnetic flux operation unit according to some embodiments of the present disclosure. 1C is a schematic diagram of a steady state compensation loop according to some embodiments of the present disclosure. 1D is a schematic diagram of a low speed compensation loop according to some embodiments of the present disclosure. FIG. 1E is a schematic diagram of a stability computing unit according to some embodiments of the present disclosure. 2A and 2B are flowcharts of motor control methods according to some embodiments of the present disclosure. FIG. 3 is a flow chart of a magnetic flux calculation routine according to some embodiments of the present disclosure. FIG. 4 is a flowchart of a steady state operation procedure according to some embodiments of the present disclosure.

Domestic storage information (please note in the order of storage institution, date and number) none Foreign deposit information (please note in the order of deposit country, institution, date and number) none

100: Motor control device

110: Magnetic flux operation unit

120: Compensation operation unit

121: steady state compensation loop

122: Low speed compensation loop

123: low pass filter

130: Stability operation unit

140: Drive arithmetic unit

141: The first coordinate conversion loop

142: Second coordinate conversion loop

143: Modulation loop

150: Feedback operation unit

151: The third coordinate conversion circuit

152: Current operation loop

153: Power calculation loop

160: Current Sensing Device

200: Motor

ω _e : frequency command

Va _: Synchronous coordinate voltage command

V _vf : Flux voltage command

V _s : driving voltage signal

V _γ : Quadrature axis voltage command

V _δ : Direct axis voltage command

V _abc : Three-phase voltage signal

V _tor : Excitation error value

V _com : virtual power error value

i _a : Synchronous coordinate feedback current

I _abc : three-phase current

Io: Excitation current setting value

i _γ : Feedback current of quadrature axis

i _δ : Direct axis feedback current

Is: current effective value

Pin: Real work feedback value

Qin: virtual work feedback value

Θ _h : Voltage compensation angle

Θ _e : drive angle

1/S: operator

Claims (15)

A motor control method for a sensorless motor, comprising: Receive a frequency command and an excitation current setting value as a motor speed command; According to the motor speed command, run a magnetic flux operation program to generate a magnetic flux voltage command; Converting the magnetic flux voltage command into a synchronous coordinate voltage command, and then generating a three-phase current to the motor; According to the three-phase current, calculate a synchronous coordinate feedback current, and then calculate a current effective value of the three-phase current; Calculate a virtual power feedback value according to the synchronous coordinate voltage command and the synchronous coordinate feedback current; According to the frequency command and the current effective value, run a steady-state operation program to calculate a virtual power command; calculating a virtual power error value between the virtual power command and the virtual power feedback value; and The magnetic flux voltage command and the virtual power error value are superimposed to adjust the synchronous coordinate voltage command, thereby changing the three-phase current. The motor control method of claim 1, further comprising: Calculate a real power feedback value according to the synchronous coordinate voltage command and the synchronous coordinate feedback current; Calculate an air-gap power variation according to the real power feedback value and the current RMS value; and According to the air-gap power variation, the synchronous coordinate voltage command is adjusted, and the three-phase current is further changed. The motor control method of claim 1, further comprising: calculating an error value between the excitation current set value and the straight-axis feedback current of the synchronous coordinate feedback current to generate an excitation error value, wherein The excitation error value is superimposed with the magnetic flux voltage command and the virtual power error value to adjust the synchronous coordinate voltage command. The motor control method as claimed in claim 1, wherein the magnetic flux calculation program comprises: establishing a first straight axis equation and a first quadrature axis equation according to the motor speed command; respectively removing the differential term parameters of the first straight-axis equation and the first quadrature-axis equation to establish a second straight-axis equation and a second quadrature-axis equation, respectively; respectively setting a quadrature axis parameter of the second quadrature axis equation and the second quadrature axis equation to zero to establish a third quadrature axis equation and a third quadrature axis equation, respectively; and The magnetic flux voltage command is calculated according to the third straight-axis equation and the third quadrature-axis equation. The motor control method as claimed in claim 4, wherein the steady state operation program comprises: According to the frequency command and the current effective value, establish a steady-state equation for a straight axis and a steady-state equation for a quadrature axis to calculate the steady-state value of the straight-axis current and a steady-state value of the quadrature axis current; adjusting a difference between the steady-state value of the direct-axis current and the steady-state value of the quadrature-axis current to fall within an error; and When the difference falls within the error, the direct-axis current steady-state value and the quadrature-axis current steady-state value are substituted into the direct-axis steady-state equation and the quadrature-axis steady-state equation to calculate the virtual work command. A motor control method for a sensorless motor, comprising: Receive a frequency command and an excitation current setting value as a motor speed command; According to the motor speed command, run a magnetic flux operation program to generate a magnetic flux voltage command; converting the magnetic flux voltage command into a synchronous coordinate voltage command, and then generating a three-phase current to the sensorless motor; According to the three-phase current, calculate a synchronous coordinate feedback current, and then calculate a current effective value of the three-phase current; Calculate a real power feedback value according to the synchronous coordinate voltage command and the synchronous coordinate feedback current; Calculate an air-gap power variation according to the real power feedback value and the current RMS value; and According to the air-gap power variation, the synchronous coordinate voltage command is adjusted, and then the three-phase current is changed to the sensorless motor. The motor control method as claimed in claim 6, further comprising: Calculate a virtual power feedback value according to the synchronous coordinate voltage command and the synchronous coordinate feedback current; According to the frequency command and the current effective value, run a steady-state operation program to calculate a virtual power command; Calculate a virtual power error value between the virtual power command and the virtual power feedback value; calculating an error value between the set value of the excitation current and the straight-axis feedback current of the synchronous coordinate feedback current to obtain an excitation error value; and The magnetic flux voltage command, the excitation error value and the virtual power error value are superimposed to adjust the synchronous coordinate voltage command, thereby changing the three-phase current to the sensorless motor. The motor control method as claimed in claim 7, wherein the magnetic flux calculation program comprises: establishing a first straight axis equation and a first quadrature axis equation according to the motor speed command; respectively removing the differential term parameters of the first straight-axis equation and the first quadrature-axis equation to establish a second straight-axis equation and a second quadrature-axis equation, respectively; respectively setting a quadrature axis parameter of the second quadrature axis equation and the second quadrature axis equation to zero to establish a third quadrature axis equation and a third quadrature axis equation, respectively; and The magnetic flux voltage command is calculated according to the third straight-axis equation and the third quadrature-axis equation. The motor control method as claimed in claim 8, wherein the steady state operation program comprises: According to the frequency command and the current effective value, establish a steady-state equation for a straight axis and a steady-state equation for a quadrature axis, so as to calculate a steady-state value of the straight-axis current and a steady-state value of the quadrature axis current; adjusting a difference between the steady-state value of the direct-axis current and the steady-state value of the quadrature-axis current to fall within an error; and When it is judged that the difference falls within the error, the direct-axis current steady-state value and the quadrature-axis current steady-state value are substituted into the direct-axis steady-state equation and the quadrature-axis steady-state equation to calculate the virtual work command. A motor control device for a sensorless motor, comprising: a magnetic flux computing unit for receiving a frequency command and an excitation current setting value to calculate a magnetic flux voltage command; a driving arithmetic unit for converting the magnetic flux voltage command into a synchronous coordinate voltage command, and then generating a three-phase current to the motor; and a feedback computing unit for calculating a synchronous coordinate feedback current according to the three-phase current and obtaining a current effective value of the three-phase current, wherein the feedback computing unit according to the synchronous coordinate voltage command and the synchronous coordinate feedback Give current, calculate a virtual work feedback value; a compensation arithmetic unit for calculating a virtual power command according to the frequency command and the current effective value; Wherein the compensation operation unit is used for calculating a virtual power error value between the virtual power command and the virtual power feedback value; The driving operation unit is also used for superimposing the magnetic flux voltage command and the virtual power error value to adjust the synchronous coordinate voltage command, thereby changing the three-phase current. The motor control device of claim 10, wherein the compensation calculation unit is further configured to calculate an error value between the excitation current setting value and the straight-axis feedback current of the synchronous coordinate feedback current, so as to obtain an excitation current The magnetic error value is superimposed with the magnetic flux voltage command and the virtual power error value to adjust the synchronous coordinate voltage command. The motor control device of claim 10, wherein the feedback computing unit is further configured to calculate a real power feedback value according to the synchronous coordinate voltage command and the synchronous coordinate feedback current. The motor control device of claim 12, further comprising: a stability calculation unit for calculating an air-gap power variation according to the real power feedback value and the current effective value; wherein the driving calculation unit also adjusts the synchronous coordinate voltage command according to the air-gap power variation, and further Change the three-phase current. The motor control device of claim 10, wherein the magnetic flux computing unit is further used to: establishing a first straight-axis equation and a first quadrature-axis equation according to the frequency command and the excitation current setting value; respectively removing the differential term parameters of the first straight-axis equation and the first quadrature-axis equation to establish a second straight-axis equation and a second quadrature-axis equation, respectively; respectively setting a quadrature axis parameter of the second quadrature axis equation and the second quadrature axis equation to zero to establish a third quadrature axis equation and a third quadrature axis equation, respectively; and The magnetic flux voltage command is calculated according to the third straight-axis equation and the third quadrature-axis equation. The motor control device of claim 10, wherein the compensation arithmetic unit is further used for: According to the frequency command and the current effective value, establish a steady-state equation for a straight axis and a steady-state equation for a quadrature axis, so as to calculate a steady-state value of the straight-axis current and a steady-state value of the quadrature axis current; adjusting a difference between the steady-state value of the direct-axis current and the steady-state value of the quadrature-axis current to fall within an error; and When it is judged that the difference falls within the error, the direct-axis current steady-state value and the quadrature-axis current steady-state value are substituted into the direct-axis steady-state equation and the quadrature-axis steady-state equation to calculate the virtual work command.

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