TWI699957B - A quasi-resonant power supply controller - Google Patents

Abstract

The invention relates to a quasi-resonant power supply controller. A quasi-resonant power supply controller is provided, which includes a primary side control module configured to receive a rectified AC input voltage and control the turn-on and turn-off of the gate. When the primary side switch is turned off, the secondary side switch is turned on until it senses After detecting the end of demagnetization, the secondary side switch is turned off. The primary side switch is connected in parallel with the primary side switch parasitic capacitance, and the switch is turned on when the voltage resonance of the primary side switch parasitic capacitance reaches the lowest value; and the secondary side control module; where the secondary side controls The module includes a control and sampling circuit, and is configured to sense the input voltage to set the reverse threshold when the secondary side switch is turned off.

Description

A quasi-resonant power supply controller

The invention relates to the field of integrated circuits. More specifically, the invention provides a method for implementing a ZVS (Zero Voltage Switch) quasi-resonant power supply controller with synchronous rectification.

The secondary-side synchronous rectifier chip can realize primary-side switching (Metal-Oxide-Semiconductor Field-Effect Transistor, MOSFET, metal oxide semiconductor field-effect transistor) under different input voltage conditions without sensing the valley voltage of the primary side. Zero voltage conduction, thus avoiding switching loss at the moment of conduction, and greatly improving the efficiency of the power supply. After achieving zero voltage turn-on, the switching frequency of the flyback converter can be increased to 200kHz or higher without affecting the system efficiency. After the operating frequency is increased, the magnetic core size of the transformer can be reduced, thereby reducing the size of the power supply.

Figure 1 is a diagram showing a typical flyback synchronous rectification quasi-resonant converter. Among them, the virtual box on the left represents the internal control structure of the primary side, and the virtual box on the right represents the internal control structure of the secondary side.

After the output voltage is divided, the secondary side feedback voltage FB voltage generated by the controllable stabilized voltage source TL431 and the optocoupler determines the turn-off time of the control signal gate. When the primary side switch is turned off, the secondary side starts to demagnetize, and the secondary side switch is turned on at this time Turn off the secondary switch until the end of demagnetization is sensed. After the demagnetization is over, series resonance occurs between the primary inductance L1 of the transformer and the parasitic capacitance C1 of the switch. When the parasitic capacitance C1 of the switch resonates to the lowest value, the voltage drop across the switch When the resonance reaches the valley voltage, the switch is turned on, so that the conduction loss of the switch can be minimized and the efficiency of the flyback converter is effectively improved.

Fig. 2 is a diagram showing the operating waveforms of the quasi-resonant switching power supply in Fig. 1. The turn-on voltage of the primary side switch is V _in -N*Vo (N is the transformer turns ratio, Vo is the output voltage). This turn-on voltage will increase with the increase of V _in , and the conduction loss will also increase. The switching frequency of the quasi-resonant power supply itself under high voltage is much higher than that under low voltage, and the valley turn-on voltage will also be very high, resulting in excessive switching losses under high input voltage, which has a greater negative impact on efficiency.

Due to the problem that the switching loss of the traditional flyback quasi-resonant converter increases as the input voltage and operating frequency increase, there are currently solutions that can add resonance to achieve zero voltage conduction and greatly reduce the switching loss.

Figure 3 is a diagram showing a typical zero voltage turn-on operating waveform. When the primary gate is turned on, the input voltage charges the primary inductor L1. When the primary side gate is turned off, the secondary side power MOS field effect transistor is turned on, and the secondary side inductor is demagnetized. When the secondary side current Is demagnetized, the secondary side switch is turned off after a delay. During this delay time Td, the output The voltage reversely charges the secondary side inductance to a certain extent. After the secondary side switch is turned off, the negative current of the secondary side is coupled to the primary side, which increases the resonance energy between the primary side’s inductance and the primary side MOS parasitic capacitance, and makes the MOS parasitic capacitance increase. The voltage Vds can be resonantly lower or even to 0V conduction. However, as the input voltage changes, the delay time Td required for Vds to resonate to 0V is different, and a fixed delay time Td cannot achieve zero voltage conduction under different input voltages.

At least in order to solve the problem of being unable to achieve zero voltage conduction under full voltage conditions, the present invention proposes a simple implementation method that allows Vds to resonate to about 0V conduction under different input voltages, and minimize the switching loss of the primary MOS.

The present invention provides a realization method of a ZVS quasi-resonant power supply controller with synchronous rectification. As an example only, some embodiments of the present invention are applied to a switching power supply. However, it will be recognized that the invention has a broader scope of applicability.

According to an embodiment of the present invention, a quasi-resonant power supply controller is provided, including: a primary side control module configured to receive a rectified AC (Alternate Current, alternating current power) input voltage and control the turn-on and turn-off of the gate , When the primary side switch is turned off, the secondary side switch is turned on until the end of demagnetization is sensed and then the secondary side switch is turned off. The primary side switch is equivalently connected in parallel with the primary side switch Parasitic capacitance, when the voltage on the parasitic capacitance of the primary side switch resonates to the lowest value, the switch is turned on; and the secondary side control module; the secondary side control module includes a control and sampling circuit configured to sense the input voltage to set the secondary side switch Reverse threshold at shutdown.

TL431: Controllable voltage source

Td: Delay time

FB: Secondary side feedback voltage

I ₁ : Secondary side reverse current

C _ds : Primary side switch parasitic capacitance

t0~t5: time

L1, Lp: Primary side inductance

Vsns: Secondary side switch drain voltage

L _s : secondary side inductance

Vth: reverse threshold

C1: Switch parasitic capacitance

Rdson: Secondary switch on impedance

V _in : input voltage

R4: resistance

Vo: output voltage

Iin: Primary current

N: Transformer turns ratio

Is: secondary side current

inv: demagnetization sensing voltage

VCS: Current sampling voltage

DEM: Demagnetization sensing module

comp: comparator

gate: Primary side switch gate signal

SR_gate: Secondary side switch gate signal

ZVS: Zero Voltage Switching

2us LEB: Timer module

I _ZVS : Control current

R: set to 0

S: set 1 end

Q: output

R1, R2, R3: external adjustment resistor

100: control and sampling circuit

Figure 1 is a diagram showing a typical flyback synchronous rectification quasi-resonant converter.

Fig. 2 is a diagram showing the operating waveforms of the quasi-resonant switching power supply in Fig. 1.

Figure 3 is a diagram showing a typical zero-voltage turn-on operating waveform.

Figure 4a shows a diagram of a working waveform for achieving zero voltage conduction according to an embodiment of the present invention.

Figure 4b shows the equivalent circuit corresponding to the different working stages shown in Figure 4a.

Figure 5 shows a schematic diagram of a zero-voltage turn-on quasi-resonant switching power supply with synchronous rectification according to an embodiment of the present invention.

Figure 6 shows the working waveforms of the key points of the secondary side of the quasi-resonant switching power supply according to Figure 5.

Figure 7 shows an implementation of the ZVS-enabled synchronous rectification controller according to an embodiment of the present invention.

Figure 8 shows a diagram representing the operating waveforms of each key point according to the implementation of Figure 7.

Figure 4a shows a diagram of a working waveform for achieving zero voltage conduction according to an embodiment of the present invention. As shown in Figure 4a, a work cycle can be divided into five work phases. Each phase of the analysis are as follows: t0 ~ t1: t0 primary time switch is turned on, the input voltage V _in to transformer primary inductance Lp charge, until the time t1, the primary gate turn-off.

t1~t2: After the primary gate is turned off, the transformer secondary inductance L _s discharges the output voltage Vo, until the discharge ends at t2, the secondary current Is is zero.

t2~t3: After the transformer secondary inductance L _s discharges to the output, since the secondary switch is still in the on state, the output voltage Vo will reversely charge the transformer negative inductance L _s to the secondary reverse current I ₁ .

t3~t4: At t3, the secondary side switch is turned off, and the secondary side reverse current is induced to the primary side of the transformer. The direction is shown by the arrow in Figure 4(b). The induced current is equal to I ₁ /N. The terminal voltage Vds=V _in +N*Vo. After the secondary side switch is turned off, the primary side inductance Lp and the primary side switch parasitic capacitance C _ds undergo series resonance. Because the voltage Vds across the primary side switch parasitic capacitance C _ds is higher than the input voltage V _in , C _ds will affect the primary side inductance Lp discharge, the voltage across C _ds decreases, the primary current Iin increases, until the time t4, the voltage Vds of the input voltage is reduced to be equal to V _in, the resonant inductor current reverse to the maximum, then the energy stored in the inductor L _s . I ₁ ² +C _ds (N. Vo) ² .

C_ds．V_in ² t4~t5: starting at t4, the primary inductor Lp is discharged, the primary switching parasitic capacitance C _{ds is} reversely charged, that is, Vds continues to decrease, until t5, the inductor current decreases to zero, and the voltage across the switching parasitic capacitance Vds resonates to The lowest value, according to energy conservation, if Vds is to resonate to 0V, the following equation L _s needs to be satisfied. I ₁ ² +C _ds (N. _{V o} ) ²

C _ds . V _in ²

Therefore, the secondary reverse current condition required to achieve zero voltage conduction is derived as

Figure 4b shows the equivalent circuit corresponding to the different working stages shown in Figure 4a.

a V_in-b，a、b數值與變壓器感量、匝比及原邊開關寄生電容大小有關。因此，可以通過感測輸入電壓調整副邊反向電流的大小以實現原邊開關的零電壓導通。 Under the condition of fixed system parameters, the magnitude of the secondary reverse current I ₁ required to realize the zero-voltage conduction of the primary side switch is only related to the input voltage V _in , and the condition I _{1 is} realized

a V _in -b, the values of a and b are related to the transformer inductance, turns ratio and the parasitic capacitance of the primary switch. Therefore, the magnitude of the reverse current of the secondary side can be adjusted by sensing the input voltage to realize the zero voltage conduction of the primary side switch.

Figure 5 shows a schematic diagram of a zero-voltage turn-on quasi-resonant switching power supply with synchronous rectification according to an embodiment of the present invention. As shown, the primary side of the control structure remains unchanged, the secondary threshold value is set reverse Vth (Vth = I ₁ .Rdson) when the secondary-side switch is turned off by sensing the input voltage Vsns V _in, Rdson represents secondary side Switch on impedance.

Figure 6 shows the working waveforms of the key points of the secondary side of the quasi-resonant switching power supply according to Figure 5. When the primary switch is turned on, the secondary switch drains the extreme voltage Vsns=V _in /N-Vo, where the output voltage Vo and the transformer turns ratio N are fixed, that is, this voltage is only related to the input voltage, so the secondary switch is turned off off time can be obtained sampling voltage Vsns input voltage V _in, and then setting the relationship between the input voltage and Vth depending on the system parameters, to achieve zero-voltage turn-on of the primary switch at different input voltages.

Figure 7 shows an implementation of the ZVS-enabled synchronous rectification controller according to an embodiment of the present invention.

Figure 8 shows a diagram representing the operating waveforms of each key point according to the implementation of Figure 7. When the primary switch is turned on, that is, when the secondary switch is turned off, Vsns=V _in /N+Vo.

According to the internal structure in Figure 7, we get

Therefore, the Vsns voltage is first divided by R ₁ and R ₂ , and the divided voltage generates a current flowing through the resistor R ₃ , and the current on the resistor R ₃ is then generated by CCCS (Current Controlled Current Source). The control current I _{ZVS of ZVS} , the control current I _ZVS flows through the resistor R ₄ , and the generated voltage is I _ZVS × R _4. Use the sample signal shown in Figure 8 to sample the voltage of the resistor R ₄ to obtain a correlation with the input voltage V _in The reverse threshold Vth when the secondary side switch is turned off. The reverse current calculation formula when the secondary side switch is turned off is as follows

The Vsns voltage and a fixed threshold (for example, as shown in the figure -300mV) are sent to the comparator, and then the timer module 2us LEB with the minimum turn-on masking time is passed, and the RS flip-flop is generated. The set 1 signal is used as the secondary side switch gate signal SR_gate to turn on the secondary side switch; the Vsns voltage and the reverse threshold Vth when the secondary side is turned off are sent to the comparator to generate the 0 set terminal R of the RS flip-flop as the secondary side. Switch gate signal SR_gate to turn off the secondary side switch.

After the output voltage is divided, the secondary side feedback voltage FB is generated through the controllable stabilized voltage source TL431 and the optocoupler. The secondary side feedback voltage FB and the current sampling voltage VCS are sent to the comparator to generate the set 0 signal of the RS trigger as the primary side switch. Pole signal gate to turn off the primary switch; the demagnetization sensing voltage inv is generated by the demagnetization sensing module DEM. The set 1 signal of the set 1 terminal S of the RS trigger is output through the output terminal Q to turn on the primary switch, which can realize the original The zero voltage of the side switch turns on.

The resistance of R4 is internally fixed, Rdson is the on-resistance of the secondary side switch, and R1, R2, and R3 are external adjustment resistors. According to the secondary reverse current conditions required for zero voltage turn-on: Just adjust the size of the external adjustment resistors R1, R2, R3 according to the transformer turns ratio N, the secondary side inductance L _s and the size of the primary side switch parasitic capacitance. Zero voltage conduction is achieved under voltage.

According to the embodiment of the present invention, the ZVS realization circuit only needs to add a simple internal control and sampling circuit on the basis of the original synchronous rectification internal control circuit, and then add three adjusting resistors to achieve zero voltage under different input voltages of different systems. Turn on, greatly improving the efficiency of the flyback converter.

The quasi-resonant power supply controller according to the embodiment of the present invention can be used to control various quasi-resonant switching power supply systems, including various topologies including flyback converters. The external circuit is simple, only need to adjust the external resistance according to different system parameters to achieve zero voltage conduction. At the same time, zero voltage conduction is achieved within the full input voltage range, which maximizes the system efficiency.

According to an example, a quasi-resonant power supply controller is provided, including: a primary side control module configured to receive a rectified AC input voltage and control the turn-on and turn-off of the gate. When the primary side switch is turned off, the turn-on secondary The side switch turns off the secondary side switch until the end of demagnetization is sensed, wherein the primary side switch is connected in parallel with the primary side switch parasitic capacitance C _ds , and the switch is turned on when the voltage on the primary side switch parasitic capacitance C _ds resonates to the lowest value; and the secondary side Control module; wherein the secondary side control module includes a control and sampling circuit 100, configured to sense the input voltage to set the reverse threshold when the secondary side switch is turned off.

The present invention can be implemented in other specific forms without departing from its spirit and essential characteristics. For example, the algorithm described in the specific embodiment can be modified, and the system architecture does not deviate from the basic spirit of the present invention. Therefore, the current embodiments are regarded as illustrative rather than restrictive in all aspects. The scope of the present invention is defined by the scope of the appended patent application instead of the above description, and the meaning and equivalent of the scope of the patent application are defined. All changes within the scope of things are thus included in the scope of the present invention.

Some or all of the elements in the various embodiments of the present invention, individually and/or in combination with at least another element, utilize one or more software elements, one or more hardware elements, and/or one of software and hardware elements Or multiple combinations to achieve. In another example, some or all of the elements in the various embodiments of the present invention are implemented in one or more circuits individually and/or in combination with at least another element, for example in one or more analog circuits and/or Implemented in one or more digital circuits. In yet another example, various embodiments and/or examples of the present invention may be combined.

Although specific embodiments of the present invention have been described, those skilled in the art will understand that there are other embodiments equivalent to the described embodiments. Therefore, it will be understood that the present invention is not limited by the specific embodiments shown, but only by the scope of the patent application.

AC: AC power

FB: Secondary side feedback voltage

L1: Primary side inductance

inv: demagnetization sensing voltage

ZVS: Zero Voltage Switching

VCS: Current sampling voltage

gate: Primary side switch gate signal

DEM: Demagnetization sensing module

Is: secondary side current

comp: comparator

Vsns: Secondary side switch drain voltage

S: set 1 end

Vth: reverse threshold

R: set to 0

2us LEB: Timer module

Q: output

SR_gate: Secondary side switch gate signal

Claims (4)

A quasi-resonant power supply controller includes: a primary side control module configured to receive the rectified AC input voltage and control the turn-on and turn-off of the gate. When the primary side switch is turned off, the secondary side switch is turned on until it is sensed Turn off the secondary side switch after the demagnetization is over, wherein the primary side switch is connected in parallel with the primary side switch parasitic capacitance, and the switch is turned on when the voltage on the primary side switch parasitic capacitance resonates to a minimum value; and the secondary side control module; The secondary side control module includes a control and sampling circuit, and is configured to sense the input voltage to set the reverse threshold when the secondary side switch is turned off, and the secondary side control module further includes one or more external adjustment resistors. , The one or more external adjustment resistors are connected to the control and sampling circuit. For the quasi-resonant power supply controller described in item 1 of the scope of patent application, the control and sampling circuit sets the reverse threshold when the secondary side switch is turned off as follows: Vth=I ₁ . In Rdson, Vth represents the reverse threshold when the secondary side switch is turned off, Rdson represents the on-resistance of the secondary side switch, and I ₁ represents the secondary side reverse current required to realize the zero-voltage conduction of the primary side switch. The quasi-resonant power supply controller described in item 1 of the scope of patent application, wherein the control and sampling circuit senses the input voltage by sensing the secondary side switch drain extreme voltage Vsns, Vsns=V _in /N-Vo where V _in represents the input voltage and the output voltage Vo and the transformer turns ratio N is fixed. According to the quasi-resonant power supply controller described in item 2 of the scope of patent application, the resistance value of the one or more external adjustment resistors is determined according to the transformer turns ratio N, the secondary side inductance L _s, and the primary side switch parasitic capacitance value.

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