TWI574275B - Improved sensing circuits for use in low power nanometer flash memory devices - Google Patents

Description

Improved sensing circuit for use in low power nano flash memory devices

An improved sensing circuit for use in a low power nano flash memory device is disclosed.

The use of floating gates to store flash memory cells in which they are charged, as well as memory arrays in which such non-volatile memory cells are formed in a semiconductor substrate, is well known in the art. In general, such floating gate memory cells have traditionally been of the split gate type or stacked gate type.

FIG. 1 shows a conventional non-volatile memory unit 10. The split gate ultra fast flash (SF) memory cell 10 includes a semiconductor substrate 1 which is of a first conductivity type, such as a P type. The substrate 1 has a surface on which a first region 2 (also known as a source line (SL)) is formed, which is of a second conductivity type, such as an N-type. A second region 3 (also known as a drain line) is formed on the surface of the substrate 1, which is also of a second conductivity type, such as an N-type. A channel region 4 is formed between the first region 2 and the second region 3. A bit line (BL) 9 is connected to the second area 3. A word line (WL) 8 (also referred to as a select gate) is located on and insulated from the first portion of the channel region 4. The word line 8 has little or no weight with the second region 3 Stack. A floating gate (FG) 5 is attached to another portion of the passage region 4. The floating gate 5 is insulated from it and adjacent to the word line 8. The float gate 5 is also adjacent to the first region 2. A coupling gate (CG) 7 (also known as a control gate) is attached to and insulated from the float gate 5. A wiper gate (EG) 6 is above the first region 2 and is adjacent to and insulated from the float gate 5 and the coupling gate 7. The wiper gate 6 is also insulated from the first region 2.

Exemplary operations of erasing and stylizing conventional non-volatile memory cells 10 are described below. The unit 10 is erased by applying a high voltage to the erase gate (EG) 6 while the other terminals are equal to zero volts through the Fowler-Nordheim tunneling mechanism. The electrons tunneling from the floating gate (FG) 5 to the erase gate (EG) 6 cause the floating gate (FG) 5 to be positively charged, causing the unit 10 to be in an ON state in the reading condition. The resulting cell erase state is known to be "1" state. Another embodiment for erasing is achieved by applying a positive voltage Vegp on the erase gate (EG) 6, applying a negative voltage Vcgn to the coupling gate (CG) 7, and making the other terminals equal to zero volts. The negative voltage Vcgn is negatively coupled to the floating gate (FG) 5, so that the positive voltage Vcgp required for erasing is small. The electrons tunneling from the floating gate (FG) 5 to the erase gate (EG) 6 cause the floating gate (FG) 5 to be positively charged, so that the unit 10 is in the reading condition (unit state "1"). On state. Alternatively, the word line (WL) 8 (Vwle) and the source line (SL) 2 (Vsle) may be negative to further reduce the positive voltage required for erasing the erase gate (FG) 5. . In this example, the magnitudes of the negative voltages Vwle and Vsle are small enough to make the p/n junctions biased. Transmitting a high voltage on the source gate (SL) 2 and applying the erase gate (EG) 6 by applying a high voltage to the coupling gate (CG) 7 through a source-side thermal electron stylization mechanism The unit 10 is programmed by applying a stylized current to the voltage and to the bit line (BL) 9. A portion of the electrons flowing through the gap between the word line (WL) 8 and the floating gate (FG) 5 obtains sufficient energy for injection To the floating gate (FG) 5, the floating gate (FG) 5 is negatively charged, causing the unit 10 to be in an off state in the read condition. The resulting unit stylized state is known to be "0" state.

The cell 10 can be suppressed during the stylization by applying a suppression voltage on the bit line (BL) 9 (e.g., if another cell in the cell column is to be programmed but the cell 10 is not to be programmed). For a description of the operation of the discrete gate flash memory and various circuit systems, see U.S. Patent No. 7,990,773 (named "Sub Volt Flash Memory System") by Hieu Van Tran et al., and U.S. Patent No. 8,072,815 by Hieu Van Tran et al. (The name is "Array of Non-Volatile Memory Cells Including Embedded Local and Global Reference Cells and Systems"), both of which are incorporated herein by reference.

2 depicts the architecture of a two-dimensional prior art flash memory system of a typical prior art. The die 12 includes: a memory array 15 for storing data and a memory array 20, the memory array optionally utilizing the memory cell 10 of FIG. 1; for other components in the die 12 and (generally a pad 35 and a pad 80 enabling electrical communication between the wire bonds (not shown), the wire bond being in turn connected to a pin (not shown) or a package bump for receiving the integrated circuit from outside the package wafer a high voltage circuit 75 for providing positive and negative voltage supply to the system; control logic 70 for providing various control functions such as redundancy and built-in self-test; analog logic 65; for self-memory The body array 15 and the memory array 20 read the data sensing circuits 60 and 61; the column decoder circuit 45 for accessing the column to be read or to be written in the memory array 15 and the memory array 20, respectively. a row decoder circuit 46; a row decoder 55 and a row decoder 56 for respectively accessing the row to be read or to be written in the memory array 15 and the memory array 20; The programming and erasing operations of the array 15 and the memory array 20 provide a boosted voltage charge pump circuit 50 and a charge pump circuit 51; the memory array 15 and the memory array 20 are shared for reading and writing (erasing/staging) Operating high voltage driver circuit 30; high voltage driver circuit 25 used by memory array 15 during read and write operations, and high voltage driver used by memory array 20 during read and write (erase/program) operations The circuit 26; and the bit line suppression voltage circuit 40 and the bit line suppression voltage circuit 41 for canceling the selection of the bit lines that do not need to be programmed during the writing operation of the memory array 15 and the memory array 20, respectively. Those functional blocks should be understood by those of ordinary skill in the art, and the block layout shown in Figure 2 is known in the prior art.

FIG. 3 depicts a conventional sensing circuit 100. Sensing circuit 100 is an example of the type of circuit that can be used as sensing circuits 60 and 61 of FIG. The sensing circuit 100 includes a memory data read block 110, a memory reference read block 120, and a differential amplifier block 130.

In the present example, the memory data reading block 110 includes a current source 111, a stacked sensing NMOS transistor 113, a bit line-embedded NMOS transistor 114, and a diode-connected sensing load PMOS transistor 112. .

In the present example, the memory reference read block 120 includes a current source 121, a reference bit line clamp NMOS transistor 124, a stacked sense NMOS transistor 123, and a diode-connected sense load PMOS transistor. 122.

In the present example, the differential amplifier block 130 includes input differential pair NMOS transistors 131 and 134, current mirror load PMOS transistors 132 and 133, and input. A PMOS transistor 135, a bias current NMOS transistor 136, an output bias NMOS transistor 137, and an output 140 are provided.

Node 116 is coupled to the selected memory unit (not shown) to be read, and node 117 is coupled to the reference memory unit (not shown) for determining the value of the selected memory unit, or (eg, A non-memory cell reference bias (eg, from a replica bias) from a bandgap or other reference circuit that adequately compensates for design or process environmental errors is used to determine the value of the selected memory cell.

The differential amplifier block 130 is for comparing signals received from the memory data read block 110 and the memory reference read block 120 for generating an output 140 indicating the value of the data stored in the selected memory unit. . These components are interconnected as shown in FIG.

During operation, the differential amplifier block 130 compares the current drawn by the memory data read block 110 (transmitted through node 116) with the current drawn by the memory reference read block 120 (transmitted through node 117) to produce an output 140. If the current drawn by the memory data read block 110 exceeds the reference current drawn from the memory reference read block 120 (meaning "1" is stored in the selected memory cell), the output 140 will be high. . If the current drawn from the memory data read block 110 is less than the current drawn from the memory reference read block 120 (meaning that "0" is stored in the selected memory cell), the output 140 will be low.

Sensing circuit 100 typically requires an operating voltage of 1.8 volts to 3.3 volts. As the size of flash memory cells and arrays shrinks, what is needed is an improvement in sensing circuit 100 that can be implemented with lower operating voltages (e.g., 1.1 volts) and lower power consumption. use. What is further needed is a sensing circuit that compensates for non-idealities such as transistor mismatch and memory array mismatch.

Several embodiments described herein provide a lower power, lower voltage sensing circuit. These embodiments use various techniques to compensate for non-idealities such as transistor mismatch and memory array mismatch.

1‧‧‧Semiconductor substrate

2‧‧‧First area

3‧‧‧Second area

4‧‧‧Channel area

5‧‧‧Float

6‧‧‧Erase the gate

7‧‧‧coupled gate

8‧‧‧ word line

9‧‧‧ bit line

10‧‧‧Non-volatile memory unit

12‧‧‧ grain

15‧‧‧Memory array

20‧‧‧ memory array

25‧‧‧High voltage driver circuit

26‧‧‧High voltage driver circuit

30‧‧‧High voltage driver circuit

35‧‧‧ pads

40‧‧‧ bit line suppression voltage circuit

41‧‧‧ bit line suppression voltage circuit

45‧‧‧ column decoder circuit

46‧‧‧ column decoder circuit

50‧‧‧Charge pump circuit

51‧‧‧Charge pump circuit

55‧‧‧ line decoder

56‧‧‧ line decoder

60‧‧‧Sensor circuit

61‧‧‧Sensor circuit

65‧‧‧ analog logic

70‧‧‧Control logic

75‧‧‧High voltage circuit

80‧‧‧ pads

100‧‧‧Sensor circuit

110‧‧‧Memory data reading block

111‧‧‧current source

112‧‧‧Diode-connected sensing load PMOS transistor

113‧‧‧Stacked sensing NMOS transistor

114‧‧‧ bit line clamp NMOS transistor

116‧‧‧ nodes

117‧‧‧ nodes

120‧‧‧Memory Reference Read Block

121‧‧‧current source

122‧‧‧Diode-connected sensing load PMOS transistor

123‧‧‧Stacked sensing NMOS transistor

124‧‧‧Reference bit line clamp NMOS transistor

130‧‧‧Differential Amplifier Block

131‧‧‧Differential to NMOS transistor

132‧‧‧current mirror load PMOS transistor

133‧‧‧current mirror load PMOS transistor

134‧‧‧Differential to NMOS transistor

135‧‧‧ Output PMOS transistor

136‧‧‧Non-current NMOS transistor

137‧‧‧Output biased NMOS transistor

140‧‧‧ Output

200‧‧‧Sensor circuit

210‧‧‧Memory data reading block

212‧‧‧Optoelectronics

216‧‧‧ nodes

220‧‧‧Memory Reference Read Block

222‧‧‧Diode-connected transistor

230‧‧‧Differential Amplifier Block

240‧‧‧ output

242‧‧‧Optoelectronics

244‧‧‧Optoelectronics

246‧‧‧Optoelectronics

250‧‧‧ switch

252‧‧‧Sensing signal

260‧‧‧ capacitor

262‧‧‧Input NMOS transistor pair

264‧‧‧ Deviation NMOS transistor

270‧‧‧ capacitor

272‧‧‧Input NMOS transistor pair

274‧‧‧Optoelectronics

278‧‧‧ Cross-coupled inverter pair

280‧‧‧ nodes

290‧‧‧ Output node

300‧‧‧Sensor circuit

310‧‧‧Memory data reading block

320‧‧‧Memory Reference Read Block

330‧‧‧Differential Amplifier Block

340‧‧‧ output

342‧‧‧Reference reading offset block

350‧‧‧ switch

352‧‧‧ signal

355‧‧‧Voltage level

360‧‧‧ capacitor

362‧‧‧Input pair

363‧‧‧Input pair of transistors

364‧‧‧ Deviation transistor

370‧‧‧ capacitor

372‧‧‧Input pair of transistors

374‧‧‧Optoelectronics

378‧‧‧ Cross-coupled inverter pair

380‧‧‧Sensor node

392‧‧‧reference node

400‧‧‧Sensor circuit

410‧‧‧Memory data reading block

411‧‧‧Stacked sensing NMOS transistor

420‧‧‧Memory Reference Read Block

42‧‧‧Stacked Sensing NMOS Transistor

430‧‧‧Differential Amplifier Block

440‧‧‧ output

442‧‧‧Reference read deviation block

450‧‧‧Switch

455‧‧‧Voltage level

460‧‧‧ capacitor

470‧‧‧ capacitor

480‧‧‧Sensor node

492‧‧‧ reference node

500‧‧‧Sensor circuit

510‧‧‧Memory data reading block

520‧‧‧Memory Reference Read Block

530‧‧‧Differential Amplifier Block

540‧‧‧ output

542‧‧‧Memory Reference Read Deviation Block

545‧‧‧NMOS transistor

550‧‧‧ switch

555‧‧‧NMOS transistor

560‧‧‧ capacitor

565‧‧‧ Cross-coupled inverter block

570‧‧‧ capacitor

580‧‧‧Sensor node

592‧‧‧ reference node

600‧‧‧Sensor circuit

610‧‧‧Memory data reading block

620‧‧‧Memory Reference Read Block

630‧‧‧Differential Amplifier Block

640‧‧‧ output

650‧‧‧ switch

660‧‧‧ capacitor

670‧‧‧ capacitor

680‧‧‧Sensor node

690‧‧‧reference node

700‧‧‧Sensor circuit

710‧‧‧Memory data reading block

720‧‧‧Memory Reference Read Block

730‧‧‧Differential Amplifier Block

740‧‧‧ Output

750‧‧‧ switch

752‧‧‧ switch

760‧‧‧ capacitor

762‧‧‧Input pair

770‧‧‧ capacitor

772‧‧‧Input pair

780‧‧‧Sensor node

790‧‧‧ reference node

800‧‧‧Sensor circuit

810‧‧‧Memory data reading block

820‧‧‧Memory Reference Read Block

830‧‧‧Differential Amplifier Block

840‧‧‧ output

850‧‧‧ switch

852‧‧‧Switch

860‧‧‧ capacitor

862‧‧‧Input pair

870‧‧‧ capacitor

872‧‧‧Input pair

880‧‧‧Sensor node

890‧‧‧ reference node

900‧‧‧Sensor circuit

910‧‧‧Memory data reading block

916‧‧‧Memory unit current

917‧‧‧Memory unit current

920‧‧‧Memory Reference Read Block

930‧‧‧Differential Amplifier Block

940‧‧‧ output

950‧‧‧ signal

954‧‧‧ switch

955‧‧‧Read reference level

956‧‧‧sum node

960‧‧‧ capacitor

970‧‧‧ capacitor

980‧‧‧Sensor node

988‧‧‧ comparator

990‧‧‧ reference node

1000‧‧‧ comparator circuit

1010‧‧‧ NMOS transistor

1020‧‧‧NMOS transistor

1030‧‧‧ Cross-coupled inverter to NMOS

1032‧‧‧ Cross-coupled inverter to PMOS

1040‧‧‧ Cross-coupled inverter to NMOS

1042‧‧‧ Cross-coupled inverter to PMOS

1050‧‧‧ PMOS transistor

1060‧‧‧Enable signal

1070‧‧‧Enable signal

1100‧‧‧ Comparator circuit

1110‧‧‧Input pair NMOS

1120‧‧‧Input to NMOS

1130‧‧‧NMOS

1132‧‧‧ bias current

1142‧‧‧cross-coupled to PMOS

1152‧‧‧cross-coupled to PMOS

1160‧‧‧Optoelectronics

1170‧‧‧Optoelectronics

1180‧‧ Enter

1190‧‧‧Enter

1200‧‧‧ Comparator circuit

1210‧‧‧Input NMOS pair

1220‧‧‧Input NMOS pair

1230‧‧‧NMOS

1240‧‧‧ Cross-coupled inverter to NMOS

1242‧‧‧ Cross-coupled inverter to PMOS

1244‧‧‧ output

1250‧‧‧ Cross-coupled inverter to NMOS

1252‧‧‧ Cross-coupled inverter to PMOS

1254‧‧‧ output

1260‧‧‧Optoelectronics

1261‧‧‧Optoelectronics

1270‧‧•Transistor

1271‧‧‧Optoelectronics

1280‧‧‧Enter

1290‧‧‧ Input

1300‧‧‧ Comparator circuit

1310‧‧‧ Cross-coupled inverter to NMOS

1312‧‧‧ Cross-coupled inverter to PMOS

1320‧‧‧ Cross-coupled inverter to NMOS

1322‧‧‧ Cross-coupled inverter to PMOS

1332‧‧‧Switch

1360‧‧‧Switch

1400‧‧‧ Comparator Circuit

1410‧‧‧Inverter NMOS

1412‧‧‧Inverter PMOS

1420‧‧‧Switch

1500‧‧‧Sensor circuit

1510‧‧‧Memory reading block

1516‧‧‧Memory unit current

1527‧‧‧Reference memory cell current

1530‧‧‧Differential Amplifier Block

1540‧‧‧ output

1550‧‧‧Switch

1554‧‧‧Switch

1555‧‧‧Read reference deviation level

1556‧‧‧ nodes

1560‧‧‧ capacitor

1580‧‧‧Sensor node

1588‧‧‧ Comparator

2010‧‧‧Signal PRECH

2020‧‧‧Signal SEN

2030‧‧‧Signal LATCH

2040‧‧‧Signal BL

2050‧‧‧Signal WL

2060‧‧‧Signal SOUT

Figure 1 depicts a conventional split gate flash memory cell.

Figure 2 depicts the layout of a conventional flash memory array.

Figure 3 depicts a conventional sensing circuit for use with a flash memory array.

4 depicts a first embodiment of a sensing circuit for use with a flash memory array.

Figure 5 depicts a second embodiment of a sensing circuit for use with a flash memory array.

Figure 6 depicts a third embodiment of a sensing circuit for use with a flash memory array.

Figure 7 depicts a fourth embodiment of a sensing circuit for use with a flash memory array.

Figure 8 depicts a fifth embodiment of a sensing circuit for use with a flash memory array.

Figure 9 depicts a sixth embodiment of a sensing circuit for use with a flash memory array.

Figure 10 depicts a seventh embodiment of a sensing circuit for use with a flash memory array.

Figure 11 depicts a seventh embodiment of a sensing circuit for use with a flash memory array.

Figure 12 depicts another embodiment of a comparator circuit for use with a flash memory array.

Figure 13 depicts another embodiment of a comparator circuit for use with a flash memory array.

Figure 14 depicts another embodiment of a comparator circuit for use with a flash memory array.

Figure 15 depicts another embodiment of a comparator circuit for use with a flash memory array.

Figure 16 depicts another embodiment of a comparator circuit for use with a flash memory array.

Figure 17 depicts another embodiment of a sensing circuit for use with a flash memory array.

Figure 18 depicts an embodiment of a sensing sequence for use with a flash memory array.

Referring to Figure 4, an embodiment is depicted. The sensing circuit 200 compensates for the transistor mismatch phenomenon and the array mismatch phenomenon. The sensing circuit 200 includes a memory data reading block 210, a memory reference reading block 220, and a differential amplifier block 230. The memory data reading block 210 has a number of components identical to those of the memory data reading block 110, and such components are not described herein. Similarly, the memory reference read block 220 has a number of components identical to those of the memory reference read block 120, which are not described herein. The memory differential amplifier block 230 includes input NMOS transistor pairs 262 and 272 that are biased by a bias NMOS transistor 264. The gates of the input NMOS transistor pairs 262 and 272 are connected to the terminals of capacitors 260 and 270, respectively. The amplifier block 230 also includes a cross-coupled inverter pair 278 whose source (virtual ground) is biased by the NMOS 264. The output of the pair of cross-coupled inverters is coupled to the drains of the input transistors 262 and 272. The memory differential amplifier block 230 includes an output stage comprised of transistors 242, 244, and 246. The drain of transistor 274 is coupled to transistor 264 drain and its gate is enabled by sense signal (P2) 252. These devices are connected as shown in FIG. Instead of the diode connection, the transistor 212 of the memory data read block 210 mirrors the reference current from the transistor 222 of the memory reference read block 220 and is compared to the data current coupled to the transmit node 216. . The comparison result is output on the node 280.

Unlike the prior art, the differential amplifier block 230 is decoupled from the memory data read block 210 and the memory reference read block 220. In particular, the input of the differential amplifier block 230 is coupled to a capacitor 260, which in turn is coupled to a memory data read block 210, specifically to the output node 280, and the difference Another input of the operational amplifier block 230 is coupled to a capacitor 270, which in turn is coupled to a memory reference read block 220, and in particular to an output node 290 of the diode-connected transistor 222. This enables the system to pre-charge the differential amplifier block 230 independently of the memory data read block 210 and the memory reference read block 220. Capacitors 260 have an exemplary value of 5fF to 80fF, and capacitor 270 has an exemplary value of 5fF to 80fF.

During the pre-charge phase, switch 250 is turned "on" prior to sensing the comparison operation. This ensures that portions of the differential amplifier block 230 coupled to the capacitor 260 are charged to the same voltage level as the portion of the differential amplifier block 230 coupled to the capacitor 270. This can be seen as the action of preamplifying the differential amplifier block 230. This also effectively acts to automatically zero (eliminate) the bias of the differential amplifier block 230, i.e., it stores the offset on the capacitor during the precharge phase and is eliminated during the sensing phase.

During the sensing phase, switch 250 is open and thus signal 252 will be turned "on". If the selected memory cell stores "0", the voltage at the sensing node 280 will rise, and if the selected memory cell stores "1", it will drop. The reference node 290 maintains a voltage level that is approximately between the high level of the sense node 280 and the low level of the sense node 280. The differential amplifier 230 then compares the sense node 280 with the reference node 290 through its voltage coupling through capacitors 260 and 270, respectively, and the result is revealed at output 240. If the selected memory unit stores "0", the output 240 will be low. If the selected memory unit stores "1", the output 240 will be high. The cross-coupled inverter pair 278 is used to be fed back during the sensing phase to speed up the sensing time. Transistor 274 is used to increase the sensing time by increasing the tail bias current of the transistor 264 in parallel, and is also used to provide a gnd (~0v) level to the cross-coupled inverter pair 278, and thus provide its output.

An advantage of this embodiment is that the transistor mismatch is mitigated by using the common initial state established by the differential amplifier 230 through the precharge phase and the decoupling of the capacitor 260 and capacitor 270. In addition, the decoupling allows the memory data read block 210 to use a higher bit line current than would be possible without decoupling. Compared to the power supply of the memory data read block 210 and the memory reference read block 220, the power supply of the differential amplifier block 230 may be further optimized (or decoupled) at different levels.

Referring to Figure 5, an embodiment is depicted. The sensing circuit 300 compensates for transistor mismatch and array mismatch. The sensing circuit 300 includes a memory data read block 310, a memory reference read block 320, and a differential amplifier block 330. The components of the memory data reading block 310, the memory reference reading block 320, and the differential amplifier block 330 have many of the same components as those of the previous embodiment, and will not be described herein. These devices are connected as shown in FIG.

The sensing circuit 300 is similar to the sensing circuit 200. The differential amplifier block 330 is decoupled from the memory data read block 310 and the memory reference read block 320. In particular, differential amplifier block 330 is coupled to capacitor 360 (which in turn is coupled to memory data read block 310), and differential amplifier block 330 is coupled to capacitor 370 (which in turn is coupled to a reference read bias) Block 342). This enables the system to be independent of the memory data read block 310 and the memory reference read block 320, while the precharge differential amplifier Block 330. An exemplary value for capacitor 360 is 5fF to 80fF, and an exemplary value for capacitor 370 is 5fF to 80fF. The differential amplifier block 330 includes a cross-coupled inverter pair 378 (via its source (virtual ground) biased via transistor 374). The differential amplifier block 330 includes input pair transistors 363 and 372 (biased via a bias transistor 364 that is different (decoupled) from the bias transistor 374). The gates of the input pair of transistors 362 and 372 are coupled to the terminals of capacitors 360 and 370, respectively.

During the pre-charge phase, switch 350 is turned "on" before the sensing operation. This ensures that portions of the differential amplifier block 330 coupled to the capacitor 360 are charged to the same voltage level as the portion of the differential amplifier block 330 coupled to the capacitor 370. This can be seen as the action of preamplifying the differential amplifier block 330. This also effectively acts to automatically zero the offset of the amplifier block 330. The placement of switch 350 in sense circuit 300 is slightly different than the placement of switch 250 in sense circuit 200. In particular, one of the switches 350 causes the sense nodes 380 and 290 to be directly coupled to the VDD power supply. Thus, at the beginning of the sensing phase, sense node 380 is at VDD. The VDD is exemplified as 1.1 volts.

Switch 350 is turned off during the sensing phase. If the selected memory cell stores "0", the voltage at the sensing node 380 will drop. If the selected memory cell stores "1", it will drop further. The reference node 392 is switched by signal 352 to a voltage level 355 that is approximately between the high level of the sense node 380 and the low level of the sense node 380. The differential amplifier 330 then compares the sense node 380 with the reference bias node 392 through its voltage coupling through capacitors 360 and 370, respectively, and The result will be revealed at output 340. If the selected memory unit stores "0", the output 340 will be low. If the selected memory unit stores "1", the output 340 will be high.

An advantage of this embodiment is that the transistor mismatch is mitigated by using the common initial state established by the differential amplifier 330 through the precharge phase and the decoupling of the capacitor 360 and capacitor 370. Additionally, the decoupling allows the memory data read block 310 to use a higher bit line current than would be possible without decoupling. Compared to the power supply of the memory data read block 310 and the memory reference read block 320, the power supply of the differential amplifier block 330 may be further optimized (or decoupled) at different levels.

Referring to Figure 6, an embodiment is depicted. The sensing circuit 400 compensates for the transistor mismatch phenomenon and the array mismatch phenomenon. The sensing circuit 400 includes a memory data read block 410, a memory reference read block 420, and a differential amplifier block 430. The components of the memory data read block 410, the memory reference read block 420, and the differential amplifier block 430 have many of the same components as those of the previous embodiment and will not be described herein. These devices are connected as shown in FIG.

The difference between the sensing circuit 400 and the sensing circuit 300 is that the memory data reading block 410 does not have a current source (such as the current source 111 shown in FIG. 3 and the following figure), and the memory reference reading area Block 420 does not have a current source (current source 121 as shown in Figure 3 and the following figure). Rather, the gate of the spliced sense NMOS transistor 411 is coupled to the bias source of voltage VC1, and the gate of the spliced sense NMOS transistor 42 is coupled to the bias source of voltage VC2. The VC1 example value is 0.6V to 1.5V, and the VC2 is illustrated. The value is 0.6V to 1.5V. The effect of these differences is that the sensing circuit 400 consumes less power than the sensing circuit 300.

The differential amplifier block 430 is decoupled from the memory data read block 410 and the memory reference read block 420. In particular, differential amplifier block 430 is coupled to capacitor 460 (which in turn is coupled to memory data read block 410), and differential amplifier block 430 is coupled to capacitor 470 (which in turn is coupled to a reference read bias) Block 442). This enables the system to pre-charge the differential amplifier block 430 independently of the memory data read block 410 and the memory reference read block 420. An exemplary value for capacitor 460 is 5fF to 80fF, and an exemplary value for capacitor 470 is 5fF to 80fF.

During the pre-charge phase, switch 450 is turned "on" before the sensing operation. This ensures that portions of the differential amplifier block 430 coupled to the capacitor 460 are charged to the same voltage level as the portion of the differential amplifier block 430 coupled to the capacitor 470. This can be seen as the action of the preamplifier of the differential amplifier block 430. One of the switches 450 causes the sense node 480 to be directly coupled to the VDD power supply. Thus, at the beginning of the sensing phase, sense node 480 is at VDD. The VDD is exemplified as 1.1 volts.

During the sensing phase, switch 450 is turned off. If the selected memory cell stores "0", the voltage at the sensing node 480 will drop. If the selected memory cell stores "1", it will drop further. The reference node 492 will switch to a voltage level 455 that is approximately between the high level of the sense node 480 and the low level of the sense node 480. The differential amplifier 430 then compares the sense node 480 with the reference bias node 492 through its voltage coupling through capacitors 460 and 470, respectively, and the result is Appeared at output 440. If the selected memory unit stores "0", the output 440 will be low. If the selected memory unit stores "1", the output 440 will be high.

One advantage of this embodiment is that the transistor mismatch is mitigated by using the common initial state established by the pre-charge phase in the differential amplifier 430 and the decoupling of the capacitor 460 and capacitor 470. Additionally, the decoupling allows the memory data read block 410 to use a higher bit line current than would be possible without decoupling.

Referring to Figure 7, another embodiment is depicted. The sensing circuit 500 compensates for transistor mismatch and array mismatch. The sensing circuit 500 includes a memory data read block 510, a memory reference read block 520, and a differential amplifier block 530. The components of the memory data reading block 510, the memory reference reading block 520, and the differential amplifier block 530 have many of the same components as those of the previous embodiment, and will not be described herein. These devices are connected as shown in FIG.

One difference of the sensing circuit 500 compared to the sensing circuit 400 is that the source of the NMOS transistor 545 is coupled to the source of the NMOS transistor 555 and is coupled to the drain of the cross-coupled inverter block 565.

The differential amplifier block 530 is decoupled from the memory data read block 510 and the memory reference read block 520. In particular, differential amplifier block 530 is coupled to capacitor 560 (which in turn is coupled to memory data read block 510), and differential amplifier block 530 is coupled to capacitor 570 (which in turn is coupled to a memory reference read) Biasing block 542). This enables the system to read block 510 and memory reference independently of memory data. Block 520 is read and pre-charged differential amplifier block 530 is pre-charged. The illustrated value of capacitor 560 is 5fF to 80fF, and the exemplary value of capacitor 570 is 5fF to 80fF.

During the pre-charge phase, switch 550 is turned "on" before the sensing operation. This ensures that portions of the differential amplifier block 530 coupled to the capacitor 560 are charged to the same voltage level as the portion of the differential amplifier block 530 coupled to the capacitor 570. This can be seen as the action of the preamplifier of the differential amplifier block 530. One of the switches 550 causes the sense node 580 to be directly coupled to the VDD power supply. Thus, at the beginning of the sensing phase, sense node 580 is at VDD. The VDD is exemplified as 1.1 volts.

During the sensing phase, switch 550 is turned off. If the selected memory cell stores "0", the voltage at the sensing node 580 will drop. If the selected memory cell stores "1", it will drop further. The reference node 592 will switch to a voltage level 555 that is approximately between the high level of the sense node 580 and the low level of the sense node 580. The differential amplifier 530 then compares the sense node 580 with the reference node 592 through its voltage coupling through capacitors 560 and 570, respectively, and the result is revealed at output 540. If the selected memory unit stores "0", the output 540 will be low. If the selected memory unit stores "1", the output 540 will be high.

An advantage of this embodiment is that the transistor mismatch is mitigated by using the common initial state established by the differential amplifier 530 through the precharge phase and the decoupling of the capacitor 560 and capacitor 570. In addition, the decoupling allows the memory data read block 510 to use a higher bit line current than would be possible without decoupling.

Referring to Figure 8, another embodiment is depicted. The sensing circuit 600 compensates for transistor mismatch and array mismatch. The sensing circuit 600 includes a memory data read block 610, a memory reference read block 620, and a differential amplifier block 630. The components of the memory data read block 610, the memory reference read block 620, and the differential amplifier block 630 have many of the same components as those of the previous embodiment and will not be described herein. These devices are connected as shown in FIG.

The differential amplifier block 630 is decoupled from the memory data read block 610 and the memory reference read block 620. In particular, differential amplifier block 630 is coupled to capacitor 660 (which in turn is coupled to memory data read block 610), and differential amplifier block 630 is coupled to capacitor 670 (which in turn is coupled to a memory reference read) Block 620). This enables the system to pre-charge the differential amplifier block 630 independently of the memory data read block 610 and the memory reference read block 620. Capacitors 660 have an exemplary value of 5fF to 80fF, and capacitor 670 has an exemplary value of 5fF to 80fF.

During the pre-charge phase, the switch 650 is turned on prior to the sensing operation. This ensures that portions of the differential amplifier block 630 coupled to the capacitor 660 are charged to the same voltage level as the portion of the differential amplifier block 630 coupled to the capacitor 670. This can be seen as the action of preamplifying the differential amplifier block 630. One of the switches 650 causes the sense node 680 to be directly coupled to the VDD power supply. Thus, at the beginning of the sensing phase, sense node 580 is at VDD. The only difference between the sense circuit 600 and the sense circuit 500 is that one of the switches 650 directly couples the reference node 690 to VDD. Therefore, reference node 690 will also be at VDD at the beginning of the sensing phase.

During the sensing phase, the switch 650 is turned off. If the selected memory cell stores "0", the voltage at the sensing node 680 will drop, and if the selected memory cell stores "1", it will speed up and further decrease. The differential amplifier 630 then compares the sense node 680 with the reference node 690 during ramp down of nodes 680 and 690 and will reveal the result at output 640. If the selected memory unit stores "0", the output 640 will be low. If the selected memory unit stores "1", the output 640 will be high. The reference node 690 ramps down to a steady state voltage level that is approximately between the high level of the sense node 680 and the low level of the sense node 680, with appropriate currents on nodes 680 and 690 or Resistive load.

An advantage of this embodiment is that the transistor mismatch is mitigated by using the common initial state established by the differential amplifier 630 through the precharge phase and the decoupling of the capacitor 660 and capacitor 670. In addition, the decoupling allows the memory data read block 610 to use a higher bit line current than would be possible without decoupling.

Referring to Figure 9, another embodiment is depicted. The sensing circuit 700 compensates for transistor mismatch and array mismatch. The sensing circuit 700 includes a memory data read block 710, a memory reference read block 720, and a differential amplifier block 730. The components of the memory data read block 710, the memory reference read block 720, and the differential amplifier block 730 have many of the same components as those of the previous embodiment, and will not be described herein. These devices are connected as shown in FIG.

The differential amplifier block 730 is decoupled from the memory data read block 710 and the memory reference read block 720. Specifically, the differential amplifier block 730 is connected to Capacitor 760 (which in turn is coupled to memory data read block 710) and differential amplifier block 730 is coupled to capacitor 770 (which in turn is coupled to memory reference read block 720). This enables the system to pre-charge the differential amplifier block 730 independently of the memory data read block 710 and the memory reference read block 720. An exemplary value for capacitor 760 is 5fF to 80fF, and an exemplary value for capacitor 770 is 5fF to 80fF.

During the pre-charge phase, switch 750 is turned "on" before the sensing operation. This ensures that portions of the differential amplifier block 730 coupled to the capacitor 760 are charged to the same voltage level as the portion of the differential amplifier block 730 coupled to the capacitor 770. This can be seen as the action of pre-charging or initializing the differential amplifier block 730. Input pairs 762 and 772 are respectively coupled to the drains of capacitors 760 and 770 to precharge to the VDD level.

During the sensing phase, switch 750 is open and switch 752 is turned "on". If the selected memory cell stores "0", the voltage at the sensing node 780 will drop. If the selected memory cell stores "1", it will drop further. The reference node 790 will be at a voltage level that is approximately between the high level of the sense node 780 and the low level of the sense node 780. The differential amplifier 730 then compares the sense node 780 with the reference node 790 through its voltage coupling through capacitors 760 and 770, respectively, and the result is revealed at output 740. If the selected memory unit stores "0", the output 740 will be low. If the selected memory unit stores "1", the output 740 will be high.

An advantage of this embodiment is that the transistor mismatch is mitigated by using a common initial state established in the differential amplifier 730 through the pre-charge phase and decoupling of the capacitor 760 and capacitor 770. In addition, compared to no decoupling As may be the case, the decoupling allows the memory data read block 710 to use a higher bit line current.

Referring to Figure 10, another embodiment is depicted. The sensing circuit 800 compensates for transistor mismatch and array mismatch. The sensing circuit 800 includes a memory data read block 810, a memory reference read block 820, and a differential amplifier block 830. The components of the memory data read block 810, the memory reference read block 820, and the differential amplifier block 830 have many of the same components as those of the previous embodiment, and will not be described herein. These devices are connected as shown in FIG.

The differential amplifier block 830 is decoupled from the memory data read block 810 and the memory reference read block 820. In particular, differential amplifier block 830 is coupled to capacitor 860 (which in turn is coupled to memory data read block 810), and differential amplifier block 830 is coupled to capacitor 870 (which in turn is coupled to a memory reference read) Block 820). This enables the system to pre-charge the differential amplifier block 830 independently of the memory data read block 810 and the memory reference read block 820. An exemplary value for capacitor 860 is 5fF to 80fF, and an exemplary value for capacitor 870 is 5fF to 80fF. Input pairs 862 and 872 are coupled to the drains of capacitors 860 and 870, respectively, to precharge to the VDD level.

During the pre-charge phase, switch 850 is turned "on" before the sensing operation. This ensures that portions of the differential amplifier block 830 coupled to the capacitor 860 are charged to the same voltage level as the portion of the differential amplifier block 830 coupled to the capacitor 870. This can be seen as the action of the differential amplifier block 830 initialization. One of the switches 850 connects the sense node 880 to VDD, and the other of the switches 850 connects the reference node 890 to VDD. Therefore, both sense node 880 and reference node 890 are at the voltage level of VDD at the beginning of the sensing phase. The VDD is exemplified as 1.1 volts.

During the sensing phase, switch 850 is open and switch 852 is turned "on". If the selected memory cell stores "0", the voltage at the sensing node 880 will drop. If the selected memory cell stores "1", it will drop further. The reference node 890 will ramp down from VDD to a voltage level that is approximately between the high level of the sense node 880 and the low level of the sense node 880. The differential amplifier 830 then compares the sense node 880 and the reference node 890 through capacitors 860 and 870, respectively, and the result is revealed at output 840. If the selected memory unit stores "0", the output 840 will be low. If the selected memory unit stores "1", the output 840 will be high.

An advantage of this embodiment is that the transistor mismatch is mitigated by using a common initial state established in the differential amplifier 830 through the pre-charge phase and decoupling of the capacitor 860 and capacitor 870. Additionally, the decoupling allows memory data read block 810 to use a higher bit line current than would be possible without decoupling.

Referring to Figure 11, another embodiment is depicted. The sensing circuit 900 compensates for transistor mismatch and array mismatch. The sensing circuit 900 includes a memory data read block 910, a memory reference read block 920, and a differential amplifier block 930. These devices are connected as shown in FIG. The memory data read block 910 includes a sense node 980 coupled to a switch controlled by signal 950 (to VDD) and to a memory cell current 916 (to ground). The memory reference read block 920 includes a reference node 990 coupled to the signal The 950 controlled switch (to ground) and coupled to the memory unit current 917 (to VDD).

The differential amplifier block 930 is decoupled from the memory data read block 910 and the memory reference read block 920. In particular, differential amplifier block 930 is coupled to capacitor 960 (which in turn is coupled to memory data read block 910), and differential amplifier block 930 is coupled to capacitor 970 (which in turn is coupled to a memory reference read) Block 920). This enables the system to operate the differential amplifier block 930 at a bias level independent of the memory data read block 910 and the memory reference read block 920. The differential amplifier block 930 includes a comparator 988 having its output 940, and a switch 954 to initialize the comparator 988. One of the terminals of the differential amplifier block 930 is coupled to the terminals of capacitors 960 and 970 at the same time. The other terminal of the differential amplifier block 930 is coupled to a read reference level 955.

During the pre-charge phase, switches 950 and 954 are turned on prior to the sensing operation. This ensures that portions of the differential amplifier block 930 coupled to the capacitor 960 are charged to the same voltage level as the portion of the differential amplifier block 930 coupled to the capacitor 970. This can be seen as the action of the differential amplifier block 930 initialization/auto-zeroing. One of the switches 850 connects the sense node 980 to VDD, while the other of the switches 850 connects the reference node 990 to GND. Therefore, the sensing node 980 and the reference node 990 are at complementary voltage levels of VDD and GND, respectively, at the beginning of the sensing phase.

During the sensing phase, switches 950 and 954 are turned off. If the selected memory cell stores "0", the voltage at the sensing node 980 will slowly drop, and if the selected memory cell stores "1", it will speed up and further decrease. Reference node 990 will The ramp rate is ramped from GND at a ramp rate that is approximately between the high ramp rate level of sense node 880 and the low ramp rate level of sense node 980. The differential amplifier 930 then compares the sum of the sense node 980 and the reference node 990 through the capacitors 960 and 970 at the sum node 956 and the reference bias node 955, respectively, and the result is revealed at output 940. If the selected memory unit stores "0", the output 940 will be low. If the selected memory unit stores "1", the output 940 will be high.

An advantage of this embodiment is that the transistor mismatch is mitigated by using the common initial state established by the pre-charge phase in the differential amplifier 930 and the decoupling of the capacitance caused by the capacitor 960 and capacitor 970. Additionally, the decoupling allows memory read blocks 910 and 920 to use higher bit line currents than would be possible without decoupling.

Referring to Figure 12, another embodiment of a comparator is depicted. Comparator circuit 1000 includes cross-coupled inverter pair NMOS 1030/PMOS 1032 and NMOS 1040/PMOS 1042, which are enabled by PMOS transistor 1050 to provide high power to VDD by enabling signal 1070. Comparator circuit 1000 includes input NMOS pairs 1010 and 1020 having respective Zener gates 1060 and 1070, and respective drains connected to the outputs of inverters 1030/1032 and 1040/1042, respectively. Signals to the gates of NMOS transistors 1010 and 1020 are provided by, for example, the front map sensing node and the reference node. The transistor 1050 is enabled during the sensing phase. The cross-coupled inverter pair provides full VDD and GND levels at the output.

Referring to Figure 13, another embodiment of a comparator is depicted. Comparator circuit 1100 includes cross-coupling pair PMOS 1142 and PMOS 1152 (the source of which is connected to VDD). Comparator circuit 1100 includes input pairs NMOS 1110 and NMOS 1120 (with inputs 1180 and 1190 on their gates, respectively). The input couples the drain of the NMOS 1110 and the NMOS 1120, respectively (the output of the comparator 1100) to the drain of the cross-coupled pair PMOS 1142 and PMOS 1152. The input is coupled to the bias current 1132 through the NMOS 1130 to the source of the NMOS 1110 and NMOS 1120. The input signals to the gates of NMOS transistors 1010 and 1020 are provided by, for example, the front image sensing node and the reference node. The transistors 1160 and 1170 are precharged to output to VDD and are turned off during the sensing phase. The cross-coupled PMOS pair 1142/1152 provides a full VDD level at the output.

Referring to Figure 14, another embodiment of a comparator is depicted. Comparator circuit 1200 includes cross-coupled inverter pair NMOS 1240/PMOS 1242 and NMOS 1250/PMOS 1252 (which are connected to VDD when powered high). Comparator circuit 1200 includes input NMOS pairs 1210 and 1220 (with inputs 1280 and 1290 on their gates, respectively). Input NMOS pairs 1210 and 1220 have their drains coupled to the sources of the cross-coupled pair NMOS 1240/1250, respectively. The source of input NMOS pair 1210 and 1220 is coupled to GND through NMOS 1230. The outputs 1244 and 1254 of the cross-coupled inverter pair 1240/1242 and 1250/1252 are the outputs of the comparator 1200. Input signals to the gates of NMOS transistors 1210 and 1220 are provided by, for example, the front image sensing node and the reference node. Transistors 1260 and 1270 are precharged to VDD and are turned off during the sensing phase. Transistors 1261 and 1271 precharge the drains of the input pair 1210 and 1220 to VDD and are turned off during the sensing phase. The cross-coupled inverter pair 1240/1242 and 1250/1252 provide a full VDD/GND level at the output.

Referring to Figure 15, another embodiment of a comparator is depicted. The comparator circuit 1300 includes a cross-coupled inverter pair NMOS 1310/PMOS 1312 and an NMOS 1320/PMOS 1322 that are connected to VDD when high power is supplied and connected to GND when low power is supplied, and the second inverter 1310/ The output of 1312 is coupled through switch 1332 to the input of inverter 1320/1322. It includes a switch 1360 to equalize the input and output of the inverter. During pre-charge, switch 1360 is turned "on" and switch 1332 is turned "off", and during sensing, switch 1360 is turned off and switch 1332 is turned "on" to establish a positive feedback path to speed up the sensing.

Referring to Figure 16, another embodiment of a comparator is depicted. Comparator circuit 1400 includes an inverter NMOS 1410 / PMOS 1412 that is coupled to VDD when powered high and to GND when powered low. It includes a switch 1420 to equalize the input and output of the inverter. Switch 1420 is turned "on" during pre-charging for equalization, and switch 1420 is turned off for amplification during sensing.

Referring to Figure 16, another sensing embodiment is depicted. The sensing circuit 1500 compensates for transistor mismatch and array mismatch. The sensing circuit 1500 includes a memory read block 1510 and a differential amplifier block 1530. These devices are connected as shown in FIG. The memory read block 1510 includes a sense node 1580 that is coupled to VDD through a reference memory cell current 1527 and to a memory cell current 1516 (to ground).

The differential amplifier block 1530 is decoupled from the memory read block 1510. In particular, differential amplifier block 1530 is coupled to capacitor 1560 (which in turn is coupled to memory read block 1510). This enables the system to read block 1510 independently of the memory, The bias amplifier block operates the bias amplifier block 1530. The differential amplifier block 1530 includes a comparator 1588 having its output 1540 and a switch 1554 for initializing the comparator 1588. One of the terminals of the differential amplifier block 1530 is coupled to the terminal of the capacitor 1560. The other terminal of the differential amplifier block 1530 is coupled to a read reference bias level 1555.

During the pre-charge phase, switches 1550 and 1554 are turned on prior to the sensing operation. This ensures that the portion of the differential amplifier block 1530 coupled to the capacitor 1560 is charged to the same bias level as the sense node 1580. This can be seen as the action of the differential amplifier block 1530 initialization/auto-zeroing.

During the sensing phase, switches 1550 and 1554 are opened. If the selected memory cell stores "0", the voltage at the sensing node 1580 will rise slowly, and if the selected memory cell stores "1", it will speed up and further decrease. The differential amplifier 1530 then compares the sense node 1580 coupled to the node 1556 through the capacitor 1560 with the reference bias node 955 and will reveal the result at the output 1540. If the selected memory unit stores "0", the output 1540 will be low. If the selected memory unit stores "1", the output 1540 will be high.

One advantage of this embodiment is that the transistor mismatch is mitigated by using a common initial state established in the differential amplifier 1530 through the precharge phase and decoupling of the energy induced by the capacitor 1560. Additionally, the decoupling allows the memory read block 1510 to use a higher bit line current than would be possible without decoupling. Comparator 1588 can be implemented as a single ended comparator instead of a differential comparator configuration.

Referring to Figure 18, one embodiment of a sensing sequence is depicted. The signal PRECH 2010 is used for pre-charging and equalization. Signal SEN 2020 is used for the sensing phase. Signal LATCH 2030 is used to latch the sense output. Signal BL 2040 is the bit line waveform of the selected memory cell, which is shown to be equal to ~VDD during the precharge period, and is shown to be stable down to a level in the sensing phase, where the high/low level and the high/low slope rate are respectively Depends on the erase or stylized state. Signal WL 2050 is the word line waveform for the selected memory cell, which is shown to be equal to 0V during precharge and is shown to be equal to a voltage level during sensing. It is shown that after pre-charging, WL 2050 is enabled (ramped) to reduce power consumption during pre-charging. After the latching phase, WL 2050 is equal to 0V. Signal SOUT 2060 is the sensed output of the sense operation, equal to 1/0 corresponding to the erase/stylization state.

An alternative embodiment is implemented in the previous figure as a single-acting amplifier instead of a differential amplifier.

A reference copy bias for replacing the reference memory current for sensing is implemented in an alternate embodiment. The reference copy bias can be implemented by band gaps, resistors, MOS devices, bipolar devices, etc. having different desired temperature coefficients and/or having different wafer characteristics and product specifications.

The citation of the present invention is not intended to limit the scope of the claims or the scope of the claims, but only to recite one or more technical features that may be covered by one or more of the claims. The above examples of materials, processes and values are for illustrative purposes only and should not be construed as limiting the scope of the claims. It should be noted that, as used herein, the terms "over" and "on" inclusively include "directly on" (no material, component or space in the middle) Set in between) and "indirectly on" (with centered materials, components or intervals set between them). Similarly, the term "adjacent" includes "directly adjacent" (without any intermediate materials, elements or spaces between them) and "indirectly adjacent" (with intermediate materials, elements or spaces between them). For example, forming an element "on top of a substrate" can include the formation of elements directly on the substrate without the presence of materials/components in between, and the indirect formation of elements on the substrate with one or more centered materials/components therebetween. presence.

200‧‧‧Sensor circuit

210‧‧‧Memory data reading block

212‧‧‧Optoelectronics

216‧‧‧ nodes

220‧‧‧Memory Reference Read Block

222‧‧‧Diode-connected transistor

230‧‧‧Differential Amplifier Block

240‧‧‧ output

242‧‧‧Optoelectronics

244‧‧‧Optoelectronics

246‧‧‧Optoelectronics

250‧‧‧ switch

252‧‧‧Sensing signal

260‧‧‧ capacitor

262‧‧‧Input NMOS transistor pair

264‧‧‧ Deviation NMOS transistor

270‧‧‧ capacitor

272‧‧‧Input NMOS transistor pair

274‧‧‧Optoelectronics

278‧‧‧ Cross-coupled inverter pair

280‧‧‧ nodes

290‧‧‧ Output node

Claims (26)

A sensing circuit for use in a memory device, comprising: a memory data reading block for sensing a selected memory unit; and a memory reference reading block for sensing a reference memory a body unit; a differential amplifier block, comprising: a first capacitor including a first terminal and a second terminal; a second capacitor including a first terminal and a second terminal, configured to be first before the sensing operation a precharge circuit for charging the second terminal of the capacitor and the second terminal of the second capacitor, and an output; wherein the first terminal of the first capacitor is connected to the memory data reading block, and the first The second terminal of the capacitor is coupled to the differential amplifier block, and the first terminal of the second capacitor is coupled to the memory reference read block, and the second terminal of the second capacitor is coupled to the difference An amplifier block; wherein the output of the differential amplifier block during the sensing operation indicates a value stored in the selected memory cell. The sensing circuit of claim 1, wherein the selected memory unit is a split gate flash memory unit. The sensing circuit of claim 2, wherein the reference memory unit is a split gate flash memory unit. The sensing circuit of claim 1, wherein the pre-charging circuit comprises a plurality of switches that are turned "on" before the sensing operation and are turned off during the sensing operation. The sensing circuit of claim 4, wherein one of the plurality of switches connects the sensing node of the memory data reading block to a voltage source when turned on. The sensing circuit of claim 5, wherein one of the plurality of switches, when turned on, causes the memory to reference the sensing node of the read block to be connected to the voltage source. The sensing circuit of claim 1, wherein the memory data reading block comprises a current source, a cascade sensing NMOS transistor, a bit line clamping NMOS transistor, and a diode connected sensing load PMOS Crystal. The sensing circuit of claim 7, wherein the memory reference read block comprises a current source, a reference bit line-embedded NMOS transistor, a cascade sense NMOS transistor, and a diode-connected sense load PMOS. Transistor. The sensing circuit of claim 8, wherein the differential amplifier block further comprises an input differential pair NMOS transistor, a current mirror load PMOS transistor, and an output PMOS transistor, a bias current NMOS transistor, and an output bias NMOS transistor. . A sensing circuit as claimed in claim 1, wherein the differential amplifier comprises a cross-coupled inverter pair in the differential input path. The sensing circuit of claim 1, wherein the memory reference read block supplies a copy reference bias. A method for determining a value stored in a selected memory cell, comprising: precharging a first terminal of the first capacitor and a first terminal of the second capacitor using a precharge circuit; and reading the block at the sensing node using the memory data Sensing the selected memory unit; Sensing the reference memory unit at the reference node using the memory reference read block; comparing the sense node to the reference node using a differential amplifier block, the differential amplifier block including the first capacitor, the second a capacitor, and an output, and wherein a second terminal of the first capacitor is coupled to the memory data read block and the first terminal of the first capacitor is coupled to the differential amplifier block, and the second capacitor a second terminal is coupled to the memory reference read block and the first terminal of the second capacitor is coupled to the differential amplifier block; and the selected memory cell is indicated at the output of the differential amplifier block The value stored in . The method of claim 12, wherein the selected memory unit is a split gate flash memory unit. The method of claim 13, wherein the reference memory unit is a split gate flash memory unit. The method of claim 12, wherein the pre-charge circuit comprises a plurality of switches, and wherein the pre-charging step comprises turning on the plurality of switches. The method of claim 15, wherein the pre-charging step comprises connecting the sensing node to the voltage source of the memory data reading block. The method of claim 16, wherein the pre-charging step further comprises connecting the sense node to the voltage source of the reference read block. The method of claim 12, wherein the memory data read block comprises a current source, a cascade sense NMOS transistor, a bit line clamp NMOS transistor, and a diode connected sense sense load PMOS transistor. The method of claim 18, wherein the memory reference read block comprises a current source, a reference bit line clamp NMOS transistor, a cascade sense NMOS transistor, and a diode-connected sense load PMOS transistor. . The method of claim 19, wherein the differential amplifier block further comprises an input differential pair NMOS transistor, a current mirror load PMOS transistor, an output PMOS transistor, a bias current NMOS transistor, and an output bias NMOS transistor. A method for determining a value stored in a selected memory cell, comprising: precharging a first terminal of the first capacitor and a first terminal of the second capacitor using a precharge circuit; and reading the block at the sensing node using the memory data Sensing the selected memory unit; sensing the reference memory unit at the reference node using the memory reference read block; comparing the sense node to the reference node using a differential amplifier block during the ramp period, the differential The amplifier block includes the first capacitor, the second capacitor, and an output, wherein a second terminal of the first capacitor is coupled to the memory data read block and the first terminal of the first capacitor is coupled to the difference An amplifier block, and a second terminal of the second capacitor is coupled to the memory reference read block and the first terminal of the second capacitor is coupled to the differential amplifier block; The value stored in the selected memory cell is indicated at the output of the differential amplifier block. The method of claim 21, wherein the selected memory unit is a split gate flash memory unit. The method of claim 21, wherein the sensing node ramps down during the sensing period. The method of claim 21, wherein the reference node ramps up during the sensing period. The method of claim 21, wherein the differential amplifier block comprises a comparator. The method of claim 25, wherein the comparator is a single comparator.