1234645 Description of invention: [Technical field to which the invention belongs] 1. The present invention relates to a temperature sensing device in a specific circuit and a method for temperature sensing. 5 [Prior art] As for high-power circuits such as power amplifiers for audio speakers and linear power supply regulators, it has the possibility of a fault condition of high on-chip current caused by an external short circuit. The dissipation of power on the chip due to these currents can result in excess temperature, which can degrade the 10 circuit characteristics on the silicon chip, and in extreme cases can even pose a fire hazard. For this reason, such power circuits are often provided with a thermal shutdown function, and the power output is disabled when the chip temperature exceeds a preset limit of, for example, 150 ° C. To implement this function, an on-chip circuit is required to detect and flag when the preset temperature threshold is exceeded. It also requires a 15 temperature detector in some microprocessor systems, such as when the microprocessor is clocked at high speed. In such a system, the limit is reached, the clock can be slowed down to reduce the supply current drawn by the microprocessor, and / or an output signal can be provided to turn on the fan. In the early days, the Zener diode voltage was resistively divided and applied 20 to the base of a common emitter bipolar transistor. To turn on the base of the bipolar transistor, the emitter voltage (Vbe) must be set to a minimum. C is reduced by about 2mV, so that the temperature is increased as it is applied at a fixed voltage (or even the rising voltage when Zener has a positive temperature coefficient tempco), a temperature will be reached, where the bipolar transistor is turned on and its set The pole current can then be used as an output. 1234645 As the supply voltage has been reduced, this method has become impractical because the typical Zener voltage, which is difficult to reliably achieve below 5 to 7V, is too large. Instead, as shown in the examples of US Pat. Nos. 3,959,713, 4,691,688, 4,574,205, and 5,099,381, the use of a band gap voltage instead of the Zener 5 voltage has become a common practice. For example, US Patent 381 describes a circuit where the band gap voltage from the Brokaw grid is compared to the Vbe multiplier voltage. To avoid instability caused by electrical and / or heat near the door monitor temperature, some local positive feedback can also be applied to provide a switching point with some hysteresis. A temperature detection circuit using a bandgap voltage source and feedback to provide hysteresis is described in US 10 5,149,199. General background knowledge in the field of temperature detection can be found in US 6,181,121, US 2002/0093325, US 6,188,270, US 6,366,071, US 5,327,028, US 4,789,819 and US 5,095,227 patents.
IEEE Journal of Solid-State Circuits, July 1996, Vol. 31, No. 7, Nos. 933--15, 937, "Micropower CMOS by A. Bakker and J.H. Huijsing"
"Temperature Sensor with Digital Output" describes a type of CMOS temperature sensing in which the current proportional to the Vbe voltage is compared to a reference current, which is compared to the base-emitter voltage reference by adding a PTAT (proportional to absolute temperature) current The temperature formed by the current source is independent and independent. The 20 sum of these two currents is approximately independent of temperature, because they have opposite temperature coefficients: positive for PTAT current; negative for vbe current. However, the circuit of Bakker and Huijsing Quite complicated (see, for example, Figure 4), and its sensitivity can be improved. Another temperature detection circuit is described in the US 5,980,106 patent, which also uses a bandgap reference. The 1A and Figure 1B shows the principle of this 1234645 circuit. Broadly speaking, the current sources 1G, 2G, which have positive and negative temperature coefficient characteristics of 12, 22, respectively, are applied to the inverting phase in the u figure, and are coupled to the-output circuit- Detect node A. As can be seen from the inspection in Figure _1B, the inverter output will be switched by the switching gate of the inverter when the voltage at point A passes the threshold _threshold 5 temperature TD. Υ§ , Ι 6 also teaches the application of feedback to the rigid node A as shown in Figure 3A of 106. A detailed degree detection circuit (Figure 4) is also described, which is based on the thermal voltage (ντ) The current Ith is combined with the current derived at the node A from the bandgap reference Ibg (eliminated at the t channel by the negative temperature coefficient k). However, the circuit of US'106 is also quite complicated and includes Changmai Ding's floating bipolar transistor. It is intended to be simpler, cheaper and easier to assemble temperature sensors. Band gap voltage often appears in circuits such as voltage regulators, but in applications such as amplifiers It is not necessary, so it does not depend on the apparent band gap of 15 volts to generate a better configuration. In addition, it has been known that it is different only by using certain reference temperatures that can be referred to by the temperature coefficient. It is basically possible to compare the temperature coefficient with a predictable absolute value, or at least a predictable relative value, to build μ degrees, and at the same time, more and more circuits are being implemented using CMOS rather than bipolar technology. Manufacturing, even if it ’s like speaker power Large traditional state of the bipolar field 20 and vice versa (see e.g. Fairchild FAN 7021). CMOS exclude many applications to use the conventional art technologies.
[Mingchi J. According to a first aspect of the present invention, it is therefore provided with a temperature sensor including: a current mirror with an input and at least two outputs; a 1234645 reference current generator with a first current The input and a first current output are configured to generate a first reference current with a positive temperature coefficient at the first current output in response to the first current input; a second reference current generator having a second The current input and a second current output are configured to generate a second reference current with a negative temperature coefficient at the second current output in response to the second current input; and wherein the first and second reference One of the generators has a respective current output coupled to the input of the current mirror; the first current input of the first reference generator and the second current input of the second reference generator are commonly coupled to the current mirrors One of the first 10 input nodes; and the other of the first and second reference generators has respective current outputs that are coupled to the second of the current mirrors to provide an electrical Sensing node; and wherein the first reference current generator comprises a bipolar transistor thermal voltage reference current source, the second reference current generator comprises a temperature-dependent reference current source semiconductor feature. 15 In this specification, the term current source includes a negative current source, which is the source in which current flows into the source (sometimes referred to as a "transducer"), and the current can therefore flow into a current source output . Broadly speaking, two reference current sources are provided, both of which intersect with the same current mirror. One of the current sources is called the bipolar transistor base emitter voltage or is substantially proportional to it (negative current is 20) The other of these current sources is called the bipolar transistor thermal voltage or is substantially proportional to it (the mathematical term is kT / q, where k is the Boltzman constant, T is the absolute temperature of Kelvin, and q is the electron Charge). This thermal voltage reference current source is sometimes referred to as a PTAT (proportional to absolute temperature) source, although in practice if the output is extrapolated back to an absolute value of 0, there may be deviations. 1234645 This configuration provides a particularly simple and elegant temperature sensing circuit, whose performance parameters can be determined fairly directly, and it is done fairly consistently in practice. In a preferred embodiment, the thermal voltage reference current source includes a pair of bipolar transistors. 'One of these transistors also provides a base emitter voltage.' The fifth second current source can be used to provide further information. Simplify and more closely lock the daddy of the two current sources together. The temperature sensing circuit is suitable for assembly with MOS, especially CMOS technology, and in this case the circuit is such that the bipolar transistor used in such current sources can be included in CMOS technology which is inherently parasitic ( Vertical or 10 lateral) devices, typically vertical PNP transistors in P-based CMOS and vertical NPN transistors in N-based CMOS. This circuit can also be configured in BiCMOS. In other embodiments, the first (positive temperature coefficient) source may use MOS instead of a bipolar transistor, such as an AVgs instead of AVbe configuration, and the 15th (negative temperature coefficient) source may include MOS. VT is the reference or low current Vgs is the source of the reference. In a preferred embodiment, the temperature sensor includes a positive feedback and this can be advantageously applied by injecting current to a common input node. This positive feedback is easy to have the result of switching behavior at the output of the current sensing node, so that it is encouraged when the output starts to change the positive feedback. This positive feedback also provides hysteresis near a threshold switching temperature. In an embodiment, the feedback may be provided in the form of a differential amplifier, or where one of a pair of transistors has an input from the current sensing node and the other has a pair of long-tailed inputs whose inputs are connected to a suitable bias. Crystal. Preferably, the inductor also includes an input 1234645 output circuit to provide substantially the same as the temperature of the circuit (more specifically, the bipolar transistors) is higher or lower than the threshold considering the hysteresis. A binary output. In a related aspect of the present invention, a method 5 for temperature-dependent signals is provided. The method uses: a current mirror with an input and at least two outputs; a first reference current generator with a first current input and a first A current output; a second reference current generator having a second current input and a second current output; and wherein one of the first and second reference generators has a respective current output coupled to the current Mirror input; the first current input of the first base 10 quasi-generator and the second current input of the second reference generator are shared to one of the first input nodes of the current mirrors; and the first The other of the first and second reference generators has respective current outputs coupled to the second of the current mirrors to provide a current sensing node; the method includes responding to a positive temperature coefficient from the shared first node Use 15 to use the first current generator to generate a first transistor thermal voltage reference current at the first current output; in response to the negative temperature coefficient from the common input node, use The second current generator generates a second reference current transistor voltage in the second current output; and the sense node in combination with these first and second reference signal to provide a current dependent of the temperature dependent signal. 20 It will be understood that the combination of these signals involves comparing them to each other or subtracting each other from each other. The temperature-dependent output signal (at the sensing node) may include a current or a voltage signal. In another aspect of the present invention, a temperature detection circuit is provided, including: a current mirror having an input and a current output after the first and second mirrors, the 10 1234645 input and the output after the first mirror are passed through separate The first and second MOS transistor channels are coupled to the respective first and second transistors to set a ratio of the current density in the first and second transistors to provide a signal from the second mirror. A positive temperature coefficient of current output; a third MOS transistor has 5 a gate connection coupled to a gate connection of the first MOS transistor and a pair of channel connections, and one of the channel connections passes through A resistor is coupled to one of the first and second transistors in common to provide a negative temperature coefficient current output at the other channel connection, where the current output is used as the first of the first transistor Based on a voltage-dependent reference, the other channel connection is coupled to 10 current outputs behind the lens to provide a temperature-dependent output. In a related aspect of the present invention, a temperature detection circuit is provided, including: a current mirror having an input and a current output after the first and second mirrors, the second and the first mirror output passing through respective first and A second M0S transistor channel is coupled to each of the first and second transistors; a third M0S transistor 15 has a gate connection coupled to a gate connection of a first M0S transistor and a For a channel connection, one of the channel connections is coupled to one of the first and second transistors via a resistor to jointly connect to provide a negative temperature coefficient current output at the other channel connection, here The current output is used as a reference for the first dependent voltage of the first transistor, and the other channel connection is coupled to the current mirror input to provide a negative temperature coefficient current from the current output after the second mirror; And the ratio of the current densities in the first and second transistors determines a positive temperature coefficient current, which is combined with the current output from the second lens to provide a temperature-dependent output. In one embodiment, the positive temperature coefficient current is a current flowing in the first MOS signal 11 1234645 crystal signal. In the special embodiment described by the latter, the first and second transistors are bipolar transistors, the first MOS transistor has its drain and gate electrodes connected together, and the second MOS transistor has A resistor is connected between its source and the second bipolar transistor. Each bipolar transistor, which would be parasitic in CMOS technology, has its base and collector connected together. A feedback circuit is preferably used so that the temperature-dependent output roughly exhibits bi-stable behavior with some hysteresis on either side of a threshold temperature. Facilities may also be included, for example, by effectively adjusting the resistor (to convert the first 10 bipolar transistor base-emitter voltage to current) and / or by effectively injecting current to the temperature-dependent output Or draw the current to adjust the threshold temperature. In a further aspect of the present invention, a method for generating a temperature-dependent signal is also provided. The method includes: using one of 15 pairs of transistors operating at different current densities to generate a thermal voltage reference positive temperature coefficient; A transistor voltage negative temperature coefficient signal; and subtracting one of the positive and negative temperature coefficient signals from the other of the signals to generate the temperature dependent signal, where the temperature dependency of the temperature dependent signal is greater than the temperature dependent Any of the subtracted signals. 20 Preferably, the transistors are bipolar transistors and the transistor voltage is a base emitter voltage. The use of a thermal voltage reference and a base-emitter voltage reference signal is preferably a current signal rather than a bandgap reference. This promotes the use of the same transistor to generate both Vbe and PTAT currents. Both are used. Furthermore, by subtracting the positive and negative temperature coefficient signals from each other, the effective temperature coefficient is increased, and the temperature dependence of the temperature-dependent signal is enhanced. Preferably, the subtracting comprises applying the positive and negative temperature coefficient signals to a detection node. A positive feedback can also be applied to the common bipolar transistor, which is the transistor used to generate the positive and negative temperature coefficient signals. 5 In a related aspect of the present invention, a circuit for generating a temperature-dependent signal is also provided. The circuit includes: the facility uses one of the transistors operating at different current densities to generate a thermal voltage reference positive temperature coefficient of the transistor; the facility uses the pair The transistor voltage produces a transistor voltage negative temperature coefficient signal; and the facility subtracts 10 of the positive and negative temperature coefficient signals from the other of the signals to generate the temperature-dependent signal, where the temperature of the temperature-dependent signal is Dependency is greater than any of these subtracted signals. These and other aspects of the present invention will now be briefly described with reference to the drawings, which are further described by way of example only, where: Figures 1A and 1B respectively show a current detection circuit based on a current source, and Thermal characteristics of the current source in the circuit of Figure 1A; Figures 2A to 2C show a self-biased reference current source, a Vbe reference current source, and a thermal voltage reference current source; Figures 3A to 3D respectively show the basis The first and second embodiments of the hysteresis-free temperature 20 detection circuit of the present invention, and the first and second embodiments of the hysteresis-required temperature detection circuit according to the present invention; FIG. A third embodiment of the temperature detection circuit; and FIG. 5 shows a fourth embodiment of the temperature detection circuit according to the present invention. 13 1234645 L Embodiment] Referring to FIG. 2A, this shows a so-called self-biased reference current source 200, which includes a current mirror 202 and a current source 204. A current is set to an input 206 of the current mirror at an output 208 of the current mirror, and the reference current source 5 204 provides an output current at an output 210, which depends on the current and current to the input 212. The output 210 may be a source or a sinking current, and is a sinking current in the illustrated example. In general, the output of this current source is approximately fixed over a range of output current but will be reduced with a small input current. The reference current source 200 utilizes a so-called modified circuit feedback technique in which 10 the current source output is connected to the current mirror input and vice versa. This circuit has a stable operating point, where (for a 1: 1 current mirror) Iout = Iin, that is, the input current to the current source is equal to the output current of the current source. This reduces the supply voltage dependency of the output current. Figures 2B and 2C show the application of the basic techniques of Figure 2A. Figure 2B shows the use of, for example, "Analysis and Design of Analogue Integrated Circuits" by PR Gray, PJ Hurst, SH Lewis, and RG Meyer, published in John Wylie 4 / E 2001. Chapter 4, Section 4 · 4 · 2 A bipolar transistor-based base-emitter voltage reference current source as described in CMOS technology. The base-emitter reference current source 220 in FIG. 2B is supplied with a positive power source Vdd and ground wires 222, 224. Transistors 226 and 228 include a current mirror equivalent to current mirror 202 in Figure 2A, transistor 228 provides the input, and transistor 226 provides the outputs. Transistors 232, 234, and 236 and resistor 238 include a current source equivalent to current source 204, and transistors 232 and 234 are configured to apply the base emitter voltage of transistor 236 (actually a diode junction voltage ) 14 1234645 Through the resistor 238, make lQut = Vbe / R238 (because the transistors 232 and 234 carry the same temperature detection circuit, and if they match, they have the same gate-source voltage). Transistor 230 provides only an additional output from the current mirror to provide a current output on line 231 equal to Iout. 5 Figure 2C shows a thermal voltage (VT) reference current source 240. The circuit in Figure 2C is similar to that in Figure 2B, and the same components are indicated by the same component numbers. Specifically, a current mirror containing transistors 226, 228, and 230 is again provided, but a different thermal voltage reference current source is used. The bipolar transistors 246, 248 operate at different current densities, for example, by providing different emitter areas to them, but they carry the same current, so that (using the Ebers-Moll formula) the difference between them is equal to ( ] <: Ding / 9) 111 (11 /】 2) = \ ^ 111 (] 1 /] 2), where VT = kT / q is the so-called thermal voltage (k, T and q are as defined above), in Representing the log with e as the base, J1 and J2 are the (base) current densities of the transistors Qpi and Qp2, respectively. At room temperature (27 ° C), VT is 25.9mV, and at 150 ° C, VT = 36.5mV. 15 Therefore, in the source 240, the output current 1 _ = ^ 丁 / 11250 111 (: [1/12), which is approximately proportional to the absolute temperature. (For the sake of simplicity, I assume that the resistor has a temperature coefficient of 0. In practice, the integrated circuit may have a temperature coefficient of about 2000 ppm / ° C, but if all resistors are made of the same material , Its temperature coefficient will be on orbit, and subsequent effects will be eliminated to at least 20 as the first order). Referring now to FIG. 3A, this shows a first embodiment of a temperature detection circuit 300 according to the present invention. This circuit is built on the basic principles described above. Referring to Fig. 3A, in broad terms, MP1, 2, 3, MN1, 2, QP1, 2 and R1 include a thermal voltage voltage reference current source similar to that shown in Fig. 2C. More 15 1234645 In detail, the MOS transistors MP1 and MP2 form a current mirror 202 having an input 302 and a wheel-out 304, which broadly corresponds to the current mirror 202 of FIG. 2A. MOS-electric body MN1 and MN2, bipolar transistors Qp 1 and Qp], and resistance cry r 1 contains a vT reference current source, which actually has one output on line 302 and 5 on line 304 One input, and therefore corresponds broadly to the current source in Fig. 2A ... 204. The MOS transistor MP3 provides an additional output from the current mirror on line 306. … MOS transistors MN2 and MN3, bipolar transistor QP2, and resistor R3 together contain a Vbe reference current source based on the base of the PNP bipolar transistor Qp2 and the 10-emitter voltage reference. Line 306 also effectively carries one output from this current source. It will be understood that this base-emitter reference current source has a different combination from the one shown in Figure 2B, because the output of the current mirror driven by the thermal voltage reference current source is not its own. Servo. It will also be understood that in the assembly of FIG. 3A, the MOS transistor MN2 and the bipolar transistor 15 Qp2 are common to both the thermal voltage reference and the Vbe reference current source. In Figure 3A, the relative size of these MOS transistors is expressed by the variable μ. It can be seen that the size ratio of the current mirror transistor MP1, MP2 and MP3 is LuMP1: MP2: MP3 = 1: 4: 4. A 4: 1 current mirror is formed so that the current passing through MP1 is 1/4 of the current passing through MP2 (and 20 1/4 of the current passing through MP23). The MOS transistors MN1, MN2 and MN3 have the same ratio, that is, MN1: MN2: MN3 = 1: 4: 4. The bipolar transistors QP1 and (^ > 2 both have their base and collector joints connected, and the size ratio QPl: QP2 = 4: 1, that is, the emitter area of the transistor QP2 is designed as the transistor qP1 1/4. Next, the operation of Figure 3A will be described. 16 1234645 Assume that the line 306 (this is the connector "0UT1") is initially connected to the voltage source from the outside, which is high enough for the MOS transistor MN3 Maintained in its saturation (fixed current) region, and low enough to keep the MOS transistor MP3 in its saturated (fixed current) region. It is also assumed that all other MOS transistors are saturated and carry 5 currents. The transistors MP1 and MP2 As mentioned previously, a 4: 1 current mirror is included, so that the current through the MP2 is 4 times the current through MP1. These currents pass through the transistors MN1 and MN2, respectively, and thus through the bipolar transistors (^^ and Qp2, respectively). Since the current through QP2 is 4 times the current through QP1, and the emitter area is 10 times that of transistor QP1, the current density of transistor QP2 is 16 times that of transistor QP1. As previously, having A pair of bipolar transistors with a current density of J1 / J2 ratio will have a Vbe of (kT / q) ln (Jl / J2) The difference is 25.9mVxln (16) in some cases, ie at T = 27. (: about 72mV, or 35.6mVxln (16), about 101mV at 150 ° C. 15 Now consider the MOS transistors MN1 and MN2. Transistor MN2 carries 4 times the current of transistor MN1, and its size is 4 times, so that] the gate-source voltage Vgs of VJN1 will be substantially the same as the gate-source voltage of transistor MN2. The gate of transistor MN1 is connected to the gate of transistor MN2. The source of transistor MN1 and the source of transistor MN2 will have the same voltage. This is the base-emitter voltage of the bipolar transistor QP2. This voltage is applied to the upper end of resistor R1, and the lower end of resistor R1 is the base-emitter voltage at the bipolar transistor QP1. Therefore, the voltage across R1 is equal to the difference between Vbe, △ Vbe = lOlmV, and the current through R1 and thus the line 302 is 101mV / Rl. This current is then shaped by the transistor MP3 at a ratio of 4: 1 17 1234645 into a mirror effect 'paid into the line 306 (that is, into or The current through the node ουτί) is equal to 404mV / Rl at 150 C, and has a positive temperature coefficient. Because this current and thermoelectricity It is proportional, VT = kT / q, so it is actually a PTAT current. Now consider the Vbe reference current source. As mentioned earlier, the voltage at the source of transistor 5 MN2 is the base of bipolar transistor QP2 The emitter voltage, and again as previously mentioned, transistor MN3 is selected to be the same size as transistor MN2. Now assuming that MN2 and MN3 have the same gate-source voltage, the voltage at the source of transistor MN3 will also be approximately equal to the base-emitter voltage of bipolar transistor QP2. Therefore, the current through r3 and thus through MN3 to spring 10 node OUT1 will be approximately (QP2Vbe) / R3. Furthermore, since Vbe has a negative temperature coefficient, typically _2mV / ° C or equivalent -3000ppm / ° C, the current will pass through MN3 to node OUT1. In the circuit shown, R1 is selected as 44kQ to set the current through MP3 to I (MP3) = 404mV / 44k0hm = 9.20uA and the current through QP2 15 to I (QP2) = 9.20uA / 4 = 2.30 uA. In a manufacturing process, this results in Vbe (QP2) = 462mV, and R3 is therefore set to 462mV / 9.20uA = 501Ω, so at 150 ° C, I (MN3) = I (MP3). _ Then if the temperature rises above 150 ° C, the current through transistor MP1 and thus through MP3 rises, and the current through transistor MN3 decreases, resulting in a current of 20 to the external voltage source from node OUT 1 result. If the temperature drops below 150 ° c, the current through transistor MP1 and thus MP3 decreases' and the current through transistor MN3 rises, resulting in the current flowing from the voltage source into the OUT1 node. If the connection between the voltage source and the node ουτί is loosened, the voltage level of this node will rise or fall respectively, and eventually make [^^ 18 1234645 and MN3 no longer saturated to balance the current. It can be seen that the υυτι node is larger than the node a in the basic configuration of Fig. 1A. '' The choice of transistor size can depend on the needs of any particular application. In terms of integrated circuit implementation, the main considerations include the chip occupied by the components: 5 products and minimize the impact of nominal mismatch between the same devices. Typically, the random deviation voltage between the 雔 polar transistors and the resistors will be less than the deviation voltage between the transistors in the circuit, and the expansion of manufacturing will be governed by the mismatch between] ^ 1 ^ 2 and] ^] ^ 1 Because this error is basically superimposed on the small quiescent voltage through R1. 10 First consider the choice of the ratio of MN2 to MN1. As the above circuit, but the ratio between MN2 and MN1 and between MP2 and MP1 is 1, it will still be available with proper adjustment of Ri. However, the current density between QP1 and QP2 is only 4 instead of 16, so you will only get half the voltage T (q) ln4 instead of (kT / q) lnl6) through R1, making the circuit for MN2 and MN1 Mismatches between 15 are more sensitive. In order to restore the current density ratio, QP1 can be 16 times of QP2, but this will take up a lot of silicon area. On the other hand, if the ratio between MN2 and MN1 and between MP2 and MP1 is 8: 1 instead of 4 :: 1, this will only increase the voltage through R1 by a factor of 1η32 / 1η16 = 1.25, but the MOS transistor is already large To reduce manufacturing tolerances and double the area. As far as the technology under consideration, 4: 1 20 is selected, but its optimum value will depend on the limitations of the particular manufacturing technology. Now consider the ratio of MN3 to MN1. As noted above, the voltage through R1 is approximately 100mV at 150 ° C and approximately 450mV through R3, and these resistors are required to pass the same current. If MP3 and MN3 are the same size as MP1 and MN1, respectively, the resistance of R3 will be approximately 4.5 19 1234645 times the resistance of R1. For best performance when using parasitic vertical transistors in CMOS, Qp] and QP2 operate optimally with a small amount of micro-current. At the same time, multiple applications have strict power pre-differentiation, and in such applications, these resistors tend to have tens of thousands of ohms and occupy a large area. Introducing 1 ^^^ 3 pairs] ^] ^ 1 of 4: 5 The ratio of 1 to R3 and R1 is similar, so the total resistor area tends to be the best. The transistors MP2 and MP3 are preferably formed of multiple units, each having a configuration similar to that of MP1. It preferably has a large channel length for medium matching and high output impedance 'but has a small channel width to length ratio W / L at 10 to maintain Vgs-Vt as large and good current medium matching. Transistor MN2 is similar to MN3 in that multiples configured for MN1 are better 'and preferably Vgs-Vt is large and has a good current medium. However, if Vgs-Vt is large, this will cause subsequent variation of Vgs (MN3) and attenuate the temperature coefficient of I (MN3) (basically put the resistance of 1 / gm (MN3) in series with 15 connected to R3) 'Therefore, these transistors should generally be designed with a sufficiently large W / L to obtain vgs-vt < 100mV (for example, at a critical temperature). Then 丨 / gm (MN3) is about 10% of R3 ′ and does not degrade the temperature sensitivity of the circuit or introduce manufacturing sensitivity due to the non-relevant and electrical characteristics of the resistor. Reviewing the above description of the operation of the circuit, it can be seen that the thermal voltage base 20 is "servo" the current mirror, and the current mirror also drives the detection node. The reference based on the base emitter uses the same transistor as the thermal voltage reference to provide a second, negative temperature coefficient output, which is subtracted from the positive temperature coefficient based voltage reference at the detection node. It will be understood that this configuration may be exchanged such that the Vbe-based benchmark is used in conjunction with the 20 1234645
The Vbe-based reference is the same as the thermal voltage reference of the transistor, which is servoed to the current mirror (which again drives the detection node) and also drives the detection node. This alternative configuration is shown in Figure 3B, where the gate-to-row connection on transistor MP1 has been moved to transistor MP3, and its output is taken from QUT2, line 3025, which is a transistor The junction of MP1 and MN1. The analysis and component values remain the same, at least for the first order. The main difference is that the current consumed by the circuit now has a negative rather than a positive temperature coefficient. The circuits described so far that do not require feedback tend to oscillate near the metastable state, and positive feedback is thus provided to provide the magnetic tape as desired. Figure 3C shows the expansion of the circuit of Figure 3A to implement this. The MOS transistors MP4 and MP9 provide further output from the current mirror, which is used as a fixed current source. The line 306 is connected to an output transistor MP5, which is a different combination from the transistor MP6, and is connected to a common current source provided by the transistor MP4 'transistor MP6 which provides positive feedback, as described in more detail below. The gate of transistor MP6 15 is connected to a bias line with a voltage similar to the voltage source previously discussed for node 306, so that when the gates of MP5 and MP6 are at the same voltage, the MN3 and MP3 are saturated. To avoid the deterioration of these temperature-dependent currents at or near the threshold temperature. Transistors MN10 and MN11 include a further current mirror, and in conjunction with the transistor "㈧ includes an output circuit 20 to substantially drive one of the output lines 310 between the power supply rail VDD and Vss (or ground) in order to drive the logic circuit. In the circuit of Figure 3C, the positive feedback system is provided by the transistors mp4, 5 and 6. At cold temperature, the node 01711 will be low, so the transistor 5 will be turned on, and it is noted that the transistor 5 When the electricity of the channel of MP5 and MP6 is 21 1234645 (determined by MP4), the transistor MP6 is turned off. As the temperature rises, the transistor MP5 starts to turn off and the transistor MP6 starts to turn on, thus driving some Current (from MP4) into transistors MN2 and QP2. This increases the voltage at the gate junctions of transistors MN2, MN1 and MN3 by Δν. At present, any changes in Vgs of MN1 and MN3 and Vbe (QPl) are omitted. Any variation of this will increase the current through the MOS transistor by the ratio △ V / (I (Rl) .Rl) = AV / (AVbe) = Z \ V / 101nA ^ 〇, and increase the current through the transistor MP1 and therefore through The current of MP3 will further encourage the rise in node OUT1. It will also increase the pass The current of R3, but with a small proportion of △ V / (I (R3) · R3) = △ V / Δ Vbe 10 = △ V / 462mV. The rise in generating I (R1) is not exactly △ V / R1 'The reason is that the extra feedback current provides a 4: 1 ratio of the current in MN2 and MN1, so that these transistors now have slightly different gate-source voltages, and the Vbe of QP1 and QP2 will also be different, but the overall The effect is still that I (MP3) is much improved compared to I (MN3). 5 This process continues until transistor MP 5 is completely turned off and transistor MP6 essentially carries the entire current through transistor MP4. At this point MP4 effectively appears in parallel with transistor MP2 and changes the ratio of the current mirror. Therefore, when the temperature finally decreases, the thermal trip point is lower than when the temperature was increasing previously Temperature, and k provides the desired hysteresis effect. It will be seen that the positive feedback does not directly set the positive or negative temperature reference current but instead changes the current by adding to the output current from transistor MP2 Current ratio in the mirror. This changes the reference and thermovoltage reference currents, but changes the thermoelectric The reference current is more, because in fact changed through transistor MN1 and MN3 and thus balance through transistor 22 1234645 MP3 and MN3 of the current. Therefore, the feedback is not directly to the order
The Vbe-based reference source may return directly to the output node ουτι, but instead of ~ return to a common node (line 304) and transistor (bipolar transistor QP2). Coming · The pole current from QP5 is compared with the fixed current through MP9 by mirrors MN10, MN11 5 to give the logic signal for swinging on the line HOT. … Figure 3D shows a feedback approach similar to the circuit applied to Figure 3B. Note that ‘the signal at the comparison node OUT2 goes lower than the temperature threshold’, so the MP5 current is now fed into node 304 to provide positive feedback. · 10 Referring now to Figure 4, this shows the same basic type of temperature detector 400 as shown in Figure 3C, and the same component numbers represent the same components. In the circuit of Fig. 4, a first 402 and a second 404 temperature adjustment circuit are provided to allow external adjustment of the threshold temperature of the circuit. The temperature adjustment circuit 402 controls the transistor MNX to inject a part of the positive temperature coefficient current into the resistor chains R3A, B, and C with the extra output of the current mirror provided by the transistor MP10. This additional pull-up current reduces the threshold temperature. The temperature adjustment circuit 404 controls the transistor MN9 to reduce the resistance of the resistor chain ruler 3 Γ or short the lower portion R3 A, which increases the Vbe / R3 current and thus raises the threshold temperature by 20 °. The temperature adjustment function provided by lines 402 and 404 can be used to change or adjust the temperature threshold, for example to provide an "early warning" function or to allow the thermal trip circuit to be implemented at room temperature when functionally testing manufactured parts . In detail, in Figure 4, the gate of the transistor Mp6 is tied to the gate of the transistor in the current mirror. As mentioned above, the gate of MP6 should be biased to an appropriate voltage so that both MP3 and MN3 are saturated when MP5 and MP6 are balanced. Here, as in the illustrated embodiment, this processing technique produces an alternative "low Vt" or PMMOS transistor with a reduced threshold voltage, and the voltage on line 5 406 is used to supply this bias Without having to force Mp4 out of this saturation region. In a process that does not require this option, the gate of Mp6 can be connected to some other suitable point. It will be understood that the circuit shown in Figures 2B and 2C has a second stable state in which all transistors are turned off. As long as a small starting current (for example, through 10 transistors 236) is sufficient to take the circuit out of this state. This can often be coupled with leakage current or capacitive current supply when powered, but a "start-up" circuit can be used to ensure that the circuit reliably leaves its zero current state. Fig. 5 shows an example 500 of a temperature detection circuit configured in accordance with Fig. 3D and containing the start-up circuit. In Fig. 5, the same 15 elements as those in Fig. 30 are indicated by the same element numbers. In the circuit in Figure 5, MN5 provides a small current to the gate of the PMOS mirror, and its gate voltage is initially pulled up to Vdd by Mp7. MN5 is turned off only every time MN4 is turned on, and it only occurs when MN3 and thus MP3 have begun to pass current. Similar techniques are applied using the circuits of Figures 3C and 4. Other solutions are familiar to circuit designers. 2 Engineers are easy to understand. No doubt how effective the change will be for those who are familiar with it. For example, although specific embodiments have been described with reference to PNP bipolar transistors, those skilled in the art will readily understand that the circuit can be reversed and NPN bipolar transistors can be used. CMOS-treated vertical parasitic transistors will typically be used, but parasitic side 24 1234645 type transistors (such as MOS transistors with collector, base, and emitter acting as the collector, base, and emitter respectively) or parasitic Diodes (because bipolar transistors are basically used to provide diode junctions) can be used in principle because the circuit is insensitive to low-beta types of such transistors. 5 In other embodiments, the bipolar transistors QP1 and QP2 can be replaced by MOS transistors with a size ratio. Preferably, these MOS transistors are operated in the subthreshold region, where they exhibit bipolar exponential IV characteristics, but even outside the subthreshold region, they will provide smaller but still positive temperature coefficient currents. . It will be understood that the invention is not limited to the described embodiments and encompasses modifications apparent to those skilled in the art that are within the spirit and scope of the patentable scope attached hereto. Simple illustration of L diagram] Figures 1A and 1B respectively show the current source-based temperature detection circuit and the thermal characteristics of the current source in the circuit of Figure 1A; 15 Figures 2A to 2C each show a self-bias voltage Reference current source, a Vbe reference current source, and a thermal voltage reference current source; Figures 3A to 3D show the first and second embodiments of the temperature detection circuit without hysteresis according to the present invention, and according to this The first and second embodiments of the temperature detection circuit requiring hysteresis of the invention; 20 FIG. 4 shows a third embodiment of the temperature detection circuit according to the present invention; and FIG. 5 shows the first and second embodiments of the temperature detection circuit according to the present invention. Four embodiments. 25 1234645 [Representative symbols for main components of the drawing] 10 ... current source 234 ... transistor 12 ... positive temperature coefficient characteristic 236 ... transistor 20 ... current source 238 ... resistor 22. .. Negative temperature coefficient characteristic 240 ... Thermal voltage reference current source 30 ... Inverter 242 ... Transistor 200 ... Self-biased reference current source 244 ... Transistor 202 ... Current mirror 246. ..Transistor 204 ... current source 248 ... transistor 206 ... input 250 ... resistor 208 ... output 300 ... temperature detection circuit 210 ... output 302 ... input 212 ··· Input 304 ... Output 220 ... Base emitter reference current source 306 ... Line 222 ... Positive power supply 308 ... Bias line 224 ... Ground line 310 ... Output line 226 ... Transistor 400 ... temperature detector 228 ... transistor 402 ... first temperature adjustment circuit 230 ... transistor 404 ... second temperature adjustment circuit 232 ... transistor 500 ... temperature detection circuit 26