TW201138320A - Analog-to-digital converter with sub-range and method thereof - Google Patents

Analog-to-digital converter with sub-range and method thereof Download PDF

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TW201138320A
TW201138320A TW99113279A TW99113279A TW201138320A TW 201138320 A TW201138320 A TW 201138320A TW 99113279 A TW99113279 A TW 99113279A TW 99113279 A TW99113279 A TW 99113279A TW 201138320 A TW201138320 A TW 201138320A
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array
capacitor
sub
overlapping
output
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TW99113279A
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Chinese (zh)
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TWI407702B (en
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Hung-Wei Chen
Hsin-Shu Chen
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Univ Nat Taiwan
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Abstract

An analog-to-digital converter with sub-range and method thereof is disclosed. By increasing overlapping capacitors within a MSB array of capacitor array, so as to reduce the accuracy requirement for comparators during the coarse conversion, as well as reduce a comparison time of the comparators heavily. The mechanism is help to improve the efficiency of conversion for analog-to-digital conversion.

Description

201138320 六、發明說明: 【發明所屬之技術領域】 其方法 本發明為有·-麵比數⑽換似及其方法,特別是指 種在電容陣财增加重疊電容之次關_比數位轉換裝置^ 【先前技術】201138320 VI. Description of the invention: [Technical field to which the invention pertains] The method of the present invention is a method for changing the ratio of the surface ratio (10), and particularly to the method of increasing the overlapping capacitance in the capacitor array. ^ [Prior Art]

近年來’隨著數位化的蓮勃發展,類比數位轉換裝置(亦稱之 為類比數位轉換器)扮演著關鍵性的重要角色。而如何使類比數位 轉換裝置具有更好_換效率則是各家廠絲欲解決的問題之 -般而言’類比數_換裝置的_是職比的輸入轉換為 相對應的數位輸出,在傳統的連續近似s(SuccessiveIn recent years, with the development of digital Lotus, analog digital conversion devices (also known as analog digital converters) have played a key role. And how to make the analog digital conversion device have better _ change efficiency is the problem that each factory wants to solve - in general, the analogy number _ change device _ is the ratio of the input to the corresponding digital output, in the traditional Continuous approximation s (Successive

Approximation,SAR)類比數位轉換器中,為了使每一位元作出正 確的判斷,故對每-位元均制姻的鮮度,相對地亦造成耗 費較多的比較時間。而為了提高分辨率,當電容陣列越大時電容 陣列穩定的制需要越長,制耗神及周邊電路複雜度則同樣 越大。 因此,便有人提出一種非二元搜尋演算法的連續近似式類比 數位轉換器(Non-Bmary Successive Approximation ADC),如:「F.In the Approximation, SAR) analog-to-digital converter, in order to make a correct judgment for each bit, the freshness of the marriage for each bit also relatively consumes a relatively long comparison time. In order to improve the resolution, the longer the capacitor array is, the longer the system needs to be stable, and the complexity of the system and the peripheral circuits are also larger. Therefore, a non-Bary Successive Approximation ADC (Non-Bmary Successive Approximation ADC) has been proposed, such as: "F.

Kuttner, A 1.2V l〇b 20MSamples/s Non-Binary Successive Approximation ADC in G‘13um CM〇s,” lsscc 吨化也上卿 PP. 176-177, 2002·」,用以解決上述問題並達到高速效果。但是, 以此方式需要使用複雜的數位控制器及溫度碼(Thenn〇meter coded)電谷陣列來貫現,以及進行錯誤校正。 m 3 201138320 有鑑於此,若能使用非二元搜尋演算法並避免使用複 位控制器及溫度碼電容陣列來實現,但同樣允許在最高位元轉 出現錯誤時進行校正,將有助於提升轉換效率。 換 - 綜上所述,可知先前技術中長期以來一直存在轉換效率不佳 - 之問題’因此實有必要提出改進的技術手段,來解決此—問題。 【發明内容】 ° 有馨於先前技術存在的問題,本發明遂揭露—種次區 比數位轉換裝置及其方法。 ' ' • 本發明所揭露之次區間的類比數位轉換裝置,包含:電容陣 列模組、轉換模組、控制模組及輸出模組。其中,電容陣列模組 接收類比訊號,並將類比訊號儲存於透過LSB陣列及陣列 後,產生電容陣列輸出’其中LSB陣列包含第一輕合電容,乂犯 陣列包含第二輕合電容及重疊電容,且第二麵合電容與重疊電容 相互並聯;轉換模組於預設的粗分(c〇arse)時脈週期中,透過第一 刖置放大n放大電容陣顺出,並將放讀的電料列輸出傳送 修至問鎖比較器’且根據問鎖比較器的比較結果產生粗分位元,並 將此粗分位元设定為決定訊號,以及於預設的細分㈣e)時脈週期 中’透過第-前置放大ϋ及第二前置放大器放大電容陣列輸出, 並毅域的f料列輸出傳送朗鎖比,且根據此閃鎖比 較器的比較結果產生細分位元,並將此細分位元設定為決定訊 號;控制模_以触決定峨喊棚設的控綱輯,並透過 此控制邏輯控制LSB陣列及MSB陣列;輸雌組⑽透過數位 錯誤技正電路將粗分位元、細分位元及重叠區域的比較結果進行 數位校正以產生數位輸出訊號。 201138320 2要訓的是’所述電容陣龍組為二進位式電容陣列。 4及的第—鮮電容的大小為“2(7·χ)· 小則為Ί”單位。另外卞㈣絲w h电合的大 控制邏輯包含重疊邏輯及標準邏輯,此 =輪=控制重4電容,而標準邏輯顧以控㈣一搞合電 容。接著,所述控制歡包含與重疊電容相同數 L S正反Θ ’用以作為暫存器並分別控制相應的重疊電容, 其中各D型正反器的”與相應的多工器電性連接,此多工巧至 少接收決定喊。在實際實施上,轉換模組在細分日娜週期中的 =了個日守脈週㈣’透過重疊邏輯計算出重疊區域,並根據此重 f區域控制重疊電容’而第—前置放大器及第二前置放大器將電 奋陣列輸纽大後,可透過增益控制多送至卩捕比較器。 至於本發明之次區間喃比數位轉齡法其步驟包括:接 收類比訊號,並賴比瓣u齡於㈣陣列及msb陣列後,產 生電料顺丨;糊設軸分(C_e)雜週射,透過第—前 置放大器放大此電容_輸出,並將放大後的電容陣列輸出傳送 至閃鎖比較器’且根據⑽味⑽味結果產生粗分位元,以 及將此粗分位元设定為決定訊號;於預設的細分(朽收)時脈週期 中,透過第一前置放大器及第二前置放大器放大電容陣列輸出, 並將放大後的電容陣列輸出傳送至閂鎖比較器,且根據閂鎖比較 态的比較結果產生細分位元,以及將此細分位元設定為決定訊 唬,接收決定訊號以執行預設的控制邏輯,並透過此控制邏輯產 生重疊區域及控制LSB陣列及MSB陣列;透過數位錯誤校正電 路將粗分位元、細分的位元及重疊區域的比較結果進行數位校正 以產生數位輸出訊號。 201138320 f述控制邏輯包含重疊邏輯及鮮賴,此重疊邏輯用以控 制重疊電容’而標準邏輯_以控制第—齡電容及第二麵合電 各。上述提及的第二叙合電容的大小為“ 2(7·χΜ,,單位,重疊電容 的大J則為1單位。另外,本發明次區間的類比數位轉換方法 .,包含,細分時脈週射的第二個時脈週期時,透過重疊邏輯計 ,出區域’並根據此重4區域控制重4電容的步驟,以及在 第則置放大減第二前置放大器將電容陣列輸出放大後透過 增益控制多送至_比較器。接著,所述重疊電容透過作 為暫存器的相同數量之D型正反器進行控制,其中各〇型正反器 的D端與相應的多工器電性連接,此多工器至少接收決定訊號。 而所述LSB陣列及所述MSB陣列則組成二進位式電容陣列。 本發明所揭露之裝置與方法如上,與先前技狀間的差異在 於本毛明是透過在電容陣列的MSB陣列中增加重疊電容,以便於 粗刀(Coarse)過程巾降低對比較器的正確性要求進*大幅減少比 較器的比較時間。 • 透過上賴技術手段’本發明可以制提高紐數位的轉換 效率之技術功效。 【實施方式】 以下將配合圖式及實施例來詳細說明本發明之實施方式,藉 此對本發明如何應用技術手段來解決技術問題並達成技術功效的 實現過程能充分理解並據以實施。 在《»兒明本發明所揭露之次區間的類比數位轉換裝置及其方法 之前,先對本發明所自行定義的名詞作說明,本發明所提及的重 疊電容(Overlapping-capacitor,0LC)是用於進行校正的電容並且 201138320 設置於MSB陣列中’這些重疊電_過相應的暫存器進行控制, 有關MSB陣列及重疊電容的詳細電路將在稍後配合圖式作詳細 說明。 … 以下配合圖式對本發明次區間的類比數位轉換裝置及其方法 作進一步說明,首先’先針對本發明次區間的類比數位轉換裝置 作說明’請參閱「第1圖」’「第1圖」為本發明次區間的類比數 位轉換裝置之方塊圖,包含:電容陣列模組1(n、轉換模組1〇2、 控制模組103及輸出模組1〇4。其中,電容陣列模組1〇1用以接收 鲁 類比祝號,並透過LSB陣列及MSB陣列產生電容陣列輸出,並 中LSB陣列包含第一耦合電容,而MSB陣列則包含第二耦合電 容及重疊電容,且第二耦合電容與重疊電容相互並聯。在實際實 施上,電容陣列模組101是由LSB陣列及MSB陣列組成的二進 位式(B inary-weighted)電容陣列。 轉換模組102用以於預設的粗分(Coarse)時脈週期中,透過第 一前置放大器(Pre-amplifier)放大電容陣列輸出,並將放大後的電 鲁 谷陣列輸出傳送至閂鎖比較器(Latch comparator),且根據此閂鎖比 較器的比較結果產生粗分位元,並將此粗分位元設定為決定訊 號;以及於預設的細分(Fine)時脈週期中,透過第一前置放大器及 第二前置放大器放大電容陣列輸出,並將放大後的電容陣列輸出 傳送至閂鎖比較器,且根據此閂鎖比較器的比較結果產生細分位 元’並將此細分位元設定為決定訊號,所述決定訊號為“〇,,或“1 ’’的數位訊號,用以決定二元搜尋演算法進行“加”或“減”的 計算。在實際實施上,電容陣列輸出可透過第一前置放大器及第 二前置放大器分成不同階段進行增益,用以降低閂鎖比較器的輸 201138320 入抵補(Input-referred offset)。舉例來說,在粗分的時脈週期時,電 容陣列輸出可透過一個第一前置放大器進行增益,而在細分的時 脈週期時,電容陣列輸出可透過一個第一前置放大器及二個第二 前置放大器進行放大(即增益),此部分的詳細電路圖將在稱後配合 圖式作說明。特別要說明的是,放大後的電容陣列輸出可先經由 增盈控制多工器(Gain control multiplexer)後,再傳送至閂鎖比較器 進4亍問鎖及比較’且本發明並未以此限定第一前置放大器及第二 前置放大器的數量。 鲁 控制模組丨〇3用以接收決定訊號以執行預設的控制邏輯控 制’並透過控制邏輯產生重疊區域及控制LSB陣列及MSB陣列。 所述控制邏輯包含重疊邏輯及標準邏輯,此重疊邏輯用以控制重 疊電容,如:「CL<0>==SC<N+1> | (SC<9>&〇LS<N>)、 CL< 1 >==SC<N>」,而標準邏輯則用以控制第一耦合電容及第二耗 & 電谷,如.「CL<〇>== SC<N+1>、CL<1>==SC<N>」,其中 n 為 位元、CL為控制邏輯、SC為狀態控制及〇LS為重疊狀態,在實 φ 際實施上’此重疊邏輯設計成可以對重疊的比較結果做第二次的 取樣’以及提供轉換模組1〇2在細分時脈週期中的第二個時脈週 - 期時,透過此重疊邏輯計算出二元搜尋演算法的重疊區域,以便 控制模組103根據此重疊區域控制重疊電容,而標準邏輯控制則 與習知的標準設計一樣。特別要說明的是,本發明並未以上述舉 例對重疊邏輯與標準邏輯作限定,在不脫離本實施例之精神和範 圍内’以等效電路所組成之相同功能,皆為本實施例之可實施手 段。另外,控制模組1〇3可包含與重疊電容相同數量的D型正反 器’用以作為暫存器並分別控制相應的重疊電容。其中,各D型 201138320 正反器的D端與相應的多工器電性連接,所述多工器至少接收轉 換模組102產生的決定訊號。 輸出模組104用以透過數位錯誤校正電路將粗分位元、細分 - 位元及重疊區域的比較結果進行數位校正以產生數位輸出訊號。 • 在實際實施上’使用二元搜尋演算法搭配重疊區域,用以檢查粗 刀位元疋否錯誤,並且於錯誤時進行數位校正。在此二元搜尋演 鼻法中女插有重璧區域的比較過程,所述重疊區域在實際實施上 是以電容陣列模組101輸入穩定的時間來計算,如果須要較長的 • 穩定時間則須要較大的重疊區域。 如「第2圖」所示,「第2圖」為本發明次區間的類比數位轉 換方法之流程圖,其步驟包括:接收類比訊號,並將類比訊號儲 存於LSB陣列及MSB陣列後,產生電容陣列輸出(步驟21〇);於 預設的粗分(Coarse)時脈週期中,透過第一前置放大器放大電容陣 列輸出,並將放大後的電容陣列輸出傳送至閂鎖比較器,且根據 閂鎖比較器的比較結果產生粗分位元,以及將此粗分位元設定為 φ 决疋訊號(步驟220);於預設的細分(Fine)時脈週期中,透過第一 前置放大ϋ及第二前置放大H放大電轉顺出,並將放大後的 •電料出傳送朗航較ϋ,且根期槪較詞比較結果 .產生細分位元,以及將此細分位元設定為決定訊號(步驟23〇);接 收決定訊號以執拥設的控觸輯,並透過此㈣賴產生重叠 區域及控制LSB陣列及MSB陣列(步驟24〇);透過數位錯誤校正 電路將粗分位元、細分位元及重疊區域的比較結果進行數位校正 以產生數位輸出訊號(步驟25〇)。在實際實施上,步驟2ι〇是透過 咖陣列及MSB陣顺成二進位式電容_來產生電容陣列輸 201138320 出而LSB P車列包含第-麵合電容,MSB陣列包含第二耗合電容 及重電谷,且第一耗合電容與重疊電容相互並聯。另外,可在 細/刀時脈週期中的第二個時脈週期時,透過重疊邏輯計算出重疊 區域’並根據此重疊區域控制重疊電容(步驟則,透過上述步驟, 即可在電谷陣列的MSB陣列中增加重疊電容,以便於粗分(c_e) 過程中降低職較ϋ紅雜要求,進而大幅減少味器的比較 時間。 以下配合「第3圖」至「第5圖」以實關的方式進行如下 說明,請先參閱「第3圖」’「第3圖」為應用本發明的電容陣列 模組之電路示意圖。電容陣列模組1〇1包含:類比訊號輸入端 301 LSB P車列310、第-搞合電容311、MSB陣列32〇、第二輕 〇電谷321、重疊電谷322及電容陣列輸出端33〇。特別要說明的 是’雖然上述以“12位元,,的電容陣列進行示意,然而,本發明 亚未以此限定電容陣列模組1〇1所包含的電子元件數量及類型。 另外,LSB陣列310及MSB陣列32〇之間更可連接整數電容 3_,而為了便於計算,故可如「第3圖」所示意額外增加凑數 電f 300a ’使整數電容獅的單位電容值能夠維持整數,以此例 而言,其凑數電容30〇a可由分別與、,,及“Vm,,連接的二個 輕合電容馳成,這兩她合電容共計為三十鮮位電容值如 此來,整數電容3〇〇b透過習知公式(例如·· “(32+3〇y31=2,,. 其中=為LSB陣列31〇的單位電容值、“3〇,,為凑數電容 ^00a的單位電谷值)進行計算後,即可得知整數電容鳥為兩個 單位電容值的輕合電容。 以“12”位元的電容陣列模組1〇1為例,此電容陣列模組收 201138320 會透過類比讯號輸入端301接收類比訊號,並且透過LSB陣列31〇 及MSB陣歹320產生電容陣列輸出,而所產生的電容陣列輸出會 透過電谷陣列輸出端330輸出至轉換模組1〇2,其中LSB陣列31〇 -包含多個第一耗合電容311 ’並且於「L1」至「L5」產生相應位 • 元。而MSB陣列320包含多個第二耦合電容321及重疊電容322, 且第二搞合電容321與重#電容322如「第3圖」所示意相互並 聯。 承上所述,在MSB陣列320中,前“6”位元是透過第二耦 • 合電容321「Mx」及重疊電容322「OLx」所形成,其中「x」代 表位元編號。其中’第二輕合電容321「Μχ」的尺寸(Sizes)為 「2(7x)-l」單位,舉例來說,第二耦合電容321「M1」的尺寸為 「2(^)-1=63」單位、第二耦合電容32丨「M2」的尺寸為「2(7·2)1=3 i」 單位、……並以此類推至第二耦合電容321「M6」的尺寸為 「2(7_6)-1=1」單位。因此,rMl+〇u」至「M6+〇L6」的第二耦合 電谷321及重疊電容322形成類似傳統連續近似式 • APPr〇ximati〇n, SAR)類比數位轉換器(Anal〇g_t〇_Digital c〇nverter, ADC)的二進位式電容陣列。 在粗分(Coarse)轉換過程中,重疊電容322「〇Lx」就像其他 - 第二耦合電容32i「Mx」執行二元搜尋演算法一樣,第二個位元 的細分轉換的結果可以改變其中一個重疊電容322的連接以執行 重唛比較。以一元搜尋演算法為例,假設類比輸入階層相當於數 值2020” ,在追蹤(Tracking)之後,相似於傳統的連續近似式類 比數位轉換益,其第一個決定點(Decisi〇np〇int) “2048”為全部輪 入範圍的一半,控制邏輯設定rM1」及r〇L1」連接至“Vrp” j 11 201138320 其導致參考來源受到大電容負載變更,因此,參考來源及電容陣 列輸出需要咖來穩定。假如_比較器在電容陣列輸出尚未完 王穩疋時進行解析位it ’將可能會產生—個錯誤的判斷。在此例 中,假設發生一個錯誤的判斷且比較器結果是以“丨,,代替“〇 ”。使「m」及「0L1」維持連接到“Vrp”,「M2」及「⑽」 連接至“Vrp”以形成第二個決定點“3〇72”。因為第二個決定點 的餘數電壓足夠大’即使電容陣列輸出尚未穩定,此閃鎖比較器 ,亦可進行板衫彳斷。「M2」及「OL2」在第二個決定點產生後從“ Vrp 切換至Vm” ’而剩下的粗分轉換則以相同的方式重複搜尋。 從第七個位元開始’電容陣列輸出要較長的時間穩定及更大 的增益放大;因此,類比電路具有高精確性以便精確轉換。不允 許在之後具有錯誤。在第七個決定點產生後,輸人(即:“2〇2〇”) 二於2143及2016之間。第七個決定點為“2〇8〇”且比較 ,器π果而為〇 。第七個結果表明輸入應介於“2〇8〇,,及“2〇16 ”之間,而在粗分轉換中,其錯誤決定只可能在“2048”。因此, 其需要在S 8位元飾檢查料點“纖”。第七她元比較器 、。果使Μ7」伙Vrp七刀換返回至“Vm”。第八個位元結果需 為 “〇”。 ,,-個重疊邏輯的函式用以在粗分轉換結果中確定第一個“ι 及第個0。在此例中,粗分位元結果為“100000”。第一 個“1”是在位元-及第一個“0”是在位元二。其意味著「〇Li」 可以透過LSB陣列310從“Vrp,’切換至“Vm,,以減少決定點。 因為’第七個位元結果表明其輸入可能小於“2048”且不大於 “2080” ’其「OL1」的暫存器閃鎖第八個位元結果且「⑽」的 is] 12 201138320 暫存态不改變。第八個位元結果為“〇,,其「〇L1」從“Vrp”切換 至“Vm”。然後’在第九個位元結果rL1」從“Vm,,切換至“Vrp 。因為「0L1」透過LSB陣列31〇降低決定點,第九個到第十 二個位元結果會透過傳統的二元搜尋演算法檢查輸入“ 2〇2〇”。 因此,在第一個位元的錯誤會被修復。 在第十二次之後,數位錯誤校正電路重新組成粗分及細分的 位几結果以形成“12”位元的輸出碼。其細分轉換輸出是由第七 個位元結果及第九她元至第十三個位元結果所域。假設錯誤 是由粗分轉換所產生,數位錯誤校正使用第七個及第八個位元結 果進行檢查。其數位錯誤校正電路的函式如下:Kuttner, A 1.2V l〇b 20MSamples/s Non-Binary Successive Approximation ADC in G'13um CM〇s,” lsscc Tenghua also Shangqing PP. 176-177, 2002·”, to solve the above problems and achieve high speed effect. However, in this way, it is necessary to use a complex digital controller and a temperature code array to perform and correct the error. m 3 201138320 For this reason, if you can use a non-binary search algorithm and avoid using the reset controller and temperature code capacitor array, but also allow the correction when the highest bit turns wrong, it will help to improve Conversion efficiency. In other words, it can be seen that there has been a problem of poor conversion efficiency in the prior art - and it is therefore necessary to propose an improved technical means to solve this problem. SUMMARY OF THE INVENTION The present invention is disclosed in the prior art. The present invention discloses a sub-region ratio digital conversion device and a method thereof. The analog-to-digital conversion device of the sub-interval disclosed in the present invention comprises: a capacitor array module, a conversion module, a control module and an output module. The capacitor array module receives the analog signal, and stores the analog signal in the LSB array and the array to generate a capacitor array output. The LSB array includes the first light-combining capacitor, and the array includes the second light-combining capacitor and the overlapping capacitor. And the second surface-capacitor capacitor and the overlapping capacitor are connected in parallel; the conversion module is in the preset coarse-divided (c〇arse) clock cycle, and the n-amplitude capacitor array is excited through the first device, and is read out. The electric energy column output transmission repairs to the lock comparator comparator 'and generates a coarse quantile according to the comparison result of the question lock comparator, and sets the coarse quantile to the decision signal, and the preset subdivision (4) e) clock During the period, the output of the capacitor array is amplified by the first preamplifier and the second preamplifier, and the output of the f-column of the domain is transmitted, and the subdivision bit is generated according to the comparison result of the flash lock comparator, and The subdivision bit is set as the decision signal; the control mode _ is used to determine the control set of the shouting shed, and the LSB array and the MSB array are controlled by the control logic; the female group (10) is coarsely divided by the digital error positive circuit. Bit, The result of the comparison of the subdivided bits and the overlap region is digitally corrected to produce a digital output signal. 201138320 2 It is mandatory that the capacitor array is a binary capacitor array. 4 and the size of the fresh capacitor is "2 (7 · χ) · small is Ί" unit. In addition, the large control logic of 卞(四)丝w h includes the overlapping logic and standard logic. This = wheel = control 4 capacitors, while the standard logic takes care of (4) one. Then, the control card includes the same number of LS positive and negative 与 as the overlap capacitor and is used as a register and respectively controls the corresponding overlapping capacitors, wherein each D-type flip-flop is electrically connected to the corresponding multiplexer. In this practical implementation, the conversion module calculates the overlapping area through the overlapping logic in the subdivision period of the subday cycle (four), and controls the overlapping capacitance according to the re-f area. The first preamplifier and the second preamplifier can be sent to the capture comparator through the gain control after the power amplifier array is large. The steps of the second interval ratio digital age method of the present invention include: receiving Analog signal, and the ratio of the 瓣 瓣 u 龄 ( ( ( ( 四 四 ms ms ms ms ms ms ms ms ms ms ms ms ms ms ms ms ms ms ms ms ms ms ms ms ms ms ms ms 丨 丨 丨 丨 丨 丨 丨 丨 丨 丨 丨 丨 丨 丨The output of the capacitor array is transmitted to the flash lock comparator' and the coarse quantile is generated according to the (10) taste (10) taste result, and the coarse position bit is set as the decision signal; in the preset subdivision (decay) clock cycle Through the first preamplifier And the second preamplifier amplifies the capacitor array output, and transmits the amplified capacitor array output to the latch comparator, and generates a subdivision bit according to the comparison result of the latch comparison state, and sets the subdivision bit to the decision Receiving a decision signal to execute a preset control logic, and generating an overlap region and controlling the LSB array and the MSB array through the control logic; comparing the coarse quantile, the subdivided bit, and the overlap region by a digital error correction circuit The result is digitally corrected to produce a digital output signal. 201138320 The control logic includes overlapping logic and a small amount of overlap logic used to control the overlapping capacitance 'and the standard logic _ to control the first-age capacitor and the second-side power. The size of the second recombination capacitor mentioned is "2 (7·χΜ, unit, the large J of the overlapping capacitance is 1 unit. In addition, the analog digital conversion method of the sub-interval of the present invention, includes, subdivided clock week When shooting the second clock cycle, through the overlapping logic meter, the output area 'and the weight 4 area according to the weight 4 area, and the first step amplification The second preamplifier amplifies the output of the capacitor array and transmits it to the comparator via the gain control. Then, the overlap capacitor is controlled by the same number of D-type flip-flops as the register, wherein each type is positive. The D terminal of the inverter is electrically connected to the corresponding multiplexer, and the multiplexer receives at least the decision signal. The LSB array and the MSB array form a binary capacitor array. The device and method disclosed by the present invention As above, the difference from the prior art is that the present invention is to increase the overlap capacitance in the MSB array of the capacitor array, so that the Coarse process towel reduces the correctness of the comparator. Comparing the time. The technical effect of the conversion efficiency of the new digits can be improved by the technical means of the present invention. [Embodiment] Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings and embodiments, thereby The process of applying technical means to solve technical problems and achieve technical efficacy can be fully understood and implemented. Before the analog digital conversion device and the method thereof in the sub-interval disclosed in the present invention, the nouns defined by the present invention are first described, and the overlapping capacitance (0LC) mentioned in the present invention is used. For the corrected capacitance and 201138320 is set in the MSB array 'these overlapped__ the corresponding register is controlled, the detailed circuit of the MSB array and the overlapping capacitor will be described later in conjunction with the drawing. The following is a description of the analog-to-digital conversion device and the method of the sub-interval of the present invention. First, the description will be made first for the analog-to-digital conversion device of the sub-interval of the present invention. Please refer to "1" and "1". The block diagram of the analog-to-digital conversion device of the sub-interval of the present invention comprises: a capacitor array module 1 (n, a conversion module 1〇2, a control module 103, and an output module 1〇4), wherein the capacitor array module 1 〇1 is used to receive the Lu analogy, and the capacitor array output is generated through the LSB array and the MSB array, and the LSB array includes the first coupling capacitor, and the MSB array includes the second coupling capacitor and the overlapping capacitor, and the second coupling capacitor In parallel with the overlapping capacitors, in actual implementation, the capacitor array module 101 is a binary-weighted capacitor array composed of an LSB array and an MSB array. The conversion module 102 is used for a preset coarse division ( Coarse), in the clock cycle, amplifying the output of the capacitor array through a first preamplifier (Pre-amplifier), and transmitting the amplified output of the electric Lugu array to a latch comparator, and According to the comparison result of the latch comparator, the coarse quantile is generated, and the coarse component is set as the decision signal; and in the preset sub-clock cycle, through the first preamplifier and the second The preamplifier amplifies the output of the capacitor array, and transmits the amplified capacitor array output to the latch comparator, and generates a subdivided bit according to the comparison result of the latch comparator and sets the subdivided bit as a decision signal. The digital signal that determines the signal is “〇,, or “1 '', is used to determine the calculation of the “add” or “minus” of the binary search algorithm. In practical implementation, the output of the capacitor array can pass through the first front The amplifier and the second preamplifier are divided into different stages for gain to reduce the input comparator's input-referred offset. For example, the capacitor array output is permeable during the coarse clock cycle. A first preamplifier performs gain, and during subdivided clock cycles, the capacitor array output is amplified by a first preamplifier and two second preamplifiers (ie, (b), the detailed circuit diagram of this part will be described in conjunction with the drawing. In particular, the amplified capacitor array output can be transmitted to the latch after passing through the Gain control multiplexer. The lock comparator enters the 4 lock and compares 'and the invention does not limit the number of the first preamplifier and the second preamplifier. The Lu control module 丨〇3 is used to receive the decision signal to perform the preset. The control logic controls 'and generates overlapping regions through the control logic and controls the LSB array and the MSB array. The control logic includes overlapping logic and standard logic, and the overlay logic is used to control overlapping capacitances, such as: "CL<0>==SC<N+1> | (SC<9>&〇LS<N>), CL<1 >==SC<N>", and standard logic is used to control the first coupling capacitor and the second consumption & Valley, for example, "CL<〇>== SC<N+1>, CL<1>==SC<N>", where n is a bit, CL is a control logic, SC is a state control, and 〇LS is Overlapping state, in real φ implementation, this overlapping logic is designed to be able to overlap As a result, the second sampling is performed, and the conversion module 1〇2 is provided to divide the overlapping region of the binary search algorithm through the overlapping logic in the second clock cycle period of the subdivided clock cycle, so as to control The module 103 controls the overlap capacitance based on this overlap region, while the standard logic control is the same as the conventional standard design. In particular, the present invention is not limited to the overlapping logic and the standard logic by the above examples, and the same functions composed of equivalent circuits are all of the embodiments without departing from the spirit and scope of the embodiments. Means can be implemented. In addition, the control module 1〇3 may include the same number of D-type flip-flops as the overlap capacitors as the registers and respectively control the respective overlapping capacitors. The D end of each D-type 201138320 flip-flop is electrically connected to a corresponding multiplexer, and the multiplexer receives at least the decision signal generated by the conversion module 102. The output module 104 is configured to perform digital correction on the comparison result of the coarse quantile, the subdivision-bit, and the overlap region by using a digital error correction circuit to generate a digital output signal. • In practice, use a binary search algorithm with overlapping regions to check for coarse tool bit errors and digital corrections for errors. In the binary search algorithm, a comparison process is performed in which the female inserts a heavy-duty region, and the overlap region is actually calculated by the capacitor array module 101 inputting a stable time, if a longer • stabilization time is required A large overlap area is required. As shown in FIG. 2, FIG. 2 is a flow chart of the analog-to-digital conversion method of the sub-interval of the present invention, and the steps include: receiving an analog signal, and storing the analog signal in the LSB array and the MSB array to generate Capacitor array output (step 21〇); in a predetermined coarse phase (Coarse) clock cycle, the capacitor array output is amplified by the first preamplifier, and the amplified capacitor array output is transmitted to the latch comparator, and Generating a coarse quantile according to the comparison result of the latch comparator, and setting the coarse component to a φ decision signal (step 220); in the preset sub-clock cycle, through the first front The magnifying ϋ and the second preamplifier H amplify the electric turn, and the enlarged electric material is sent to the voyage, and the root period is compared with the word comparison result. The subdivision bit is generated, and the subdivision bit is set. To determine the signal (step 23〇); receive the decision signal to implement the control touch, and generate the overlap region and control the LSB array and the MSB array through the fourth (step 24); the coarse error is determined by the digital error correction circuit Bits, subdivisions, and overlapping regions The comparison result is digitally corrected to generate a digital output signal (step 25A). In practical implementation, step 2 ι is to generate a capacitor array through the café array and the MSB array into a binary capacitor _ to generate a capacitor array input 201138320 and the LSB P train column includes a first-to-surface capacitance, the MSB array includes a second consuming capacitor and The electric grid is re-powered, and the first consumable capacitor and the overlapping capacitor are connected in parallel with each other. In addition, during the second clock cycle in the fine/knife clock cycle, the overlap region is calculated by the overlap logic and the overlap capacitance is controlled according to the overlap region (step, through the above steps, the array can be in the valley array) The overlapping capacitance is added to the MSB array to reduce the requirement of the ambiguous redness in the process of coarse division (c_e), thereby greatly reducing the comparison time of the succulent. The following is in conjunction with "3rd" to "5th". The method is as follows. Please refer to "3" and "3" for the circuit diagram of the capacitor array module to which the present invention is applied. The capacitor array module 1〇1 includes: analog signal input terminal 301 LSB P car The column 310, the first-combining capacitor 311, the MSB array 32〇, the second tapping valley 321 , the overlapping grid 322, and the capacitor array output terminal 33. In particular, the above description is '12 bits, The capacitor array is illustrated. However, the present invention does not limit the number and type of electronic components included in the capacitor array module 101. In addition, the integer capacitance 3_ can be connected between the LSB array 310 and the MSB array 32〇, For the sake of convenience Therefore, as shown in "Figure 3", it is possible to add an additional number of electric power f 300a ' to make the unit capacitance value of the integer capacitance lion maintain an integer. For example, the capacitance 30 〇a can be respectively and And "Vm,, the connection of the two light-combined capacitors, the two her combined capacitance is a total of thirty fresh capacitor values, the integer capacitance 3〇〇b through the conventional formula (for example "(32+3 〇y31=2,,. where = is the unit capacitance value of 31LS of the LSB array, “3〇, the unit electric valley value of the capacitance ^00a”, and then the integer capacitance bird is two For example, the capacitor array module 1〇1 of the “12” bit is used as an example. The capacitor array module receives 201138320 and receives the analog signal through the analog signal input terminal 301, and passes through the LSB array 31. And the MSB array 320 generates a capacitor array output, and the generated capacitor array output is output to the conversion module 1〇2 through the grid array output terminal 330, wherein the LSB array 31〇 includes a plurality of first consumable capacitors 311' And generate corresponding bits in "L1" to "L5", and MSB array 320 contains The second coupling capacitor 321 and the overlapping capacitor 322, and the second engaging capacitor 321 and the heavy capacitor 322 are connected in parallel with each other as shown in FIG. 3. As mentioned above, in the MSB array 320, the first "6" bit The element is formed by the second coupling capacitor 321 "Mx" and the overlapping capacitor 322 "OLx", wherein "x" represents the bit number. The size of the second light combining capacitor 321 "Μχ" is "Sizes". For the 2(7x)-l" unit, for example, the size of the second coupling capacitor 321 "M1" is "2 (^) - 1 = 63" unit, and the size of the second coupling capacitor 32 丨 "M2" is "2" (7·2) 1=3 i" The unit, ... and so on, the size of the second coupling capacitor 321 "M6" is "2 (7_6) - 1 = 1". Therefore, the second coupled valley 321 and the overlapping capacitor 322 of rMl+〇u" to "M6+〇L6" form a similar traditional continuous approximation; APPr〇ximati〇n, SAR) analog digital converter (Anal〇g_t〇_Digital c 〇nverter, ADC) A binary capacitor array. In the Coarse conversion process, the overlapping capacitance 322 "〇Lx" is the same as the other - the second coupling capacitance 32i "Mx" performs the binary search algorithm, and the result of the subdivision conversion of the second bit can be changed. A connection of overlapping capacitors 322 is performed to perform a repeat comparison. Taking the unary search algorithm as an example, suppose the analog input hierarchy is equivalent to the value 2020". After tracking, it is similar to the traditional continuous approximation analog digital conversion benefit, and its first decision point (Decisi〇np〇int) “2048” is half of the total wheeling range, and the control logic sets rM1” and r〇L1” to “Vrp” j 11 201138320. This causes the reference source to be changed by the large capacitive load. Therefore, the reference source and capacitor array output require coffee. stable. If the _ comparator is to resolve the bit it's when the output of the capacitor array has not been completed, it may result in a wrong decision. In this example, suppose an erroneous judgment occurs and the comparator result is "丨, instead of "〇". Keep "m" and "0L1" connected to "Vrp", "M2" and "(10)" are connected to "Vrp" to form the second decision point "3〇72". Because the residual voltage of the second decision point is large enough 'even if the output of the capacitor array is not stable, the flash lock comparator can also be cut off. "M2" and "OL2" are switched from "Vrp to Vm" after the second decision point is generated, and the remaining coarse conversion is repeated in the same manner. Starting from the seventh bit, the capacitor array output takes longer to stabilize and gain amplification; therefore, the analog circuit has high accuracy for accurate conversion. It is not allowed to have errors afterwards. After the seventh decision point is produced, the loser (ie: “2〇2〇”) is between 2143 and 2016. The seventh decision point is “2〇8〇” and the comparison is π and 〇. The seventh result indicates that the input should be between "2〇8〇,, and "2〇16", and in the coarse-segment conversion, the wrong decision is only possible at "2048". Therefore, it needs to be in the S8 position. Yuan ornaments check the material point "fibre". The seventh her-yuan comparator, the fruit makes the Μ7" gang Vrp seven knife change back to "Vm". The eighth bit result needs to be “〇”. ,, - An overlapping logic function is used to determine the first "ι and the first 0" in the coarse conversion result. In this example, the coarse quantile result is "100000". The first "1" is The bit--and the first "0" are in bit 2. This means that "〇Li" can be switched from "Vrp," to "Vm" through the LSB array 310 to reduce the decision point. Because 'the seventh bit result indicates that its input may be less than "2048" and not greater than "2080" 'its "OL1" register flash locks the eighth bit result and "(10)" is] 12 201138320 The state does not change. The eighth bit result is "〇, and its "〇L1" is switched from "Vrp" to "Vm". Then 'in the ninth bit result rL1' is switched from "Vm," to "Vrp." Since "0L1" lowers the decision point through the LSB array 31, the ninth to the twelfth bit result is checked by the conventional binary search algorithm to input "2〇2〇". Therefore, the error in the first bit will be fixed. After the twelfth second, the digital error correction circuit recomposes the coarse and subdivided bit results to form an output code of "12" bits. The subdivision conversion output is the result of the seventh bit result and the ninth to thirteenth bit result. Assuming that the error is due to a coarse-to-segment conversion, the digital error correction is checked using the seventh and eighth bit results. The function of its digital error correction circuit is as follows:

Output = Coarse*64+Fine-32 if 7th & 8th results are 00Output = Coarse*64+Fine-32 if 7th & 8th results are 00

Output = Coarse*64+Fine if 7th &8th results are 01 or l〇Output = Coarse*64+Fine if 7th &8th results are 01 or l〇

Output = Coarse*64+Fine+32 if 7th & 8th results are 11 雖然此方式需要額外的一個位元比較重疊,但是電容陣列輪 出的穩定時間在粗分轉換可大幅減少,進而導致更高的轉換速度。 接下來,請參閱「第4圖」’「第4圖」為應用本發明的轉換 杈組之電路示意圖。其轉換模組1〇2包含:輸入端4〇1、第一前置 放大器411、第二前置放大器412、閂鎖比較器43〇及輸出端44〇。 前面提到,轉換模組102可透過第一前置放大器411及第二前置 放大器412分成不同階段進行增益(即訊號放大),其實際連接方式 如「第4圖」所示意。輸入端4〇1與電容陣列模組1〇1的電容陣 列輸出端’雜連接’肋接收電料列輸出。縣,在粗分 時脈週期中僅透過第-前置放大器411放大電容陣列輸出,以達 成快速轉換的目的。之後,在細分時脈週期中再透過第一前置放 201138320 大器411及一個串連的第二前置放大器似放大電容陣列輸出, 且將放大後的電容陣m出傳送至_比較器。在實際實施 上’更*可透過二組增益控制多工器接收放大後的電容陣列輸 出,接著由這二組增益控制多工器42〇輸出至问鎖比較器以 產生決魏號,並且透過輸料_傳送難生的歧訊號至控 制模組103。 —如「第5圖」所示意,「第5圖」為細本發明的暫存器之電 路:意圖’暫存器500包含··多工器5i〇、D型正反器別及控制 1^520其中’夕工器51〇具有三條輸入線分別用以接收工作 電壓_)、與轉換模組102的輸出端電性連接,用以接收轉 換換組102所產生的決定訊號’以及與D型正反器別的〇端電 ^生連接。除此之外’多工器510的輸出線則與D型正反器530的 D端電性連接’且控制線與控制邏輯52〇電性連接。而在d型正 2 =的部分更如「第5圖」所示意連接有時脈及取樣—帅 ^實際實施上,控制邏輯剩以控制多工器训的輸入,例如: ,=L為“00”時’位元“N”狀態維持先前的狀態;當江為如 時,取樣輸出端440的決定訊號;當江為“ 1〇 則設定為“1”(即與“Vdd” f性連接 舒 器500重置為“〇,,。 *追縱時’所有暫存 前面提到,重疊電容322透過相應的暫存器5〇〇 戶㈣暫存ϋ 500即是由「第5圖」所示意的電路所組成。: 貫化上,重疊電容322與暫存_具有相同的 ^ 假設職陣列汹中的重疊電容切之數量為六個^來^ 娜的數量亦同樣為六個,每-暫存器㈣分別控制相應的重= IS1 14 201138320 容 322。 —綜上所述,可知本發明與先前技術之間的差異在於透過在電 谷陣列的MSB陣财増加重疊電容,讀雌分(c_e)過程中 '降倾比㈣的正雜要求,藉纟此—技術手段可贿決先前技 *術所存在關題’進而大幅減少比較器的比較時間, 比數位的轉換效率之技術功效。 冋 雖然本發明以前述之實關揭露如上,然其並相以限定本 發8月,任何熟習相像技藝者,在不脫離本發明之精神和範圍内, 擊當可魅許之絲觸飾,耻本發明之翻倾細須視本說 明書所附之申請專利範圍所界定者為準。 【圖式簡單說明】 第1圖為本發明次區_類比數位轉換裝置之方塊圖。 第2圖為本發a狀㈣的航數轉換方法之流程圖。 第3圖為顧本發_電料賴組之f路示意圖。 第4圖為應用本發明的轉換模組之電路示意圖。 私第5圖為應用本發明的暫存器之電路示意圖。 【主要元件符號說明】 101電容陣列模組 102轉換模組 103控制模組 1〇4輸出模組 300a湊數電容 300b整數電容 類比訊號輸入端 m 201138320Output = Coarse*64+Fine+32 if 7th & 8th results are 11 Although this method requires an extra bit to overlap, the stabilization time of the capacitor array rotation can be greatly reduced in the coarse conversion, which leads to higher Conversion speed. Next, please refer to "Fig. 4" and "Fig. 4" for a circuit diagram of a conversion group to which the present invention is applied. The conversion module 1〇2 includes an input terminal 4〇1, a first preamplifier 411, a second preamplifier 412, a latch comparator 43〇, and an output terminal 44〇. As mentioned above, the conversion module 102 can be divided into different stages for gain (i.e., signal amplification) through the first preamplifier 411 and the second preamplifier 412, and the actual connection manner is as shown in FIG. The input terminal 4〇1 and the capacitor array output terminal 'hybrid connection' of the capacitor array module 1〇1 receive the output of the electric energy column. In the county, the capacitor array output is amplified only by the pre-amplifier 411 in the coarse clock cycle to achieve fast conversion. Then, in the subdivided clock cycle, the first preamplifier is placed through the first preamplifier 201138320 and the second preamplifier is connected to the amplified capacitor array output, and the amplified capacitor array m is sent to the _ comparator. In actual implementation, the output of the amplified capacitor array can be received by the two sets of gain control multiplexers, and then outputted by the two sets of gain control multiplexers 42 to the lock comparator to generate a sigma number, and The feed_transmits the uncomfortable signal to the control module 103. - as shown in "5", "5th" is a circuit of the register of the present invention: the intention of the register 500 includes a multiplexer 5i, a D-type flip-flop and a control 1 ^520, wherein the "Holiday device 51" has three input lines for receiving the working voltage _), and is electrically connected to the output end of the conversion module 102 for receiving the decision signal generated by the conversion group 102 and D The other types of positive and negative poles are connected. In addition, the output line of the multiplexer 510 is electrically connected to the D terminal of the D-type flip-flop 530 and the control line is electrically connected to the control logic 52. In the case where the d-type positive 2 = is more like the "figure 5", it is intended to connect the time pulse and the sampling - handsome ^ actual implementation, the control logic is left to control the input of the multiplexer training, for example: , =L is " The 00" bit "N" state maintains the previous state; when Jiang is like the time, the decision signal of the output terminal 440 is sampled; when the river is "1", it is set to "1" (that is, it is connected with "Vdd" f Shushu 500 is reset to "〇,,. * When tracking" all temporary storage mentioned above, overlapping capacitor 322 through the corresponding register 5 (4) temporary storage ϋ 500 is the "figure 5" The schematic circuit is composed of: In the above, the overlapping capacitor 322 and the temporary storage _ have the same ^ The number of overlapping capacitances in the hypothetical array 为 is six, and the number of the same is six, each - The register (4) respectively controls the corresponding weight = IS1 14 201138320 322. - In summary, it can be seen that the difference between the present invention and the prior art is that the female component is read by adding the overlapping capacitance in the MSB array of the grid array ( C_e) In the process of 'falling down ratio (four) of the mixed requirements, by this - technical means can bribe the previous technology * surgery exists The title 'in turn greatly reduces the comparison time of the comparator, the technical effect of the conversion efficiency over the digital. 冋 Although the present invention is disclosed above in the above-mentioned real terms, it is intended to limit the hair of the present, any familiar artisan, Without departing from the spirit and scope of the present invention, the singularity of the singularity of the singularity of the present invention is determined by the scope of the patent application attached to the present specification. [Simplified illustration] The figure is a block diagram of the sub-region analog-to-digital conversion device of the present invention. Fig. 2 is a flow chart of the method for converting the number of digits of the a-type (four) of the present invention. Fig. 3 is a schematic diagram of the f-way of the Gubenfa_electric material group. The figure shows a circuit diagram of a conversion module to which the present invention is applied. Private figure 5 is a circuit diagram of a register to which the present invention is applied. [Main Symbol Description] 101 Capacitor Array Module 102 Conversion Module 103 Control Module 1〇 4 output module 300a to make up the number of capacitors 300b integer capacitance analog signal input terminal m 201138320

310 LSB陣列 311第一耦合電容 320 MSB陣列 321第二耦合電容 322重疊電容 330電容陣列輸出端 401輸入端 411第一前置放大器 412第二前置放大器 420增益控制多工器 430閂鎖比較器 440輸出端 5〇〇暫存器 510多工器 520控制邏輯 530 D型正反器 步驟21G接收-類比訊號’並將軸比訊賴存於一 陣列及-MSB陣列後’產生一電容陣列輸出 步驟220於預設的粗分(C。織)時脈週期中,透過一第一前 置放大器放大該電容陣列輸出,並將放大後的該 電容陣列輸出傳送至-閃鎖比較器,且根據該閃 鎖比較态的比較結果產生—粗分位元,以及將該 粗分位元設定為一決定訊號 Λ 步驟23〇於預設的細分(Fine)時脈週期中,透過該第—前置 201138320 放大器及至少一第二前置放大器放大該電容陣列 輸出,並將放大後的該電容陣列輸出傳送至該閂 鎖比較器,且根據該閂鎖比較器的比較結果產生 一細分位元,以及將該細分位元設定為該決定訊 號 步驟231在細分時脈週期中的第二個時脈週期時,透過該 重疊邏輯計算出該重疊區域’並根據該重疊區域 控制該至少一重疊電容 步驟240接收該決定訊號以執行預設的一控制邏輯,並透 過該控制邏輯產生一重疊區域及控制該LSB陣列 及該MSB陣列 步驟250透過一數位錯誤校正電路將該粗分位元、該細分 位元及該重疊區域的比較結果進行數位校正以產 生一數位輸出訊號310 LSB array 311 first coupling capacitor 320 MSB array 321 second coupling capacitor 322 overlap capacitor 330 capacitor array output 401 input 411 first preamplifier 412 second preamplifier 420 gain control multiplexer 430 latch comparator 440 output terminal 5 〇〇 register 510 multiplexer 520 control logic 530 D-type flip-flop step 21G receive - analog signal 'and store the analog signal in an array and - MSB array' to generate a capacitor array output Step 220: in a preset coarse (C. woven) clock cycle, amplifying the capacitor array output through a first preamplifier, and transmitting the amplified capacitor array output to the -flash lock comparator, and according to The comparison result of the flash lock comparison state generates a coarse division bit, and the coarse division bit is set as a decision signal. Step 23 is in a preset sub-period (Fine) clock cycle, through the first pre-position The 201138320 amplifier and the at least one second preamplifier amplify the capacitor array output, and transmit the amplified capacitor array output to the latch comparator, and generate a subdivision according to the comparison result of the latch comparator And setting the subdivision bit to the second clock cycle in the subdivided clock cycle by the decision signal step 231, calculating the overlap region by the overlap logic and controlling the at least one overlap according to the overlap region The capacitor step 240 receives the decision signal to execute a predetermined control logic, and generates an overlap region through the control logic and controls the LSB array and the MSB array step 250 to transmit the coarse quantile through a digital error correction circuit. The comparison result of the subdivided bit and the overlapping area is digitally corrected to generate a digital output signal

IS] 17IS] 17

Claims (1)

201138320 七、申請專利範圍: 1. 一種次區間的類比數位轉換裝置,包含: -電容陣職組,肋接收-類比訊號並將該類比訊 號儲存於—LSB _及—msb _後,產生—電容陣列輸 出’其中該LSB陣列包含至少一第一耗合電容,言亥MSB陣 列包含至少一第二耦合電容及至少一重疊電容,且該至少一 第二耦合電容與該至少一重疊電容相互並聯; 一轉換模組,用以於預設的粗分(c〇arse)時脈週期中, 透過-第-前置放大ϋ放大該電轉顺丨,並將放大後的 該電容陣顺出傳送至—⑽比較器,且根據朗鎖比較器 的比較結果產生-粗分位元’並將該粗分位元設定為一決定 訊號,以及於預設的細分(Fine)時脈週期甲,透過該第一前 置放大器及至少一第二前置放大器放大該電容陣列輸出,並 將放大後的該電容陣列輸出傳送至該閂鎖比較器,且根據該 問鎖比較H的比較絲產生-細分位元,並觸細分位元設 定為該決定訊號; 一控制模組,用以接收該決定訊號以執行預設的一控制 邏輯,並透過該控制邏輯產生一重疊區域及控制該LSB陣 列及該MSB陣列;及 一輸出模組’用以透過一數位錯誤校正電路將該粗分位 元、δ亥細分位元及該重疊區域的比較結果進行數位校正以產 生一數位輸出訊號。 2.如申請專利範圍第丨項所述之次區間的類比數位轉換裝 置,其t該電容陣列模組為二進位式電容陣列。 18 201138320 3. :ί(申明專利範圍第1項所述之次區間的類比 置’其中該至少-第二鶴合電容社 ^裝 少一重叠電容的大小Μ單位。 1早位’该至 4. 如申請專利範圍第i項所述之次區間的類 置’其中該控觸輯包含—重疊賴 、裝 邏輯用以控制該至少—重疊電容,;邏輯’該重疊 少一第一綱姆肋控制該至 & =申=專利範圍第!項所述之次區間的類比數位轉換裝 ,、中雜細組包含與該至少—重疊餘_㈣ 6 =反主^用以作為暫存器並分別控制相應的該重疊電容。 明專利靶圍第5項所述之次區間的類 各該D型正反器之D端與相應的-多工器= λ 接,该夕工器至少接收該決定訊號。 ^申::利範圍第i項所述之次區間的類比數位轉換裝 士厂中雜換模組在細分時脈週期中的第二個時脈週期 =,透過遠重疊邏輯計算出該重叠區域,並根據該 控制該至少一重疊電容。 埤 如申請專利範圍第!項所述之次區間的類比 其中該第-前置放大器及該至少—第二前置放大器^ 電容陣列輸出放大後,透過至少—增雜制多卫 閂鎖比較器。 系 一種次區間的類比數位轉換方法,該方法包括: 接收一類比訊號,並將該類比訊號儲存於—lsb陣列 及一 MSB陣列後,產生一電容陣列輸出; 9 201138320 於預設的粗分(Coarse)時脈週期中,透過一第一前置放 大杰放大該電容陣列輸出,並將放大後的該電容陣列輸出傳 送至一閂鎖比較器’且根據該閂鎖比較器的比較結果產生一 , 粗分位元,以及將該粗分位元設定為一決定訊號; . 於預設的細分(Fine)時脈週期中,透過該第一前置放大 益及至少一第二前置放大器放大該電容陣列輸出,並將放大 後的該電容陣列輸出傳送至該閂鎖比較器,且根據該閂鎖比 較器的比較結果產生一細分位元,以及將該細分位元設定為 _ 該決定訊號; 接收該決定訊號以執行預設的一控制邏輯,並透過該控 制邏輯產生-重疊區域及控制該LSB陣列及該MSB陣列; 及 透過一數位錯誤校正電路將該粗分位元、該細分位元及 该重疊區域的比較結果進行數位校正以產生一數位輸出吨 號。 。 Φ 1〇’如申凊專利範圍第9項所述之次區間的類比數位轉換方 去,其中該控制邏輯包含一重疊邏輯及一標準邏輯,該重疊 邏輯肋控制至少-重疊電容’該標準邏輯用以控制至少一 - 第一耦合電容及至少一第二耦合電容。 u.如申清專利範圍第10項所述之次區間的類比數位轉換方 法’其中該方法更包含在細分時脈週期中的第二個時脈週期 時,透過該重疊邏輯計算出該重疊區域,並根據該重叠區域 控制該至少一重疊電容的步驟。 12.如申請專利範圍第K)項所述之次區間的類比數位轉換方 20 [ S] 201138320 〃中^亥至少一第二輕合 少—重#電容的大小為I單位。小為單位,該至 如申請專利範圍第u項所述 法,其中該至少1疊電容 ^的類比數位轉換方 型正反器進行控制。 *“、暫存器的相同數量之D 申料·_13項職之趨 法,其令各該D型正反器之D端 1數位轉換方 接,該多工器至少接收該決定訊號:應、的—多工器電性連201138320 VII. Patent application scope: 1. A sub-interval analog-to-digital conversion device, including: - Capacitor array, rib receiving-analog signal and storing the analog signal in -LSB _ and -msb _, generating - capacitor Array output 'where the LSB array includes at least one first consumable capacitance, the MSB array includes at least one second coupling capacitor and at least one overlapping capacitor, and the at least one second coupling capacitor and the at least one overlapping capacitor are connected in parallel with each other; a conversion module for amplifying the electrical rotation through the -first preamplifier during a predetermined coarse clock period, and transmitting the amplified capacitor array to the - (10) a comparator, and generating a coarse register based on the comparison result of the lock master and setting the coarse component as a decision signal, and in a preset subdivision (Fine) clock cycle A, through the A preamplifier and at least one second preamplifier amplify the output of the capacitor array, and transmit the amplified output of the capacitor array to the latch comparator, and compare the H of the comparison lock according to the Q lock to generate a subdivision bit And the touch sub-division bit is set as the decision signal; a control module is configured to receive the decision signal to execute a preset control logic, and generate an overlap region and control the LSB array and the MSB array through the control logic And an output module 'for performing a digital correction on the comparison result of the coarse quantile, the delta subdivision bit, and the overlap region through a digital error correction circuit to generate a digital output signal. 2. The analog-to-digital conversion device of the sub-interval described in the scope of the patent application, wherein the capacitor array module is a binary capacitor array. 18 201138320 3. :ί (Representation of the analogy of the sub-interval described in item 1 of the patent scope) where the at least-second Hehe capacitors are installed with less than one overlapping capacitor size. 1 early position 'to 4 The class of the sub-interval described in item i of the patent application scope, wherein the control touch contains - overlapping, the logic is used to control the at least - overlapping capacitance; the logic 'the overlap is less than the first one Controlling the analog-to-digital conversion package of the sub-interval described in the &=== patent scope item!, the middle-mechanical group contains the at least-overlapped residual_(four) 6=reverse master^ is used as a temporary register The respective overlapping capacitors are respectively controlled. The D-end of each of the D-type flip-flops in the sub-interval described in Item 5 of the patent target is connected with the corresponding-multiplexer=λ, and the evening device receives at least the Determining the signal. ^申:: The second time interval in the subdivision clock cycle of the analogy digital converter in the sub-interval of the sub-interval described in item i of the interest range =, calculated by far overlapping logic The overlapping area, and controlling the at least one overlapping capacitance according to the control. The analogy of the sub-interval described in the scope of the patent item is that the output of the first preamplifier and the at least-second preamplifier capacitor array is amplified, and at least the polynomial latch comparator is added. The analog-to-digital conversion method of the sub-interval includes: receiving an analog signal and storing the analog signal in the -lsb array and an MSB array to generate a capacitor array output; 9 201138320 in a preset coarse coordinate (Coarse) In the clock cycle, the capacitor array output is amplified by a first preamplifier, and the amplified capacitor array output is transmitted to a latch comparator 'and a result is obtained according to the comparison result of the latch comparator. a quantile, and setting the coarse quantile as a decision signal; in a preset sub-clock cycle, the capacitor is amplified by the first preamplifier and the at least one second preamplifier Array output, and transmitting the amplified capacitor array output to the latch comparator, and generating a subdivided bit according to the comparison result of the latch comparator, and dividing the subdivision Set to _ the decision signal; receive the decision signal to execute a preset control logic, and generate the overlap region and control the LSB array and the MSB array through the control logic; and pass the coarse division through a digital error correction circuit The comparison result of the bit element, the subdivided bit and the overlapping area is digitally corrected to generate a digital output tonne. Φ 1〇', as in the analogy digital conversion of the sub-interval described in claim 9 of the patent scope, The control logic includes an overlay logic and a standard logic. The overlay logic rib controls at least the overlap capacitor. The standard logic controls the at least one first coupling capacitor and the at least one second coupling capacitor. u. If the analog digital conversion method of the sub-interval described in item 10 of the patent scope is applied, wherein the method further includes the second clock cycle in the subdivided clock cycle, the overlapping region is calculated by the overlapping logic. And controlling the at least one overlapping capacitance according to the overlapping area. 12. The analog-to-digital conversion side of the sub-interval described in item K) of the patent application 20 [S] 201138320 〃中^海At least one second light-to-weight# The size of the capacitor is I unit. The unit is small, and the method is as described in claim 5, wherein the at least one stack of capacitances is controlled by an analog digital conversion type flip-flop. * ", the same number of D register of the register, _13 job, which makes the D-terminal 1 digit conversion of each D-type flip-flop, the multiplexer receives at least the decision signal: Multiplexer electrical connection 15. Γ=1 利範圍第9項所述之次區間的類比數位轉換方 法,討該LSB陣列及該MSB陣列組成二進位式電容陣列。 16. 如申请專利範圍第9項所述之次區間的類比數位轉換方 法,其中該第-前置放大器及該至少_第二前置放大器將該 電容陣列輸出放大後’透過至少一增益控制多工器傳送至該 閂鎖比較器。15. Γ=1 The analog digital conversion method of the sub-interval described in item 9 of the range is to discuss that the LSB array and the MSB array form a binary capacitance array. 16. The analog-to-digital conversion method of the sub-interval described in claim 9 wherein the first preamplifier and the at least second preamplifier amplify the output of the capacitor array by transmitting at least one gain control The tool is transferred to the latch comparator. 21 [S]21 [S]
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