TW201131894A - Methods for reducing near-field radiation and specific absorption rate (SAR) values in communications devices - Google Patents

Abstract

A method is provided for reducing near-field radiation and specific absorption rate (SAR) values in a communications device. The communications device includes a multimode antenna structure transmitting and receiving electromagnetic signals and circuitry for processing signals communicated to and from the antenna structure. The antenna structure includes: a plurality of antenna ports operatively coupled to the circuitry; a plurality of antenna elements, each operatively coupled to a different one of the antenna ports; and one or more connecting elements electrically connecting the antenna elements at a location on each antenna element that is spaced apart from an antenna port coupled thereto to form a single radiating structure and such that electrical currents on one antenna element flow to a connected neighboring antenna element and generally bypass the antenna port coupled to the neighboring antenna element, the electrical currents flowing through the one antenna element and the neighboring antenna element being generally equal in magnitude, such that an antenna mode excited by one antenna port is generally electrically isolated from a mode excited by another antenna port at a given desired signal frequency range and the antenna structure generates diverse antenna patterns. The method includes adjusting the relative phase between signals fed to neighboring antenna ports of the antenna structure such that a signal fed to the one antenna port has a different phase than a signal fed to the neighboring antenna port to provide antenna pattern control and to increase gain in a selected direction toward a receive point. The method features using a transmit power lower than the transmit power used in a non-pattern control operation of the antenna structure such that the communications device obtains generally equivalent wireless link performance with the receive point using reduced transmit power compared to the non-pattern control operation, thereby reducing the specific absorption rate.

Description

201131894 VI. Invention Description: [No. Partial continuation of Case No. 750,196, US Patent Application No. 12/750,196 is the United States filed on April 8, 2008 under the name "Multimode Antenna Structure" (issued as US Patent No. 7,688,273) The continuation of the patent application No. 12/099, 320, and the US patent application No. 12/099, 320 is June 27, 2007. The name of the application is "Multimode Antenna Structure" (issued as US Patent No. 7,688,275) Part of the continuation of U.S. Patent Application Serial No. 11/769,565, and U.S. Patent Application Serial No. 11/769,565 is based on U.S. Provisional Patent entitled "Multimode Antenna Structure" filed on April 20, 2007. Application No. 60/925,394 and the name of the application filed on May 8, 2007 are also US Provisional Patent Application No. 60/916 of "Multimode Antenna Structure". Case No. 655. This application is also based on US Provisional Patent Application No. 61/181,176, which was also filed on May 26, 2009 and is also entitled "Multimode Antenna Structure". Each of the above-identified applications is hereby incorporated by reference. The present invention is directed to a method for reducing near field radiation and specific absorption values in a communication device. BACKGROUND OF THE INVENTION The present invention relates generally to wireless communication devices and, more particularly, to methods for reducing near field light shot and specific absorption rate (sar) values in such devices. Many communication devices have multiple antennas that are tightly sealed (e.g., less than four-quarters apart) and that can operate simultaneously in the same frequency band. Common examples of such communication devices include portable communication products such as cellular phones, personal digital assistants (PDAs), and data cards for wireless network devices or personal computers (PCs) (such as multi-input and output (ΜΙΜΟ )) and mobile wireless communication | standard protocols such as wireless LAN 8〇2·11η and such as 8〇2_1ΜΜΜΑΧ), Η_Α and 1xEvd (^3g data communication) require multiple antennas to operate simultaneously. SUMMARY OF THE INVENTION Overview of Embodiments of the Invention According to one or more embodiments, a method for reducing near field radiation and specific absorption rate (SAR) values in a communication device is provided. The communication device includes a multimode antenna structure for transmitting and receiving electromagnetic signals, and circuitry for processing signals transmitted to the antenna structure and signals from the antenna structure. The antenna structure includes: a plurality of antennas operatively coupled to the circuit; a plurality of antenna elements, each antenna element being operatively coupled to a different one of the antennas; and one or more a plurality of connecting elements electrically connecting the corresponding antenna elements at a position on each of the antenna elements, each antenna element being separated by an antenna 耦 coupled thereto to form a single radiating structure, and causing an antenna element The current flows to a connected adjacent antenna element of 201131894, and typically does not flow to the antenna 被 coupled to the adjacent antenna element, and flows through the one antenna element and the adjacent antenna element The currents are generally equal in magnitude, such that an antenna pattern excited by one antenna 在一 within a given desired signal frequency range is generally electrically isolated from a pattern excited by another antenna ,, and the antenna structure is generated Diversity antenna field type. The method includes adjusting a relative phase between signals fed to adjacent antennas of the antenna structure such that a signal fed to the one antenna is compared to a signal fed to the adjacent antenna The signal has a different phase to provide antenna field control and increase the gain in a selected direction toward a receiving point. The method is characterized by using a transmission power lower than the transmission power used in a non-field type control operation of the antenna structure, such that the communication device acquires the transmission power using the transmission power lower than the non-field type control operation The receiving point is roughly equivalent to the wireless link performance, thereby reducing the SAR. In accordance with one or more additional embodiments, a method for reducing near field radiation and specific absorption rate (SAR) values in a communication device is provided. The communication device includes an antenna array for transmitting and receiving electromagnetic signals, and circuitry for processing signals transmitted to the antenna array and signals from the antenna array. The antenna array includes a plurality of radiating elements, each radiating element having an antenna port operatively coupled to the circuit. The method includes adjusting a relative phase of signals fed to the antennas of the antenna array such that a signal fed to one antenna has a signal compared to a signal fed to another antenna A different phase provides antenna field control and increases the gain in a selected direction toward a receiving point. The method 6 201131894 is characterized in that a transmission power lower than the transmission power used in a non-field type control operation of the antenna array is used, so that the communication device acquires using a transmission power lower than the non-field type control operation. The wireless link performance is approximately equivalent to the receiving point, thereby reducing the specific absorption rate. BRIEF DESCRIPTION OF THE DRAWINGS Figure 1A illustrates an antenna structure having two parallel dipoles. Figure 1B illustrates the current generated by a dipole excitation in the antenna structure of Figure 1A. Figure 1C illustrates a model corresponding to the antenna structure of Figure 1A. Fig. 1D is a diagram illustrating the scattering parameters of the antenna structure of Fig. 1C. Figure 1E is a diagram illustrating the current ratio of the antenna structure of Figure 1C. Fig. 1F is a diagram illustrating the gain pattern of the antenna structure of Fig. 1C. Figure 1G is a diagram illustrating the envelope correlation of the antenna structure of Figure 1C. Figure 2A illustrates an antenna structure connected to two parallel dipoles through a connecting element in accordance with one or more embodiments of the present invention. Figure 2B illustrates a model corresponding to the antenna structure of Figure 2A. Figure 2C is a diagram illustrating the scattering parameters of the antenna structure of Figure 2B. Figure 2D is a diagram illustrating the scattering parameters of the antenna structure of Figure 2B with lumped element impedance matching at the two turns of the antenna structure. Figure 2E is a diagram illustrating the current ratio of the antenna structure of Figure 2B. 201131894 Figure 2F is a diagram illustrating the gain field pattern of the antenna structure of Figure 2B. Figure 2G is a diagram illustrating the envelope correlation of the antenna structure of Figure 2B. Figure 3A illustrates an antenna structure connected to two parallel dipoles by meandering connecting elements in accordance with one or more embodiments of the present invention. Fig. 3B is a diagram showing the scattering parameters of the antenna structure of Fig. 3A. Figure 3C is a diagram illustrating the current ratio of the antenna structure of Figure 3A. Figure 3D is an illustration of the gain field pattern of the antenna structure of Figure 3A. Figure 3E is a diagram illustrating the envelope correlation of the antenna structure of Figure 3A. Figure 4 illustrates an antenna structure of a ground or ground network in accordance with one or more embodiments of the present invention. Figure 5 illustrates a balanced antenna structure in accordance with one or more embodiments of the present invention. Figure 6A illustrates an antenna structure in accordance with one or more embodiments of the present invention. Figure 6B is a diagram showing the scattering parameters of a particular dipole width antenna structure of Figure 6A. Figure 6C is a diagram showing the scattering parameters of the antenna structure of another dipole width in Fig. 6A. Figure 7 illustrates an antenna structure fabricated on a printed circuit board in accordance with one or more embodiments of the present invention. 8 201131894 Figure 8A illustrates an antenna structure with apex resonance in accordance with one or more embodiments of the present invention. Fig. 8B is a diagram illustrating the scattering parameters of the antenna structure of Fig. 8A. Figure 9 illustrates a tunable antenna structure in accordance with one or more embodiments of the present invention. 10A and 10B illustrate an antenna structure having connecting elements pointing in different positions along the length of the antenna element in accordance with one or more embodiments of the present invention. Figures 10C and 10D are diagrams illustrating the scattering parameters of the antenna structures of Figures 1A and 10B, respectively. The first diagram illustrates an antenna structure including a connecting element having a switch in accordance with one or more embodiments of the present invention. Figure 12 illustrates an antenna structure having a connection element to which a filter is coupled, in accordance with one or more embodiments of the present invention. Figure 13 illustrates an antenna structure having two connecting elements, some of which are coupled to the connecting elements, in accordance with one or more embodiments of the present invention. Figure 14 illustrates an antenna structure having an adjustable frequency connection element in accordance with one or more embodiments of the present invention. Figure 15 illustrates an antenna structure mounted on a PCB assembly in accordance with one or more embodiments of the present invention. Figure 16 illustrates another antenna structure mounted on a PCB assembly in accordance with one or more embodiments of the present invention. 201131894 Figure 17 illustrates an alternative antenna structure that can be mounted on a PCB assembly in accordance with one or more embodiments of the present invention. Figure 18A illustrates a three mode antenna structure in accordance with one or more embodiments of the present invention. Figure 18B is a diagram illustrating the gain pattern of the antenna structure of Figure 18A. Figure 19 illustrates an antenna and power amplifier combiner application of an antenna structure in accordance with one or more embodiments of the present invention. 20A and 20B illustrate a multimode antenna structure that can be used, for example, in a WiMAX USB or ExpressCard/34 device, in accordance with one or more additional embodiments of the present invention. Fig. 20C illustrates a test combination used to measure the performance of the antennas of Figs. 20A and 20B. Figs. 20D to 20J illustrate test results of the antennas of Figs. 20A and 20B. 21A and 21B illustrate a multimode antenna structure that can be used, for example, in a WiMAX USB server dongle, in accordance with one or more alternative embodiments of the present invention. 22A and 22B illustrate a multimode antenna structure that can be used, for example, in a WiMAX USB server key, in accordance with one or more alternative embodiments of the present invention. Figure 23A illustrates a test combination used to measure the antenna performance of Figures 21A and 21B. Figure 2 3 B to 2 3 K illustrates the test measurements of the antennas of Figures 21A and 21B. 10 201131894 Results. Figure 24 is a schematic block diagram of an antenna structure having a beam steering mechanism in accordance with one or more embodiments of the present invention. Figures 25A through 25G illustrate the test measurements of the antenna of Figure 25A. Figure 26 illustrates the gain advantage of an antenna structure as a function of the phase angle difference between the feed points, in accordance with one or more embodiments of the present invention. Figure 2 7A is a schematic diagram illustrating the structure of a simple dual-band branch monopole antenna. Figure 27B illustrates the current distribution in the antenna structure of Figure 27A. Figure 27C is a schematic diagram illustrating a spurline band stop filter. Figures 27D and 27E are test results illustrating frequency suppression in the antenna structure of Figure 27A. Figure 28 is a schematic diagram showing the structure of a band-stopped antenna in accordance with one or more embodiments of the present invention. Figure 29A illustrates an alternate antenna structure having a resisting slot in accordance with one or more embodiments of the present invention. Figures 29B and 29C illustrate test measurement results for the antenna structure of Figure 29A. Figure 30 illustrates an exemplary USB server key with a two-turn antenna configuration for field control applications in the 1900 MHz band.

Figure 31 illustrates the SAR value determined by simulating the device of Figure 30. [Embodiment] 1 11 201131894 DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS In accordance with various embodiments of the present invention, a multimode antenna structure is provided for transmitting and receiving electromagnetic signals in a communication device. The communication devices include circuitry for processing signals to and from an antenna structure. The antenna structure includes a plurality of antennas operatively coupled to the circuit and a plurality of antenna elements, each antenna element being operatively coupled to a different antenna. The antenna structure also includes one or more connection elements electrically coupled to the antenna elements such that an antenna pattern excited by one antenna 在一 in a given signal frequency range is generally associated with a pattern excited by another antenna 埠Electrically isolated. In addition, the antenna pattern produced by the pupils exhibits well-defined field diversity with low correlation. Antenna structures in accordance with various embodiments of the present invention are particularly useful in communication devices that require multiple antennas to be closely packed together (e.g., less than a quarter wavelength apart) 'including more than one of the antennas being used simultaneously And especially devices that are used simultaneously in the same frequency band. Common examples of devices in which such antenna structures can be used include portable communication products such as cellular handsets, PDAs, and wireless network devices or Pc data cards. These antenna structures are standard protocols such as 系统IΜ Ο system architecture and mobile wireless communication devices that require multiple antennas to operate simultaneously (such as 802.11n for wireless LAN and 3G data such as 8〇2.16e(WiMAX), HSDPA, and lxEVDO). It is also especially useful in communication). The 1A-1G diagram illustrates the operation of an antenna structure 100. The eighth diagram schematically illustrates an antenna structure 100 having two parallel antennas, particularly parallel dipoles 1〇2, 1〇4 of length L, which are separated by a distance (1) Separated, and 12 201131894 is not connected through any connecting elements. The dipoles 1G2, 1G4 have a fundamental resonant frequency approximately corresponding to L = X/2. Each dipole is connected to operate at the same frequency - Independent transmit/receive system. The system connection can have the same characteristic impedance z〇 for both antennas, in this case 50 ohms. When a dipole is transmitting a signal, the signal transmitted through the dipole Some of them will be directly coupled into adjacent dipoles. The maximum amount of coupling occurs around the half-wave resonance frequency of the individual dipoles and increases with a smaller separation distance d. For example, for d<;a/3, large coupling amplitude = 0.1 or -10dB 'For γλ/8, the coupling amplitude is greater than _5. When a dipole is transmitting a signal, some of the signals transmitted through the dipole Will be directly coupled to the adjacent dipole. Maximum amount of coupling through the sling Near the half-wave resonance frequency of the individual dipoles, and increases with a smaller separation distance d. For example, for d<w3, the coupling amplitude is greater than 0·1 or -10 dB 'for d<X/8' _ Width is greater than _5 (^. It is expected that there is no coupling (ie 'complete isolation') or reduce the coupling between the antennas. For example, if the coupling is _1 〇 (18, then 1% The transmission power is lost because the amount of power is directly coupled into the adjacent antenna. There may also be unfavorable system effects, such as being connected to the adjacent antenna, receiving saturation and reducing sensitivity or being connected. The performance of the transmitter to the adjacent antenna is degraded. The induced current on the adjacent antenna distorts the gain pattern compared to the gain pattern generated by the individual dipole. This effect is known to be reduced by the even The correlation between the gain field patterns generated by the poles. Therefore, although the coupling can provide some field type diversity, it has the disadvantageous system as described above = 13 201131894

o These antennas do not function independently due to tight coupling' and can therefore be considered an antenna system with two pairs of terminals or ports corresponding to two different gain field types. The use of 任-埠 essentially involves the entire structure including two dipoles. The parasitic excitation of adjacent dipoles enables diversity to be achieved at close dipole spacing, but the current that is excited on the dipole is transmitted via the source impedance, indicating mutual coupling between the turns. Fig. 1C illustrates a model dipole pair for simulation corresponding to the antenna structure 1 所示 shown in Fig. 1. In this example, the dipoles 102, 104 have a square cross section of 1 mm x 1 mm and a length (L) of 56 mm. These sizes produce a center resonance frequency of 2.45 GHz when attached to a 50 ohm source. The free space wavelength at the s Xuan frequency is 122 mm. A plan view of the scattering parameters S11 and S12 of a 10 mm or approximately λ/12 separation distance (d) is shown in Fig. 1D. Due to symmetry and reciprocity, S22 = S11, S12 = S21. For the sake of simplicity, only S11 and S12 are shown and discussed. In this combination, the coupling between the dipoles represented by S12 reaches a maximum of -3.7 dB. Figure 1E shows the ratio of the vertical current on the dipole 10 4 of the antenna structure to the vertical current on the dipole 1 〇 2 under the condition that the 埠 106 is excited and the 埠 108 is passively terminated (marked as " Amplitude Ι2/ΙΓ). The current ratio (dipole 104/dipole 102) is a frequency at a maximum corresponding to a frequency having a phase difference of 180 degrees between the dipole currents and is only slightly higher in frequency. The frequency at the maximum coupling point shown in the 1D graph. Figure 1F shows the azimuth gain field at several frequencies excited by 埠106. The 201131894 type is more omnidirectional and is due to the changing coupling scarf. Field value and phase change with frequency. Due to symmetry, the % type generated by the excitation of 埠108 will be the image of the field pattern excited by 埠106. Therefore, the field pattern is uneven from left to right, and its gain amplitude The value is more variable. The calculation of the correlation coefficient between the field types provides a quantitative description of the field diversity. The 1G graph shows the correlation between the 埠106 and 埠1〇8 antenna field types. Sex is much lower than the ideal dipole model through Clark Predicted correlation. This is due to differences in the field pattern introduced by mutual coupling. 2A-2F illustrates the operation of an exemplary two-turn antenna structure 200 in accordance with one or more embodiments of the present invention. The two-turn antenna structure 200 includes two spaced apart resonant antenna elements 202, 204 and provides low field type correlation and low coupling between the turns 206, 208. Figure 2A schematically illustrates the two turns antenna structure 200. The structure is similar to the antenna structure 1'' including the dipole pair shown in FIG. 1B but additionally includes a level between the dipoles on each side of 埠2〇6, 2〇8 Conductive connection elements 21, 212. The two turns 206, 208 are located at the same position as the antenna structure of Figure 1. When a turn is excited, the combined structure exhibits a similar resonance to the structure without the attached dipole pair , but the coupling is significantly reduced, and the field diversity is significantly increased. An exemplary model of an antenna structure 2 有一 with a 10 mm dipole spacing is shown in Figure 2B. The structure generally has the same as shown in Figure 1C. Antenna structure 100 with the same geometry But with two additional horizontal connection elements 210, 212 electrically connecting the antenna elements and slightly above and slightly below the turns. The structure is shown at the same frequency as the unattached dipoles - 15 201131894 Strong resonance, but there are very different scattering parameters as shown in Figure 2C. There is a deep detachment (dr〇p-〇ut) (below -20dB) in the coupling, and there is a drift in the input impedance, as by As indicated by S11. In this example, the optimal impedance matching (sii minimum) does not coincide with the minimum coupling (S12 minimum). A matching network can be used to improve input impedance matching and further reach as in 2D. The low coupling shown in the figure. In this example, a lumped component matching network contains a series of inductors, and then a shunt capacitor is added between each turn and the structure. Figure 2E shows the ratio between the current on the dipole element 2〇4 and the current on the dipole element 202 generated by the excitation of 埠206 (identified as "amplitude Ι2/ΙΓ" in this figure). Below the resonant frequency, this current is actually greater than the current on the dipole element 204. Near the resonance, as the frequency increases, the current on the dipole element 220 decreases relative to the current on the dipole element 2 〇2. The minimum coupling point (2 44 GHz in this case) occurs near the frequency where the currents on the two dipole elements are substantially equal in magnitude. At this frequency, the current on the dipole element 204 The phase is less than about 160 degrees behind the phase of the current on the dipole element 202. Unlike the dipole of the unconnected element of Figure 1C, the current on the antenna element 204 of the combined antenna structure 200 of Figure 2B is not forced through the termination impedance of the 埠208. Conversely, a resonant mode is generated in which current flows from the antenna element 204, through the connecting elements 210, 212, and then up to the antenna element 202, as indicated by the arrows shown in Figure 2A. (Note that this current represents half of the resonance In the other half of the resonance period, the current direction is reversed. The resonance mode of the combined structure has the following characteristics: (1) The current on the antenna element 204 is large 16 201131894 The portion does not flow to the 埠 208, thereby allowing the 埠 206 Higher isolation between 208, and (2) the magnitudes of the currents on the two antenna elements 202, 204 are approximately equal, which allows for different and uncorrelated gain patterns, as described in further detail below. The magnitudes of the currents on the antenna elements are nearly equal, and a more directional field pattern is produced (as shown in Figure 2F) than in the case of antenna structure 100 with uncoupled dipoles in Figure 1C. When the currents are equal, the condition that the field type is zero in the x (or phi = 0) direction is that the phase of the current on the dipole 204 lags behind the phase of the current on the dipole 202 by π-kd (where 1 ί = 2 π) /λ, λ is the effective wavelength.) In this case, the magnetic field propagating from the dipole 204 in the phi=0 direction will be 180 degrees from the phase of the dipole 202, so the combination of the two will be in phi There is a zero value in the =0 direction. The model example in Figure 2B , d is 10mm or an effective electrical length λ/12. In this case, kd is equal to π/6 or 30 degrees, so for phi=0 is zero, Phi=180 has one of the maximum gains of directional azimuthal radiation field. The condition of the type is that the current on the dipole 204 lags the current on the dipole 202 by 150 degrees. At resonance, the currents approach the passing condition (as shown in Figure 2E), which explains the directionality of the field pattern. In the case of dipole 204 excitation, the radiation pattern is a mirror image of those radiation patterns relative to the 2F map, and the maximum gain is in the phi=0 direction. As shown in Figure 2G, the two pupils are generated. The difference in the antenna pattern has an associated low prediction envelope correlation. Therefore, the combined antenna structure has two turns that are isolated from each other and produce a gain field with low correlation. Therefore, the frequency response of the coupling depends on the 17 201131894 characteristics of the connecting elements 210, 212 'including its impedance and electrical length. The frequency and bandwidth at which a desired isolation amount can be maintained in accordance with one or more embodiments of the present invention is controlled by suitably assembling the connecting elements. The method of grouping and connecting is to change the physical length of the connecting element. An example of this is shown by the multimode antenna structure 300 of Figure 3A, in which a meander is added to the cross-connect of the connecting elements 310, 312. ~ 〇 This has the general effect of increasing the electrical length and impedance of the connection between the two antenna elements 302, 304. The performance characteristics of this structure include the scattering parameters, current ratio, gain pattern, and field type correlation as shown in 3B, 3C, 3D, and 3E, respectively. In this embodiment, changing the length of the body does not significantly change the resonant frequency of the structure, but S12 has a significant change, with a larger bandwidth and a larger minimum than the structure without the meandering portion. It is therefore possible to optimize and improve the isolation performance by changing the electrical characteristics of the connecting elements. An exemplary multimode antenna structure in accordance with various embodiments of the present invention can be designed from a ground or ground grid 402 (as shown by antenna structure 400 in FIG. 4), or as a balanced structure (eg, through FIG. 5) The antenna structure 500 in the display is activated. In either case, each antenna structure includes two or more antenna elements (402, 404 in FIG. 4, and 502, 504 in FIG. 5) and one or more conductive connection elements (Fig. 4) 4〇6 in the middle, and 506, 5〇8) in Fig. 5. For ease of explanation, only one two-inch structure is illustrated in this example diagram. However, it is possible to extend the structure to include more than two turns in accordance with various embodiments of the present invention. A missing link η connection to the antenna structure or 埠 (418, 412 in Figure 4, and 510, 512 in Figure 5 18 201131894) is provided at each antenna element. The connecting elements provide electrical connections between the two antenna elements at the frequency or within the frequency range of interest. Although the antenna is a structure physically or electrically, the direct operation can be interpreted by considering it as two separate days ",,,. For a line structure such as a one-day line 100 that does not include a connection element line structure, the structure 106 of the structure can be said to be connected to the antenna 1〇4.纨—, person... and 'in the case of such a combined structure such as the antenna structure 400, 埠418 β +fe/1 is broken to be associated with an antenna pattern, and 埠412 may be referred to as another antenna^(4) The components are designed to resonate at the expected age frequency or frequency range. When the antenna element has an electrical length of one of the wavelengths of the Rtb., the lowest resonance occurs. Therefore, in the case of a non-equilibrium combination, the design of a component of a soap is an eight-mode model—wavelength unipolar 0 uses a higher order 様 "b. For example, the structure formed by a quarter-wavelength monopole also exhibits a high mode barrier with a dual mode line. , % times the fundamental frequency - the frequency _ Russia again. Thus, higher order modes can be developed to produce a more than 9 Γ / 2 in the same 'in-balanced combination, the antenna elements can, however, be complementary quadruplets in the mid-wavelength feed dipole - wavelength element. The complex structure can also be formed by an antenna element of resonance = type at a desired frequency or frequency range. Other possible antenna element sets are packaged. Zigzag shape limits:: rotating coil, wide-band planar shape, chip antenna, form. 'And inductive shunting such as Planar Inverted F Antenna (PIFA) according to the invention> The element does not have an antenna structure of the same structure or antenna type of the same type. 19 201131894 = The line elements should each have one or more embodiments in accordance with the present invention in a desired range of operating frequency ratios, one element having the same geometry. This is simple for design: it is preferable to 'in particular for the county The connection: The demand is the same. The bandwidth and resonance solution of the line structure of "Life &" can be controlled by the antenna element f width and resonance frequency. Therefore, a wider bandwidth component can be used for Γ = 6A, and the pattern of the combined structure illustrated in Figure 6C produces a wider bandwidth. Fig. 6A illustrates a multimode antenna structure _ including two transmission elements _, _ connected to dipoles 602, 604. The haptics 602, 604 each have a width (w) and a length (6) and are spaced apart by a distance (4). Figure 6B illustrates scattering parameters for structures having the following exemplary sizes, where W = 1 mm, L = 57.2 mm, and (iv) _. The πth diagram illustrates scattering parameters for structures having the following exemplary sizes, center w, leg, L, .4 mm, and d = 1 mm. As shown, increasing W' from "to 1" and usually maintaining another size (4) produces a wider isolation bandwidth and impedance bandwidth of the antenna structure. Increasing the isolation between the antenna elements increases the isolation bandwidth and impedance bandwidth of an antenna structure. Typically, the connecting element is in the high current region of the combined resonant structure. Therefore, it is preferable to have a high conductivity for the -connecting element. If they are operated as separate antennas, they will be located at the feeding point of the line elements of the day 2011. A matching component or structure can be used to match the 槔 impedance to the desired system impedance. According to one or more embodiments of the present invention, the multimode antenna structure may be a planar structure incorporated into, for example, a printed circuit board as shown in Fig. 7. In this example, antenna structure 700 includes antenna elements 7, 2, 704 that are connected through ports 706 at ports 708, 710. The antenna structure is fabricated on a printed circuit board substrate 712. The antenna elements shown in this figure are simple quarter-wave monopoles. However, the antenna elements can be any geometry that produces an equivalent effective electrical length. In accordance with one or more embodiments of the present invention, an antenna element having a dual resonant frequency can be used to produce a combined antenna structure having a dual resonant frequency, thereby having a dual operating frequency. Figure 8A shows an exemplary model of a multimode dipole structure 800 in which the dipole antenna elements 802, 804 are divided into two unequal length finger configurations 806, 8 〇 8 and 810, 812, respectively. The dipole antenna elements have a resonant frequency associated with each of the two different finger configuration lengths, thus exhibiting a double resonance. Likewise, the multimode antenna structure using the dual resonant dipole arms exhibits two frequency bands in which high isolation (or small S21) is obtained as shown in Fig. 8B. In accordance with one or more embodiments of the present invention, a multimode antenna structure 900 is shown in FIG. 9 having variable length antenna elements 902, 904 forming a frequency modulated antenna. This can be accomplished by varying the effective electrical length of the antenna elements, such as by a controllable device of each of the antenna elements 902, 904, RF switches 906, 908. In this example, the switch can be opened (by operating the controllable device) to produce a shorter electrical length (for higher frequency operation), 21 201131894 or closed to produce a longer electrical length (for operation) Lower frequency). The operating band of the antenna structure 900, including the high isolation feature, is adjusted for uniformity by adjusting the two antenna elements. The method can be used with a variety of methods for varying the effective electrical length of the antenna elements, including, for example, using a controllable dielectric material, having a device such as a MEMs, a varactor or a tunable dielectric capacitor. One of the antenna elements of the variable capacitor' and the parasitic element is turned on or off. In accordance with one or more embodiments of the present invention, the or the connecting elements provide an electrical connection between the antenna elements, wherein the antenna elements have an electrical length approximately equal to the electrical distance between the elements. In this case, and when the connecting elements are attached to the ends of the antenna elements, the turns are isolated at a frequency near the resonant frequency of the antenna elements. This configuration produces near perfect isolation at a specific frequency. Alternatively, as previously discussed, the electrical length of the connecting element can be increased to expand over the bandwidth over which a particular value is isolated. For example, a straight connection between antenna elements can produce a minimum of s25 dB at a particular frequency. <-10dB bandwidth can be ioomHz. A new response can be obtained by increasing the electrical length, with the minimum S21 being increased to -i5dB, but S21 The bandwidth of <-1 OdB can be increased to 150 MHz. Various other multimode antenna structures in accordance with one or more embodiments of the present invention are possible. For example, the connecting element can have a different geometry, and it can be constructed to include components that alter the structural characteristics of the antenna. These components may include, for example, active inductor and capacitor components, resonator or filter structures or active components such as phase shifters. 22 201131894 In accordance with one or more embodiments of the present invention, the position of the connecting element along the length of the antenna element can be varied to adjust the characteristics of the antenna structure. The frequency bands on which the turns are isolated may be shifted upwards in frequency by moving the attachment points of the connecting elements on the antenna elements away from the turns and near the distal ends of the antenna elements. Figures 10A and 10B illustrate multimode antenna structures 1000, 1002, respectively, each having a connection element electrically connected to the antenna elements. In the antenna structure 1000 of Fig. 10A, the connecting member 1004 is located in the structure such that the gap between the connecting member 1004 and the upper edge 1006 of the ground plane is 3 mm. Figure 10C shows the scattering parameters of the structure, showing that the high isolation is at frequency 1. It was obtained at 15 GHz. A combined capacitor/series inductor matching network is used to provide impedance matching at M5GHz. Figure 10D shows the scattering parameters of structure 1002 of Figure 10B, wherein the gap between connecting element 1008 and the upper edge 1010 of the ground plane is 19 mm. The antenna structure 1002 of Fig. 10B is presented at approximately 1. There is a high isolation operating band at 50 GHz. Figure 11 is a schematic illustration of a multimode antenna structure 1100 in accordance with one or more additional embodiments of the present invention. The antenna structure 1100 includes two or more connection elements 1102, 1104, each of which electrically connects the antenna elements 1106, 1108. (For ease of illustration, only two connecting elements are shown in this figure, it being understood that the use of more than two connecting elements is also contemplated.) The connecting elements 1102, 1104 are along the other such antenna elements 1106, 1108 They are separated. Each of the connecting elements 1102, 1104 includes a switch 1112, 1110. The peak isolation frequency can be selected by controlling the switches 1110, 1112. For example, a frequency fl can be selected by closing the switch 1110 and opening the switch 23 201131894 1112. A different frequency f2 can be selected by closing switch 1112 and opening switch 1110. Figure 12 illustrates a multimode antenna structure 1200 in accordance with one or more alternative embodiments of the present invention. The antenna structure 1200 includes a connection element 1202 to which a filter 12〇4 is operatively coupled. The filter 1204 can be a selected low pass or band pass filter such that the connections of the connecting elements between the antenna elements 1206, 1208 are only effective in a desired frequency band such as a high isolation resonant frequency. . At higher frequencies, the structure will function as two separate antenna elements that are not coupled through the conductive connection elements and open between them. Figure 13 illustrates a multimode antenna structure 1300 in accordance with one or more alternative embodiments of the present invention. The antenna structure 1300 includes two or more connection elements 13A2, 1304 that include filters 1306, 1308, respectively. (For ease of illustration, only two connecting elements are shown in this figure, it being understood that the use of more than two connecting elements is also contemplated.) In one possible embodiment, the antenna structure 1300 is in the connecting element 13 4 (which is adjacent to the antenna 埠) has a low pass filter 1308, and a high pass filter 1306' on the connecting element 13 〇 2 to produce an antenna structure having two high isolation bands, ie, Dual band structure. Figure 14 illustrates a multimode antenna structure 1400 in accordance with one or more alternative embodiments of the present invention. The antenna structure 14A includes one or more connection elements 14A2 having a frequency tunable element 1406 operatively coupled thereto. The antenna structure 1400 also includes antenna elements 14A, 141A. The adjustable frequency component 1406 changes the delay or phase of the electrical connection or changes the reactance of the electrical connection. The amplitude and frequency response of the scattering parameters S21/S12 are affected by changes in electrical delay or impedance, so an antenna structure can be used to adapt or generally optimize isolation for a particular frequency. Figure 15 illustrates a multimode antenna structure 150 in accordance with one or more alternative embodiments of the present invention. The multimode antenna structure 丨5 〇 〇 can be used, for example, in a WIMAX USB server key. The antenna structure 15A can be used, for example, to operate in the WiMAX band from 2300 to 2700 MHz. The sinusoidal antenna structure 1500 includes two antenna elements 1502, 1504 that are connected through a conductive connecting element 丨5〇6. The antenna elements include slots for increasing the electrical length of the components to achieve a desired operating frequency range. In this example, the antenna structure is optimally used for a center frequency of 235 〇 MHz. The length of the slots can be reduced to achieve a higher center frequency. The antenna structure is mounted on a printed circuit board assembly 15〇8. A two component lumped element match is provided at each antenna feed. The antenna structure 1500 can be fabricated, for example, from a metal stamping. It can be 0. Made of 2mm thick copper alloy sheet. The antenna structure 包括5〇〇 includes a pickup feature 510 on the connecting element at the center of the structure, which can be used in an automated pick-and-place assembly process. The antenna structure is also compatible with the reflow combination. Figure 16 illustrates a multimode antenna structure 1600 in accordance with one or more alternative embodiments of the present invention. As with the antenna structure 1500 of Figure 15, the antenna structure 1600 can be used, for example, in a wiMAX USB server key. The antenna structure can be configured to operate, for example, in a WiMAX band from 2300 to 2700 MHz. 25 201131894 The antenna structure 1600 includes two antenna elements 1602, 1604, each of which includes a meandering monopole. The length of the meandering part determines the center frequency. This exemplary design shown in this figure is best used for a center frequency of 2350 MHz. In order to obtain a higher center frequency, the length of the meandering portion can be reduced. A connecting element 1606 electrically connects the antenna elements. A two-component lumped element match is provided at each antenna feed. The antenna structure can be fabricated, for example, of copper as a flexible printed circuit (FPC) mounted on a plastic carrier 1608. The antenna structure can be produced by a metal portion of the fpc. The plastic carrier provides mechanical support and makes installation to a PCB assembly 1610 easier. Alternatively, the antenna structure may be formed of a metal plate. Figure 17 illustrates a multimode antenna structure 1700 in accordance with another embodiment of the present invention. The ahai antenna design can be used, for example, in the usb, Express 34 and Express 54 data card formats. The exemplary antenna structure shown in this figure is designed to operate at frequencies from 2_3 to 6 GHz. The antenna structure can be fabricated, for example, from a metal plate or from an FPC on a plastic carrier 1702. Figure 18A illustrates a multimode antenna structure 1800 in accordance with another embodiment of the present invention. The antenna structure 18A includes a three-mode antenna having three turns. In this configuration, the three monopole antenna elements 18 〇 2, 18 〇 4, 1806 are connected using a connecting element 1808 comprising a conductive loop connecting one of the adjacent antenna elements. The antenna elements are balanced by a common ground grid or a single hollow conductive cylindrical sleeve 1810. The antenna has three coaxial windings 1812, 1814, 1816 for connecting the antenna structure to a communication device. The 26 201131894 coaxial cables 1812, 1814, 1816 pass through the hollow sleeve 1810. The antenna assembly can be constructed from a single flexible printed circuit wrapped in a cylinder and can be packaged in a cylindrical plastic housing to provide a single antenna combination in place of three separate antennas. In an exemplary configuration, the diameter of the cylinder is l〇mm and the total length of the antenna is 56 mm to 2. 45GHz operates with high isolation between turns. The antenna structure can be, for example, a multi-antenna radio system such as a chirp or at 2. 4 to 2. 80. Operating in the 5 GHz band. The 211N system is used together. In addition to the isolation from 埠, each 埠 advantageously produces a different gain pattern as shown in Figure 18B. However, this is only a specific example, it being understood that the structure can be scaled to operate at any desired frequency. It will also be appreciated that the methods previously described in the context of two antennas for adjusting, manipulating bandwidth and generating a multi-band structure can be applied to the multi-turn structure. Although the above embodiment is shown as a true cylinder, it is possible to use other configurations having three antenna elements and connecting elements that produce the same advantages. This includes, but is not limited to, configurations in which there are straight connections such that the connecting elements form a triangle or another polygonal geometry. It is also possible to construct a similar structure by similarly connecting three independent dipole elements instead of three monopole elements with a common ground net. At the same time, although the symmetric configuration of the antenna elements advantageously produces equivalent performance from each turn, for example, the same bandwidth, isolation, impedance matching, it is possible to arrange the antenna elements asymmetrically or at unequal intervals depending on the application. of. Figure 19 illustrates the use of a multimode antenna structure 1900 in a combiner application in accordance with one or more embodiments of the present invention. As shown in Figure 31 201131894, the transmit signal can be applied simultaneously to the two antennas of the antenna structure 1900. In this configuration, the multimode antenna can function as an antenna and power amplifier combiner. The high isolation between the antenna turns limits the interaction between the two amplifiers 1902, 1904, which is known to have undesired effects such as signal distortion or loss of efficiency. The impedance matching at 1906 can be provided at these antennas. 20A and 20B illustrate a multimode antenna structure 2000 in accordance with one or more alternative embodiments of the present invention. The antenna structure 2 can also be used in, for example, a WiMAX USB or ExpressCard/34 device. The antenna structure can be configured to operate, for example, in the WiMAX band from 23 00 to 6000 MHz. The antenna structure 2000 includes two antenna elements 2 〇〇 1, 2004, each antenna element comprising a wide unipolar. A connecting element 2〇〇2 electrically connects the antenna elements. Slots (or other slits) 2005 were used to improve input impedance matching above 5000 MHz. The exemplary design shown in this figure is best used to cover frequencies from 2300 to 6000 MHz. The antenna structure 2000 can be used, for example, in the manufacture of metal stampings. It can be 〇. Made of 2mm thick copper alloy sheet. The antenna structure 2 大 on the connecting member 2002 generally includes a pick-up body 2〇〇3 at the center of mass of the structure, which can be used in an automated pick-and-place assembly process. The antenna structure is also compatible with the signature recombination combination. The feed point 2〇〇6 of the antenna provides a connection point to the RF circuit on a PCB, and also functions as a structure mounting support for the antenna to the PCB. Additional contact points 2〇〇7 provide structural support. Figure 20C illustrates a test set used to measure the performance of the antenna 2 28 201131894 2010. This figure also shows the coordinate reference for the far field field type. The antenna 2〇〇〇 is mounted on a 3〇x88mm PCB 2011 representing an ExpressCard/34 device. The ground portion of PCB 2011 is attached to a larger metal plate 2〇12 (165x254 mm in this example) to represent the ground grid size of a typical notebook computer. The test on PCB 2011 was connected to the antenna via a 5Oohm ribbon transmission line in 2014 and 2016. Figure 20D shows the VS WR measured during these tests 、 2014, 2016. Figure 20E shows the coupling (S2丄 or S12) measured between these test turns. The VSWR and coupling are advantageously low over a wide frequency range (e.g., from 2300 to 6000 MHz). Figure 20F shows the radiation efficiencies measured with reference to these tests 埠 2014 (埠1), 2016 (埠2). Figure 20G shows the correlation calculated between the radiation pattern generated by the excitation test 埠2014(埠1) and those generated by the excitation test 埠2016(埠2). At the frequencies of interest, the radiation efficiency is advantageously high and the correlation between the field types is advantageously low. Figure 20H shows the far field gain field generated by the excitation test 埠2014(埠1) or test埠2016(埠2) at a frequency of 2500MHz. Figures 201 and 20J show the same field measurements at frequencies of 3500 and 5200 MHz, respectively. The field patterns produced by the test 埠 2014 (埠 1) in the φ = 0 or XZ plane and in the θ = 90 or XY plane are different and are complementary to those of the test 埠 2016 (埠 2). 21 and 21 illustrate a multimode antenna structure 2100 in accordance with one or more alternative embodiments of the present invention. The antenna structure 2100 can be used, for example, in a WiMAX USB server record. The antenna structure can be used, for example, to operate in the WiMAX band from 2300 to 2400 MHz. 29 201131894 The antenna structure 2100 includes two antenna elements 2102, 2104, each of which includes a meandering monopole. The length of the meandering portion determines the center frequency. Other curved structures such as, for example, spiral coils and loops can be used to provide a desired electrical length. The exemplary design shown in this figure is best used for a 2350 MHz center frequency. A connecting element 2106 (shown in Figure 21B) electrically connects the antenna elements 21A2, 2104. A two-component lumped element match is provided at each antenna feed. The antenna structure can be made, for example, of copper as a flexible printed circuit (FPC) 21〇3 mounted on a plastic carrier 2101. The antenna structure can be produced by a metal portion of the FPC 2103. The plastic carrier 21〇1 provides mounting pins or pins 2107 for attaching the antenna to a PCB assembly (not shown), and for fixing the FPC 2103 To the pin 2105 on the carrier 2101. The metal portion of 2103 includes an exposed portion or pad 2108 for electrically contacting the antenna with circuitry on the PCB. In order to achieve a higher center frequency, the electrical length of the elements 2102, 2104 can be reduced. Figures 22A and 22B illustrate a multimode antenna structure 22〇0 that is optimally designed for a 2600 MHz center frequency. The electrical lengths of the elements 22A2, 2204 are shorter than the electrical lengths of the elements 2102, 2104 of Figures 21A and 21B because the metallization at the ends of the elements 2202, 2204 has been removed and the width of the component feed end is increase. Figure 23A illustrates a test combination 23 00 using antenna 2100 of Figures 21A and 21B along with a far field field coordinate reference. Figure 2 3 B shows the VSWR measured at test 埠 2302 (埠1), 2304 (埠2). Figure 23C shows the coupling measured between the tests 埠 2302 (埠1), 2304 (埠2) (S21 30 201131894 or S12). The VSWR and coupling are advantageously low at frequencies of interest (e.g., from 2300 to 2400 MHz). Figure 23D shows the radiation efficiency measured with reference to the test cartridges. Figure 23E shows the correlation between the radiation pattern generated by the excitation test 埠 2302 (埠 1) and those generated by the excitation test 埠 2 3 04 (埠 2). At the frequencies of interest, the radiation efficiency is advantageously high and the correlation between the field types is advantageously low. Figure 23F shows the far field gain pattern generated by the excitation test 2302 (埠I) or test 槔 2304 (埠2) at a frequency of 2400 MHz. The field patterns produced by the test 埠 2302 (埠 1) in the φ = 〇 or XZ plane and in the θ = 90 or ΧΥ plane are different and are complementary to those of the test 埠 2304 (埠 2). Figure 23G shows the VSWR measured at the test 埠 of antenna 2200 in place of combination 2300 of antenna 2100. Figure 23 shows the coupling measured between these test ( (S21 or S12). The VSWR and coupling are advantageously low at the frequency of interest (eg, from 2500 to 2700 ΜΗΖ). Figure 231 shows the radiation efficiency measured with reference to the test 《. Figure 23 J shows the transmitted excitation test 埠 2302 (埠1 The correlation between the generated radiation pattern and those generated by the excitation test 埠 2304 (埠 2). At the frequencies of interest, the radiation efficiency is advantageously high and the correlation between the field types is advantageously low. Figure 23 shows the far-field gain pattern generated by exciting test 埠 2302 (槔1) or test 槔 2304 (埠2) at a frequency of 2600MHz. The field patterns produced by the test 埠23〇2(埠丨) in the φ=〇 or ΧΖ plane and in the θ=90 or ΧΥ plane are different and complement those of the test 埠 2304 (埠2). One or more additional embodiments of the present invention are directed to beam pattern control 31 201131894 techniques for null steering and beam pointing purposes. When the technique is applied to a conventional array antenna (including separate antenna elements separated by small portions of a wavelength), each element of the array antenna is fed with a signal that is a reference signal or The phase shift of the wavelength is deformed. For a non-uniform linear array with equal excitation, the resulting beam pattern can be described by an array factor F, which depends on the phase of each individual element and the element spacing d between the elements. yV-l ^ = XexP[7^(>0rf cos^ + a)] where 'β=2π/λ, N=total number of components#, α=phase shift between successive elements, and θ=starts from the array axis The angle can be adjusted to a different direction by controlling the phase a equal to the maximum value of the value aj 'F, thereby controlling the direction in which a maximum signal is broadcast or received. The inter-element spacing in conventional array antennas is typically on the order of 1/4 wavelength' and the antennas can be tightly coupled with nearly the same polarization direction. This advantageously reduces coupling between components because coupling can cause problems in several array antenna designs and performance. For example, problems such as field distortion and scanning dead zones (see Stutzman, Antenna Theory and Design, Wiley 1998's 122-128 and 135-136 and pages 466-472) may be due to excessive inter-element coupling and a given number of components. A reduction in the maximum gain that can be achieved. Beam field control techniques can advantageously be applied to all of the multimode antenna structures described herein. The antenna structures have connections that are connected through one or more connection elements and exhibit high isolation between multiple feed points. Antenna element 32 201131894 The phase between the bees in the structure of the antenna can be used to control the antenna pattern. It has been found that since the coupling between the feed points is reduced, a higher peak gain is achievable in a given direction when the antenna is used as a simple beamforming array. Thus, a greater gain can be achieved in a selected direction by a high isolation antenna structure that utilizes phase control of the carrier signal represented at its feed end in accordance with various embodiments of the present invention. In cell phone applications where the antennas are separated by an interval of less than 1/4 wavelength, the mutual coupling effect in conventional antennas reduces the radiation efficiency of the array, thus reducing the maximum achievable gain. The direction of the maximum gain produced by the antenna pattern can be controlled by controlling the phase of the carrier signal supplied to each feed point of a high isolation antenna in accordance with various embodiments. An advantage of, for example, 3 dB gain obtained by beam steering is advantageous, particularly in portable device applications where the beam pattern is fixed and the device orientation is randomly controlled by the user. As shown, for example, in a schematic block diagram of Fig. 24 illustrating a field type control device 24A in accordance with various embodiments, a relative phase shift α through phase shifter 24〇2 is applied to each antenna feed applied thereto. The RF signals of the 2404, 2408. The signals are fed to the antenna 天线 of the antenna structure 2410, respectively. Phase shifter 2402 can include standard phase shifting components such as, for example, electronically controlled phase shifting devices or standard phase shifting networks. 25A-25G for different phase differences a between the two feeds of the antenna, providing an antenna pattern produced by a closely spaced 2_D conventional array of dipole antennas and high by various embodiments in accordance with the present invention Isolation Antenna 33 201131894 A comparison of antenna field patterns produced by a 2-D array. In the 25A-25G diagram, the curve shows the antenna pattern of θ = 90 degrees. In the figure the solid line represents the antenna pattern produced by the isolated feed single element antenna according to various embodiments, and the dashed line represents the antenna pattern produced by two independent monopole conventional antennas, wherein the conventional antenna is Equal to the separation of the single element by a distance of the width of the isolated feed structure. Therefore, the conventional antenna and the high-isolation antenna are generally of equal size. In all of the cases shown in the figures, when compared to the two independent conventional dipoles, 'the peak gain produced by the high isolation antennas according to various embodiments yields a larger gain margin, while Provide azimuth control of the beam field. This behavior makes it possible to use a high-isolation antenna in the transmission or reception in which a particular direction is required or an additional gain. This direction can be made by adjusting the relative phase between the signal of the decision point. This is particularly advantageous for applications such as, for example, the pure point of the base station. The combined high isolation antenna provides greater advantages when compared to two similar antenna elements. In the 25A® towel (10)*, which is based on various realities, each of which is shown in Figure 25, which is shown in Figure 25, is a combination of the various embodiments according to various embodiments. The azimuthal field pattern of the shift (α = 6 〇 (the θ = 90 graph with a 60-degree phase difference between the feed points) shows a larger peak gain (in the dip). As shown in Fig. 25D, the combined dipole according to various embodiments uses a shifted azimuthal field pattern (α = 9 〇 (9 相位 phase difference between feed points) Θ-90 Figure) shows an even larger peak gain (at _〇). As shown in Fig. 25, the combined dipole according to various embodiments uses a shifted azimuth field type (α = 12 〇 (120 degree phase difference between feed points). The lobes (θ = 9 〇 in φ = 18 〇) show a larger peak gain (at Φ^ο). As shown in Figure 25F, the combined dipole according to various embodiments uses an even larger lobes (at φ = 180) with a = 150 (150 degree phase difference between feed points). A shifted azimuthal field pattern (θ = 9 〇 map) shows a larger peak gain (in φ = 〇). As shown in Figure 25G, the combined dipole according to various embodiments uses an a = 18 0 (18 0 degree phase difference between feed points) double lobe azimuth field (θ = 90 Figure) shows a larger peak gain (in φ = 〇 & 18 〇). Figure 26 illustrates the ideal gain advantage of the combined high isolation antenna of two independent dipoles in accordance with one or more embodiments as a function of the phase angle difference between the feed points of a two feed point antenna array. In accordance with one or more embodiments of the present invention, the increased gain obtained by field mode control using an antenna structure that is connected to two parallel dipoles through a meandering connection element can be used to improve the range or reliability of a wireless link. Alternatively, the increased gain may allow a portable or other device to obtain equivalent wireless link performance with reduced transmission power. For example, by the field type 35 201131894 ^Dragon (10) (10) - the average transmission increase will allow the transmission power to be the same as the chain performance. The reduction in transmission power is advantageous in several aspects. First, the τ-receiving wireless device usually needs to meet the second absorption rate (SAR) regulatory limit, but does not make some compromises in performance. The reduction in the power of the transmission wheel allows the SAR peak to correspond to the factory> and the performance is compromised. In addition to this, the reduction in transmission power reduces the burden on the output ’, allowing for designs with lower power and higher linearity. In addition, the reduction in transmission power is beneficial for extending the life of the f-pool of portable or other devices and reducing their heat dissipation requirements. While phase control is used to produce the desired far field gain increase, the phase (four) change can also change the near field and affect the SAR value. In order to achieve a SAR (four) material less, the increase in antenna far-field gain should be greater than the increase in any S-value. Through experiments, the towel has found that the actual (10) value is relatively small with respect to the phase gain compared to the far field gain. A New USB ship with a two-bee antenna structure for field-type control applications in the 19 〇OMHz band is shown in the wipes. As shown in Fig. 31, the sar value determined by simulating the structure of the third diagram is relatively independent of the phase-inducing _(four) phase, so that the benefit of the measured SAR peak reduction is The relative phase value is achieved while providing full azimuth control of the beam pattern. The techniques described herein for reducing the near field light level and the sAR value are preferably compared to the above described high impedance multimode antenna structure having a 7L piece of electrically connected antenna elements. _ use. However, such techniques are also useful for use with antenna arrays that include a plurality of radiating elements that are steerable to provide antenna field type 36 201131894 control and that increase the gain in a selected direction. Further embodiments of the present invention provide an increased high isolation multimode antenna structure between multi-band antennas 接近 that are operating close to each other over a given frequency range. In these embodiments, a band stop is incorporated into one of the antenna elements of the antenna structure to provide reduced coupling at the frequency to which the slot is adjusted. Figure 27A schematically illustrates a simple dual band spur monopole antenna 2700. The antenna 2700 includes a strip stop 2702 that defines two branch resonators 2704 ' 2706. The antenna is driven by a signal generator 2708. Depending on the frequency at which antenna 2700 is being driven, various current distributions are implemented on the two dividers 2704, 2706. As shown in Fig. 27A, the physical size of the groove 2702 is defined: the width of the household is Ws' and the length is Ls. When the excitation frequency satisfies the condition Ls = l 〇 / 4, the groove characteristics become resonance. At this point the current distribution concentrates around the shorted portion of the slot, as shown in Figure 27b. The current flowing through the branch resonators 2704 ' 2706 is approximately equal, and the sides of the eight slots 2702 point in opposite directions. This causes the antenna structure 27 to behave in a manner similar to the tributary band stop filter 2720 (shown schematically in Figure 27C) which converts the antenna input impedance down to significantly below the nominal source impedance. This large impedance mismatch results in a very high VSWR as shown in Figures 27D and 27E, resulting in the desired frequency rejection. The band stop technique can be applied to an antenna system having two (or more) antenna elements operating close to each other, wherein one antenna element needs to pass a signal of 37 201131894 with a desired frequency and the other does not. In one or more embodiments, one of the two antenna elements includes a strip stop and the other does not. Figure 28 is an illustration of an antenna structure 28A including a first antenna element 2802, a second antenna element 28A4, and a connecting element 2806. The antenna structure 2800 includes 埠 2808 and 2810 at antenna elements 2802 and 2804, respectively. In this example, a signal generator drives the antenna element 2802 at 埠28〇8, while a meter is coupled to the 埠281埠 to measure the current of 埠2810. However, it should be understood that either or both of the two turns can be driven by the signal generator. The antenna element 28〇2 includes a strip stop 2812 defining two branch resonators 2814, 2816. In this embodiment, the branch resonators comprise the main transmitting portion of the antenna structure and the antenna element 2804 comprises a diversity receiving portion of the antenna structure. Due to the large mismatch at the antenna element 2802埠 with the resistive slot 2812, the mutual coupling of the antenna element 2802 with the gate of the diversity receiving antenna element 28〇4 (actually matching at the resonant frequency of the slot) will Very small, so it will produce a fairly high degree of isolation. Figure 29 is a perspective view of a multimode antenna structure 2900 including a multi-band sub-receiver antenna system using the band-stop technique in the G P S band in accordance with one or more additional embodiments of the present invention. It is 1575. 42 MHz with a bandwidth of 20 MHz. The antenna structure 2900 is in a dielectric substrate 2902 in which the substrate is formed as a layer on the electrolyte carrier 2904. The antenna structure 2900 includes a GPS tape and slot 38 201131894 2906 on the primary transmit antenna element 2908 of the antenna structure 2900. The antenna structure 2900 also includes a diversity receive antenna element 2910, and a connection element 2912 that connects the diversity receive antenna element 2910 and the primary transmit antenna element 2908. A GPS receiver (not shown) is coupled to the diversity receive antenna element 2910. In order to generally minimize antenna coupling from the primary transmit antenna element 2908 at these frequencies, and generally maximize the diversity antenna radiation efficiency, the primary antenna element 2908 includes a band stop slot 2906 and is near the center of the GPS band. It is adjusted to one quarter of the electrical wavelength. The diversity receive antenna element 2910 does not include such a stop slot, but includes a GPS antenna element that is properly matched to the primary antenna source impedance so that there will typically be maximum power conversion between the GPS receiver and the GPS receiver. Although the two antenna elements 2908, 2910 coexist in close proximity, since the high VSWR of the slot 2906 at the primary transmit antenna element 2908 reduces the coupling to the source impedance of the primary antenna element when the slot 2906 is adjusted to frequency, Isolation is provided between the two antenna elements 2908, 2910 at the GPS frequency. The mismatch between the two antenna elements 2908, 2910 in the GPS band is large enough to decouple the antenna elements to meet the isolation requirements of the system design, as shown in Figures 29B and 29C. The antenna structure, antenna element and connecting element described herein in accordance with various embodiments of the present invention preferably form a single integrated radiating structure such that a signal fed to either of the turns excites the entire antenna structure to radiate as a unitary body. Instead of being a separate radiating structure. Thus, the techniques described herein provide isolation of the antenna , without using a decoupling network at the antenna feed point. It will be understood that, although the invention has been described above in terms of the specific embodiments of the invention, the above-described embodiments are only provided by way of illustration and not limitation. Various other embodiments, including but not limited to the following, are also within the scope of the appended claims. For example, the elements or components of the various multi-mode antenna structures described herein may be further divided into additional components or combined to form fewer components for performing the same function. While the preferred embodiment of the invention has been described, I: Schematic description of the figure 3 Figure 1A illustrates an antenna structure with two parallel dipoles. Figure 1B illustrates the current generated by a dipole excitation in the antenna structure of Figure 1A. Figure 1C illustrates a model corresponding to the antenna structure of Figure 1A. Figure 1D is a diagram illustrating the scattering parameters of the antenna structure of Figure 1C. Figure 1E is a diagram illustrating the current ratio of the antenna structure of Figure 1C. Figure 1F is an illustration of the gain field pattern of the antenna structure of Figure 1C. Figure 1G is a diagram illustrating the envelope correlation of the antenna structure of Figure 1C. Figure 2A illustrates an antenna structure connected to two parallel dipoles through a connecting element in accordance with one or more embodiments of the present invention. Figure 2B illustrates a model corresponding to the antenna structure of Figure 2A. Fig. 2C is a diagram illustrating the scattering parameters of the antenna structure of Fig. 2B. Figure 2D is a diagram illustrating the scattering parameters of the antenna structure of Figure 2B. 40 201131894, where there is lumped element impedance matching at the two turns of the antenna structure. Figure 2E is a diagram illustrating the current ratio of the antenna structure of Figure 2B. Fig. 2F is a diagram illustrating the gain field pattern of the antenna structure of Fig. 2B. Figure 2G is a diagram illustrating the envelope correlation of the antenna structure of Figure 2B. Figure 3A illustrates an antenna structure connected to two parallel dipoles through a meandering connecting element in accordance with one or more embodiments of the present invention. Fig. 3B is a diagram showing the scattering parameters of the antenna structure of Fig. 3A. Figure 3C is a diagram illustrating the current ratio of the antenna structure of Figure 3A. Figure 3D is an illustration of the gain field pattern of the antenna structure of Figure 3A. Figure 3E is a diagram illustrating the envelope correlation of the antenna structure of Figure 3A. Figure 4 illustrates an antenna structure of a ground or ground network in accordance with one or more embodiments of the present invention. Figure 5 illustrates a balanced antenna structure in accordance with one or more embodiments of the present invention. Figure 6A illustrates an antenna structure in accordance with one or more embodiments of the present invention. Figure 6B is a diagram showing the scattering parameters of a particular dipole width antenna structure of Figure 6A. Figure 6C is a diagram showing the scattering parameters of the antenna structure of another dipole width in Fig. 6A. 41 201131894 Figure 7 illustrates an antenna structure fabricated on a printed circuit board in accordance with one or more embodiments of the present invention. Figure 8A illustrates an antenna structure having apex resonance in accordance with one or more embodiments of the present invention. Fig. 8B is a diagram illustrating the scattering parameters of the antenna structure of Fig. 8A. Figure 9 illustrates a tunable antenna structure in accordance with one or more embodiments of the present invention. 10A and 10B illustrate an antenna structure having connecting elements pointing in different positions along the length of the antenna element in accordance with one or more embodiments of the present invention. Figures 10C and 10D are diagrams illustrating the scattering parameters of the antenna structures of Figures 10A and 10B, respectively. Figure 11 illustrates an antenna structure including a connecting element having a switch in accordance with one or more embodiments of the present invention. Figure 12 illustrates an antenna structure having a connection element to which a filter is coupled, in accordance with one or more embodiments of the present invention. Figure 13 illustrates an antenna structure having two connecting elements, some of which are coupled to the connecting elements, in accordance with one or more embodiments of the present invention. Figure 14 illustrates an antenna structure having an adjustable frequency connection element in accordance with one or more embodiments of the present invention. Figure 15 illustrates an antenna structure mounted on a PCB assembly in accordance with one or more embodiments of the present invention. 42 201131894 Figure 16 illustrates another antenna structure mounted on a PCB assembly in accordance with one or more embodiments of the present invention. Figure 17 illustrates an alternate antenna structure that can be mounted on a PCB assembly in accordance with one or more embodiments of the present invention. Figure 18A illustrates a three-phase antenna structure in accordance with one or more embodiments of the present invention. Figure 18B is a diagram illustrating the gain pattern of the antenna structure of Figure 18A. Figure 19 illustrates an antenna and power amplifier combiner application of an antenna structure in accordance with one or more embodiments of the present invention. 20A and 20B illustrate a multimode antenna structure that can be used, for example, in a WiMAX USB or ExpressCard/34 device, in accordance with one or more additional embodiments of the present invention. Figure 20C illustrates a test combination used to measure the performance of the antennas of Figures 20A and 20B. Figures 20D through 20J illustrate test measurements of the antennas of Figures 20A and 20B. 21A and 21B illustrate a multimode antenna structure that can be used, for example, in a WiMAX USB server dongle, in accordance with one or more alternative embodiments of the present invention. 22A and 22B illustrate a multimode antenna structure that can be used, for example, in a WiMAX USB server key, in accordance with one or more alternative embodiments of the present invention. Figure 23A illustrates a test combination that can be used to measure the antenna properties of Figures 21A and 21B. Figures 23B to 23K illustrate the test measurement results of the antennas of Figs. 21A and 21B. Figure 24 is a schematic block diagram of an antenna structure having a beam steering mechanism in accordance with one or more embodiments of the present invention. Figures 25A through 25G illustrate the test measurements of the antenna of Figure 25A. Figure 26 illustrates the gain advantage of an antenna structure as a function of the phase angle difference between the feed points, in accordance with one or more embodiments of the present invention. Figure 27A is a schematic diagram showing the structure of a simple dual band spur monopole antenna. Figure 27B illustrates the current distribution in the antenna structure of Figure 27A. Figure 27C is a schematic diagram illustrating a spurline band stop filter. Figures 27D and 27E are test results illustrating frequency suppression in the antenna structure of Figure 27A. Figure 28 is a schematic diagram showing the structure of a band-stopped antenna in accordance with one or more embodiments of the present invention. Figure 29A illustrates an alternate antenna structure having a resisting slot in accordance with one or more embodiments of the present invention. Figures 29B and 29C illustrate test measurement results for the antenna structure of Figure 29A. Figure 30 illustrates an exemplary U S B server key with a two-turn antenna configuration for field control applications in the 1900 MHz band.

Figure 31 illustrates the SAR 44 201131894 value determined by simulating the device of Figure 30. [Description of main component symbols] 100.. . Antenna structure 102.. Dipole 104.. Dipole 106.. .埠108...埠200.. . Two-antenna antenna structure 202.. Components 204.. Components 206".埠208".埠210.. . Connection element 212.. Connection element 300.. . Multimode antenna structure 302.. Antenna element 304.. Antenna element 310.. . Connection element 312.. Connecting element 400.. Antenna structure 402.. Antenna element 404.. Antenna element 406.. Antenna element 412···埠418...埠500.. Antenna structure 502.. Antenna element 504.. Antenna Element 506.. Antenna Element 508.. Antenna Element 510...埠512···埠600.. . Multimode Antenna Structure 602.. Dipole 604.. Dipole 606.. Connection Element 608. . Connecting element 700.. Antenna structure 702.. Antenna element 704.. Antenna element 706.. Connecting element 708.. .埠710.••埠712... Printed circuit board substrate 800.. Mode dipole structure 802.. dipole antenna element 45 201131894 804.. dipole antenna element 806.. finger structure 808.. finger structure 810.. finger structure 812.. finger structure 902.. . Antenna Element 904.. . Antenna Element 906.. .R F switch 908.. RF switch 1000... Multimode antenna structure 1002.. Multimode antenna structure 1004.. Connection element 1006.. Ground plane upper edge 1008.. Connection element 1010.. . 1100.. . Multimode antenna structure 1102.. Connecting element 1104.. Connecting element 1106.. Antenna element 1108.. Antenna element 1110.. Switch 1112... Switch 1200.. Multimode antenna structure 1202. . Connecting element 1204.. Filter 1206.. Antenna element 1208.. Antenna element 1300.. Multimode antenna structure 1302.. Connecting element 1304.. Connecting element 1306.. Filter 1308. . Filter 1400.. Multimode Antenna Structure 1402.. Connecting Element 1406.. Adjustable Frequency Element 1408.. Antenna Element 1410.. Antenna Element 1500.. Multimode Antenna Structure 1502.. Antenna Element 1504.. Antenna Element 1506.. Conductive Connection Element 1508.. Printed Circuit Board Assembly 1510.. Pickup Body 1600.. Multimode Antenna Structure 1602.. Antenna Element 1604.. Antenna Element 1606. . Connecting element 1608.. plastic carrier 46 201131894 1610.. .PCB combination 1700.. .Multimode antenna structure 1702...Plastic carrier 1800.. .Multimode Line structure 1802... three monopole antenna elements 1804.. three monopole antenna elements 1806.. three monopole antenna elements 1808.. connecting elements 1810.. sleeve 1812.. coaxial cable 1814.. coaxial Cable 1816.. . Coaxial cable 1900.. Multimode antenna structure 1902.. Amplifier 1904.. Amplifier 2000.. Multimode antenna structure 2001.. Antenna component 2002.. . Connecting component 2003.. Pickup Shape 2004.. . Antenna Element 2005.. . Slot 2006.. Feed Point 2007.. . Contact Point 2010.. Test Combination

2011.. .PCB 2012.. .Metal Plate 2014...Test 埠2016.. .Test 埠2100···Multimode Antenna Structure 2101..Plastic Carrier 2102.. .Antenna Element 2103.. .Flexible Printed Circuit (FPC 2104.. . Antenna element 2105.. . Pin 2106.. . Connecting element 2107.. . Pin 2108.. Pad 2200... Multimode antenna structure 2202.. Antenna element 2204.. Antenna element 23 00...test combination 2302...test 埠2304..test 璋2400.. field type control device 2402.. phase shifter 2404.. antenna feed 2408.. antenna feed 2410.. antenna Structure 47 201131894 2700.. . monopole antenna 2702.. with stop groove 2704.. branch resonator 2706.. branch resonator 2708.. signal generator 2720.. branch line band rejection filter 2800.. Antenna structure 2802.. 1st antenna element 2804.. 2nd antenna element 2806.. . Connecting element 2808 •"埠2810.. .埠2812.. .Restriction groove 2814.. branch resonance 2816.. branch resonator 2900.. multimode antenna structure 2902.. elastic film dielectric substrate 2904.. electrolyte carrier 2906.. with stop groove 2908.. antenna element 2910.. antenna element 2912.. . Connecting element 51 1.. . Scattering parameters 512.. . Scattering parameters 521.. . Scattering parameters 48

Claims (1)

201131894 VII. Patent application scope: 1. A method for reducing near-field radiation and specific absorption rate (SAR) values in a communication device, the communication device comprising a multi-mode antenna structure for transmitting and receiving electromagnetic signals, and for processing a circuit for transmitting signals to the antenna structure and signals from the antenna structure, the antenna structure comprising: a plurality of antennas operatively coupled to the circuit; a plurality of antenna elements, each of the antenna elements being operatively Coupling to a different one of the antennas; and one or more connecting elements, respectively electrically connecting the corresponding antenna elements at a position on each of the antenna elements, each antenna element being coupled to the day The turns are spaced apart to form a single radiating structure, and such that current on one antenna element flows to a connected adjacent antenna element, and typically does not flow to the antenna port coupled to the adjacent antenna element, And the currents flowing through the one antenna element and the adjacent antenna elements are generally equal in magnitude, thus a given desired signal frequency An antenna pattern excited by an antenna 一般 is generally electrically isolated from a pattern excited by another antenna ,, and the antenna structure produces a diversity antenna pattern; the method comprises the steps of: adjusting the feed to the antenna The relative phase between the signals of adjacent antennas of the structure such that a signal fed to the one antenna has a different phase than a signal fed to the adjacent antenna to provide an antenna Field mode control and increasing gain in a selected direction toward a receiving point; and using a transmission power lower than the transmission power used in a non-field type control operation of the antenna structure, such that the communication device uses a low 49 In 201131894, the transmission power of the non-field control operation is used to obtain the wireless link performance substantially equal to the receiving point, thereby reducing the specific absorption rate. 2. The method of claim 2, wherein adjusting the relative phase between the signals comprises using an electronically controlled phase shifting device to adjust the relative phase between the signals. 3. The method of claim i, wherein adjusting the relative phase of the signal comprises using a __ phase shifting network to adjust the relative phase between the signals. 4. If you apply for a patent scope! The method of claim wherein adjusting the relative phase between the signals comprises adjusting the relative phase between the signals by controlling a phase of a carrier signal provided by each of the plurality of antenna bars. 5. The method of claim i, wherein the communication device is a cellular handset, a personal digital assistant (pDA), a wireless network device, or a personal computer (PC) data card. 6. If you apply for a patent scope! The method of claim wherein the antenna elements comprise a helical coil, a broadband planar profile, a wafer antenna, a meander profile, a loop or an inductive shunt. 7. The method of claim i wherein the multimode antenna structure comprises a planar structure fabricated on a printed circuit board substrate. 8. The method of claim i, wherein the multimode antenna structure comprises a metal stamping member, and the metal stamping member comprises a picking body at a centroid of the metal stamping member for an automated inspection _ Used in the assembly process. The method of claim 1, wherein the multimode antenna structure comprises an elastic printed circuit mounted on a plastic carrier or a plastic housing of a device. 10. The method of claim 1, wherein the receiving point is a base station, an mobile terminal, or a router. 11. The method of claim 1, wherein adjusting the relative phase between signals comprises dynamically adjusting the relative phase between signals fed to adjacent antennas to maintain a destination that reaches the receiving point. Generally optimized communication links. 12. A method of reducing near field radiation and specific absorption rate (SAR) values in a communication device, the communication device comprising an antenna array for transmitting and receiving electromagnetic signals, and for processing transmission to the antenna array a circuit and a signal from the antenna array, the antenna array including a plurality of radiating elements, each radiating element having an antenna 可 operatively coupled to the circuit; the method comprising the steps of: adjusting is fed to the The relative phase between the signals of the antennas of the antenna array such that a signal fed to one antenna has a different phase than a signal fed to the other antenna to provide an antenna Field mode control and increasing gain in a selected direction toward a receiving point; and using a transmission power lower than the transmission power used in a non-field type control operation of the antenna array such that the communication device is used below The non-field type controls the transmission power of the operation to obtain a wireless link performance substantially equivalent to the reception point, thereby reducing the specific absorption rate. 51 201131894 4 The method of claim n, wherein adjusting the relative phase of the signals comprises using an electronically controlled phase shifting device to adjust the relative phase between the signals. The method of claim 12, wherein adjusting the relative phase between the signals comprises using a phase shifting network to adjust the relative phase between the signals. 15. The method of claim 2, wherein the relative phase between the adjustment signals comprises the phase of the carrier money provided by each antenna material controlled by the plurality of antenna bees. Adjust the relative phase between the signals. 16. The method of claim 12, wherein the communication device is a cellular phone, a PDA, a wireless network device or a pC data card. 17. The method of claim 12, wherein the radiating elements comprise a helical coil, a broadband planar shape, a wafer antenna, a meander profile, a loop or an inductive shunt. 8. The method of claim 12, wherein the antenna array comprises a planar structure fabricated on a printed circuit board substrate. 19. The method of claim 12, wherein the antenna array comprises a metal stamping member, and the metal stamping member comprises a picking body at a centroid of the metal stamping member, in an automated 捡-type Used in the assembly process. 20. The method of claim 12, wherein the antenna array comprises an elastic printed circuit mounted on a plastic carrier or a plastic housing of a device. The method of claim 12, wherein the receiving point is a base station, an mobile terminal, or a router. 22. The method of claim 12, wherein adjusting the relative phase between the signals comprises dynamically adjusting the relative phase between signals fed to the antennas to maintain one of the receiving points. Generally optimized communication links. 53