JP5616955B2 - Multimode antenna structure - Google Patents

Description

Citation to Related Application This application claims priority from US Provisional Application No. 61 / 161,669, filed March 19, 2009 and entitled “Multi-Mode Antenna Structure”. The application is incorporated herein by reference. This application is a continuation-in-part of U.S. Patent Application No. 12 / 099,320 (issued as U.S. Patent No. 7,688,273) filed on April 8, 2008 entitled "Multi-Mode Antenna Structure". US patent application Ser. No. 12 / 099,320 is filed from US patent application Ser. No. 11 / 769,565 (issued as US Pat.No. 7,688,275) filed June 27, 2007 entitled “Multi-Mode Antenna Structure”. US Patent Application No. 11 / 769,565, which is a continuation-in-part of U.S. Patent, is a provisional application of U.S. Provisional Application No. 60 / 925,394 filed April 20, 2007 entitled "Multi-Mode Antenna Structure" and This is also based on US Provisional Patent Application No. 60 / 916,655 filed on March 8, 2007 entitled “Multimode Antenna Structure”, which is incorporated by reference in its entirety.

  The present invention relates generally to wireless communication devices, and more particularly to antennas used in such devices.

  Many communication devices are equipped with multiple antennas that are mounted in close proximity (ie, separated by less than a quarter of a wavelength) and that can operate simultaneously within the same frequency band. Common examples of such communication devices include portable communication products such as mobile handsets, personal digital assistants (PDAs), and wireless network devices or data cards for personal computers (PCs). Many system architectures (e.g. mimos (MIMO)) and standard protocols for mobile wireless communication devices (e.g. 802.11n for wireless LAN, 3G data communication such as 802.11e (WiMAX), HSDPA, and 1xEVDO) Requires many antennas to work.

  According to one or more embodiments, a multi-mode antenna structure for transmitting and receiving electromagnetic signals in a communication device is provided. The communication device includes a circuit for processing a signal transmitted and received by the antenna structure. The antenna structure includes a plurality of antenna ports coupled to the circuit, a plurality of antenna elements each operably coupled to a different one of the antenna ports, and a plurality of connection elements. Each of the connection elements electrically connects adjacent antenna elements, and the antenna elements and the connection elements are disposed along the periphery of the antenna structure and form a single radiation structure. The current on one antenna element flows to the connected adjacent antenna element and largely bypasses the antenna port coupled to the adjacent antenna element, and the antenna mode excited by one antenna port is given by In the desired signal frequency range, it is generally electrically isolated from the modes excited by other antenna ports, and the antenna structure produces various antenna patterns.

  In accordance with one or more embodiments of the present invention, a multimode antenna structure for transmitting and receiving electromagnetic signals in a communication device is provided. The communication device includes a circuit for processing a signal transmitted and received by the antenna structure. The antenna structure includes a plurality of antenna ports coupled to the circuit and a plurality of antenna elements each operably coupled to a different one of the antenna ports. The plurality of antenna elements are arranged along the periphery of the antenna structure. The antenna structure also includes a connection element that electrically connects the antenna elements to a common point to form a single radiating structure. The current on one antenna element flows to another antenna element and generally bypasses the antenna port coupled to the other antenna element, and the antenna mode excited by one antenna port is a given desired signal frequency. In range, it is generally electrically isolated from modes excited by another antenna port, and the antenna structure produces a variety of antenna patterns.

  In accordance with one or more embodiments of the present invention, a multimode antenna structure for transmitting and receiving electromagnetic signals in a communication device is provided. The communication device includes a circuit for processing a signal transmitted and received by the antenna structure. The antenna structure includes a plurality of antenna ports coupled to the circuit and a plurality of antenna elements each operably coupled to a different one of the antenna ports. Each antenna element includes upper and lower planar portions that are generally parallel and spaced apart, and side portions that connect the upper and lower planar portions. The antenna structure also includes one or more connecting elements, each connecting element electrically connecting an adjacent antenna element in one of the planar portions such that the antenna elements form a single radiating structure. Connect. Current on one antenna element flows to one connected adjacent antenna element and generally bypasses the antenna port coupled to the adjacent antenna element. The currents flowing through the one antenna element and the adjacent antenna elements are approximately equal in magnitude, and an antenna mode excited by one antenna port is excited by another antenna port in a given desired signal frequency range It is generally electrically separated from the mode and the antenna structure produces a variety of antenna patterns.

  Various embodiments of the invention are described in the following detailed description. As will become apparent below, the invention is capable of other and different embodiments, and its several details can be modified in various ways without departing therefrom. Accordingly, the drawings and description are to be regarded as illustrative in nature and not in a limiting or restrictive sense, and it is to be understood that the scope of the present application is indicated in the claims.

An antenna structure with two parallel dipoles is shown. FIG. 3 shows a current flow generated from excitation of one dipole in the antenna structure of FIG. 2 shows a model corresponding to the antenna structure shown in FIG. 2 is a graph showing scattering parameters related to the antenna structure of FIG. 1 (C). 2 is a graph showing a current ratio related to the antenna structure of FIG. 3 is a graph showing a gain pattern related to the antenna structure of FIG. 1 (C). 2 is a graph showing an envelope correlation for the antenna structure of FIG. Fig. 3 shows an antenna structure comprising two parallel dipoles connected by a connecting element according to one or more embodiments of the invention. A model corresponding to the antenna structure of FIG. 3 is a graph showing scattering parameters related to the antenna structure of FIG. 2 (B). 3 is a graph showing scattering parameters for the antenna structure of FIG. 2 (B) showing lumped element impedance matching at both ports. 3 is a graph showing a current ratio regarding the antenna structure of FIG. 2 (B). 3 is a graph showing a gain pattern related to the antenna structure of FIG. 2 (B). 3 is a graph showing an envelope correlation for the antenna structure of FIG. 2 (B). Fig. 4 shows an antenna structure with two parallel dipoles connected by meander connection elements according to one or more embodiments of the invention. 4 is a graph showing scattering parameters related to the antenna structure of FIG. 4 is a graph showing a current ratio regarding the antenna structure of FIG. FIG. 4 is a graph showing a gain pattern related to the antenna structure of FIG. FIG. 4 is a graph showing an envelope correlation regarding the antenna structure of FIG. 1 illustrates an antenna structure with ground or counterpoise according to one or more embodiments of the present invention. 1 illustrates a balanced antenna structure according to one or more embodiments of the present invention. 1 illustrates an antenna structure according to one or more embodiments of the present invention. FIG. 7 is a graph showing scattering parameters for the antenna structure of FIG. 6 (A) for a specific dipole width dimension. FIG. 7 is a graph showing scattering parameters for the antenna structure of FIG. 6 (A) for another dipole width dimension. 2 illustrates an antenna structure fabricated on a printed circuit board according to one or more embodiments of the present invention. 1 illustrates an antenna structure with double resonance according to one or more embodiments of the present invention. FIG. 9 is a graph showing scattering parameters related to the antenna structure of FIG. 8 (A). FIG. 1 illustrates a tunable antenna structure according to one or more embodiments of the present invention. FIG. 6 shows an antenna structure with connecting elements arranged at different positions along the length of the antenna element according to one or more embodiments of the invention. FIG. 6 shows an antenna structure with connecting elements arranged at different positions along the length of the antenna element according to one or more embodiments of the invention. 11 is a graph showing scattering parameters for FIGS. 10 (A) and 10 (B), respectively. 11 is a graph showing scattering parameters for FIGS. 10 (A) and 10 (B), respectively. 1 illustrates an antenna structure including a connecting element with a switch according to one or more embodiments of the present invention. 1 illustrates an antenna structure including a connecting element to which a filter according to one or more embodiments of the invention is coupled; Fig. 3 shows an antenna structure comprising two connecting elements to which a filter according to one or more embodiments of the invention is coupled. 1 illustrates an antenna structure including a tunable connection element according to one or more embodiments of the present invention. 1 illustrates an antenna structure mounted on a PCB assembly according to one or more embodiments of the present invention. FIG. 6 illustrates another antenna structure mounted on a PCB assembly according to one or more embodiments of the present invention. FIG. 6 illustrates an alternative antenna structure that can be mounted on a PCB assembly in accordance with one or more embodiments of the present invention. 3 illustrates a three-mode antenna structure according to one or more embodiments of the present invention. FIG. 19 is a graph showing a gain pattern related to the antenna structure of FIG. 2 illustrates an antenna and power amplifier combiner application of an antenna structure according to one or more embodiments of the present invention. FIG. 6 illustrates a multi-mode antenna structure that can be used in, for example, a WiMAX USB or Express Card / 34 device, according to one or more further embodiments of the present invention. FIG. 6 illustrates a multi-mode antenna structure that can be used in, for example, a WiMAX USB or Express Card / 34 device, according to one or more further embodiments of the present invention. FIG. 21 shows a test assembly used to measure the performance of the antennas of FIGS. 20 (A) and (B). The test measurement results of the antennas of FIGS. 20 (A) and (B) are shown. The test measurement results of the antennas of FIGS. 20 (A) and (B) are shown. The test measurement results of the antennas of FIGS. 20 (A) and (B) are shown. The test measurement results of the antennas of FIGS. 20 (A) and (B) are shown. The test measurement results of the antennas of FIGS. 20 (A) and (B) are shown. The test measurement results of the antennas of FIGS. 20 (A) and (B) are shown. The test measurement results of the antennas of FIGS. 20 (A) and (B) are shown. FIG. 6 illustrates a multi-mode antenna structure that can be used, for example, with a WiMAX USB dongle, according to one or more alternative embodiments of the present invention. FIG. 6 illustrates a multi-mode antenna structure that can be used, for example, with a WiMAX USB dongle, according to one or more alternative embodiments of the present invention. FIG. 6 illustrates a multi-mode antenna structure that can be used, for example, with a WiMAX USB dongle, according to one or more alternative embodiments of the present invention. FIG. 6 illustrates a multi-mode antenna structure that can be used, for example, with a WiMAX USB dongle, according to one or more alternative embodiments of the present invention. FIG. 22 shows a test assembly used to measure the performance of the antennas of FIGS. 21 (A) and (B). The test measurement results of the antennas of FIGS. 21 (A) and (B) are shown. The test measurement results of the antennas of FIGS. 21 (A) and (B) are shown. The test measurement results of the antennas of FIGS. 21 (A) and (B) are shown. The test measurement results of the antennas of FIGS. 21 (A) and (B) are shown. The test measurement results of the antennas of FIGS. 21 (A) and (B) are shown. The test measurement results of the antennas of FIGS. 21 (A) and (B) are shown. The test measurement results of the antennas of FIGS. 21 (A) and (B) are shown. The test measurement results of the antennas of FIGS. 21 (A) and (B) are shown. The test measurement results of the antennas of FIGS. 21 (A) and (B) are shown. The test measurement results of the antennas of FIGS. 21 (A) and (B) are shown. 1 shows a block schematic diagram of an antenna structure with a beam steering mechanism according to one or more embodiments of the present invention. A test measurement result of the antenna of FIG. 25 (A) is shown. A test measurement result of the antenna of FIG. 25 (A) is shown. A test measurement result of the antenna of FIG. 25 (A) is shown. A test measurement result of the antenna of FIG. 25 (A) is shown. A test measurement result of the antenna of FIG. 25 (A) is shown. A test measurement result of the antenna of FIG. 25 (A) is shown. A test measurement result of the antenna of FIG. 25 (A) is shown. FIG. 6 illustrates the gain merit of an antenna structure according to one or more embodiments of the present invention as a function of the phase angle difference between feed points. 1 is a schematic diagram illustrating a simple dual-band branch line monopole antenna structure. FIG. 27 shows a current distribution in the antenna structure of FIG. 6 is a schematic diagram illustrating a branch line band elimination filter. FIG. 28 shows frequency rejection in the antenna structure of FIG. FIG. 28 shows frequency rejection in the antenna structure of FIG. 1 is a schematic illustration of an antenna structure with band elimination slots according to one or more embodiments of the present invention. FIG. 6 illustrates an alternative antenna structure with band elimination slots according to one or more embodiments of the present invention. A test measurement result of the antenna structure of FIG. 29 (A) is shown. A test measurement result of the antenna structure of FIG. 29 (A) is shown. FIG. 3 illustrates an exemplary cylindrical antenna with three ports operable in a single frequency band according to one or more embodiments of the present invention. A cross-sectional view of the antenna of FIG. FIG. 31 is a graph showing a voltage standing wave ratio of the antenna of FIG. FIG. 31 is a graph showing coupling between ports of the antenna of FIG. 30 (A). FIG. FIG. 31 is a graph showing the realized radiation efficiency of the antenna of FIG. 30 (A). FIG. FIG. 31 is a graph showing a correlation between antenna patterns of the antenna of FIG. FIG. 31 is a graph showing a radiation pattern on the azimuth plane of the antenna of FIG. 30 (A). FIG. FIG. 31 is a graph showing a radiation pattern on the azimuth plane of the antenna of FIG. 30 (A) with and without a cable choke. FIG. 31 is a graph showing a radiation pattern on the φ = 90 elevation plane of the antenna of FIG. 30 (A) with and without a cable choke. FIG. 3 illustrates a stamped metal antenna with three ports operable in a single frequency band according to one or more embodiments of the present invention. FIG. 32 shows a PCB assembly using the antenna of FIG. FIG. 32 is a graph showing a voltage standing wave ratio of the antenna of FIG. FIG. 32 is a graph showing the inter-port coupling of the antenna of FIG. FIG. 32 is a graph showing the realized radiation efficiency of the antenna of FIG. FIG. 32 is a graph showing a correlation between antenna patterns of the antenna of FIG. FIG. 32 is a graph showing a radiation pattern on the azimuth plane of the antenna of FIG. 1 illustrates a cylindrical antenna with three ports operable in multiple frequency bands according to one or more embodiments of the present invention. FIG. 33 shows an antenna assembly with a cable using the antenna of FIG. FIG. 33 shows an antenna assembly with a cable using the antenna of FIG. FIG. 33 is a graph showing scattering parameters of the antenna of FIG. FIG. 33 is a graph showing the actual radiation efficiency of the antenna of FIG. 32 (A) in different frequency ranges. FIG. 33 is a graph showing the actual radiation efficiency of the antenna of FIG. 32 (A) in different frequency ranges. 33 is a graph showing peak gains of the antenna of FIG. 32 (A) in different frequency ranges. 33 is a graph showing peak gains of the antenna of FIG. 32 (A) in different frequency ranges. FIG. 4 illustrates a multimode antenna with four ports operable in a single frequency band according to one or more embodiments of the present invention. 34 is a graph showing a voltage standing wave ratio of the antenna of FIG. 34 is a graph showing the inter-port coupling of the antenna of FIG. FIG. 34 is a graph showing the realized radiation efficiency of the antenna of FIG. 1 illustrates a stamped metal antenna with two ports operable in a single frequency band according to one or more embodiments of the present invention. A top view of the antenna of FIG. 34 (A) is shown. A bottom view of the antenna of FIG. 34 (A) is shown. FIG. 34 shows a test assembly using the antenna of FIG. FIG. 35 is a graph showing a voltage standing wave ratio of the antenna of FIG. FIG. 35 is a graph showing the inter-port coupling of the antenna of FIG. 34 (A). FIG. 35 is a graph showing the actual radiation efficiency of the antenna of FIG. FIG. 35 is a graph showing a correlation between antenna patterns of the antenna of FIG. FIG. 35 is a graph showing a radiation pattern on an azimuth plane generated by port 1 of the test assembly of FIG. 34 (D). FIG. 35 is a graph showing a radiation pattern on the φ = 0 elevation plane generated by port 1 of the test assembly of FIG. 34 (D). FIG. 35 is a graph showing a radiation pattern at an elevation angle plane of φ = 90 generated by the port 1 of the test assembly of FIG. 34 (D). FIG. 35 is a graph showing a radiation pattern on an azimuth plane generated by port 2 of the test assembly of FIG. 34 (D). FIG. 35 is a graph showing a radiation pattern on the φ = 0 elevation plane generated by port 2 of the test assembly of FIG. 34 (D). FIG. 35 is a graph showing a radiation pattern on the φ = 90 elevation plane generated by port 2 of the test assembly of FIG. 34 (D). 1 illustrates a stamped metal antenna with two ports operable in multiple frequency bands according to one or more embodiments of the present invention. FIG. 35 shows high and low frequency modes of the antenna of FIG. FIG. 36 shows a test assembly using the antenna of FIG. FIG. 36 is a graph showing scattering parameters of the antenna of FIG. FIG. 36 is a graph showing the realized radiation efficiency of the antenna of FIG. FIG. 36 is a graph showing a radiation pattern in an azimuth plane generated by port 1 of the test assembly of FIG. 35 (C) at 2450 MHz. FIG. 36 is a graph showing a radiation pattern in the φ = 0 elevation plane generated by the port 1 of the test assembly of FIG. 35 (C) at 2450 MHz. FIG. 36 is a graph showing a radiation pattern on the φ = 90 elevation plane generated by the port 1 of the test assembly of FIG. 35 (C) at 2450 MHz. FIG. 36 is a graph showing a radiation pattern in an azimuth plane generated by the port 1 of the test assembly of FIG. 35 (C) at 5150 MHz. FIG. 36 is a graph showing a radiation pattern in the φ = 0 elevation plane generated by the port 1 of the test assembly of FIG. 35 (C) at 5150 MHz. FIG. 36 is a graph showing a radiation pattern on the φ = 90 elevation plane generated by the port 1 of the test assembly of FIG. 35 (C) at 5150 MHz. Fig. 4 illustrates an antenna with four ports operable in multiple frequency bands according to one or more embodiments of the present invention. FIG. 37 is a graph showing scattering parameters of the antenna of FIG. FIG. 37 is a graph showing the realized radiation efficiency of the antenna of FIG.

  In accordance with various embodiments of the present invention, a multimode antenna structure for transmitting and receiving electromagnetic signals in a communication device is provided. These communication devices include circuits for processing signals transmitted and received by the antenna structure. The antenna structure includes a plurality of antenna ports operably coupled to the circuit and a plurality of antenna elements each operably coupled to a different antenna port. The antenna structure further includes one or more connecting elements that electrically connect the antenna elements, the connecting elements having an antenna mode excited by a single antenna port in a given signal frequency range. The antenna elements are electrically connected such that they are generally electrically separated from the mode excited by the other antenna port. Further, the antenna pattern formed by the ports exhibits well-defined pattern diversity with low correlation.

  An antenna structure according to various embodiments of the present invention implements multiple antennas in close proximity (eg, 1/4 wavelength), including communication devices that use multiple antennas simultaneously and particularly within the same frequency band. It is particularly useful in communication devices that need to be less than (separated). Common examples of such devices that can use such an antenna structure are portable handsets, PDAs, and wireless network devices for PCs or their portable communication products such as data cards. These antenna structures are standard protocols for mobile wireless communication devices that require MIMO or many antennas that operate simultaneously (for example, 802.11n for wireless LAN, 802.16e (WiMAX), HSDPA, and 1xEVDO It is also particularly useful in system architectures such as 3G data communications)).

1A to 1G show the operation of the antenna structure 100. FIG. FIG. 1A schematically shows an antenna structure 100 comprising two parallel antennas, specifically parallel dipoles 102, 104 of length L. The dipoles 102 and 104 are separated by a distance d and are not connected by a connecting element. Dipoles 102, 104 have a fundamental resonance frequency approximately equal to L = λ / 2. Each dipole is connected to an independent transmit / receive system, which can operate at the same frequency. This system can have the same characteristic impedance z ₀ for both antennas, in this example 50 ohms.

When one dipole is transmitting a signal, a portion of the signal transmitted by this dipole is directly coupled to the adjacent dipole. The maximum amount of coupling occurs near the half-wave resonance frequency of this dipole and increases as the separation distance d decreases. For example, for d <λ / 3, the magnitude of coupling is greater than 0.1 or -10 dB, d
<For λ / 8, the magnitude of coupling is greater than -5 dB.

  It is desirable to eliminate coupling between antennas (ie, complete separation) or reduce coupling. If the coupling is, for example, -10 dB, 10% of the transmitted power is lost because the amount of power is directly coupled to the adjacent antenna. In addition, undesirable system effects may occur, such as saturation or sensitivity suppression of a receiver connected to an adjacent antenna, and a decrease in performance of a transmitter connected to the adjacent antenna. The current induced in the adjacent antenna distorts the gain pattern as compared to the current generated by individual dipoles. This effect is known to reduce the correlation between gain patterns due to these dipoles. Thus, the combination may cause some pattern diversity, but it also creates undesirable system effects as described above.

  Due to such close coupling, the antennas do not operate independently, and they can be considered as an antenna system with two pairs of terminals or ports corresponding to two different gain patterns. The use of either port will essentially involve the entire structure including both dipoles. Diversity is achieved in the close separation of the dipoles due to parasitic excitation of adjacent antennas, but the current excited by the dipole passes through the source impedance and thus appears as a mutual coupling between the ports.

  FIG. 1C shows a model dipole pair corresponding to the antenna structure 100 shown in FIG. 1 used for the simulation. In this example, the dipoles 102 and 104 have a square cross section of 1 mm × 1 mm and a length (L) of 56 mm. These dimensions produce a central resonant frequency of 2.45 GHz when connected to a 50 ohm source. The free space wavelength at this frequency is 122 mm. The scattering parameters S11 and S12 for a separation distance (d) of 10 mm, ie approximately λ / 12, are shown in FIG. 1D. Due to symmetry and reciprocity, S22 = S11 and S12 = S21. For simplicity, S11 and S12 are shown and described. In this configuration, the coupling between dipoles represented by S12 reaches a maximum of -3.7 dB.

  FIG. 1E shows the ratio of the vertical current in the dipole 104 of the antenna structure to the vertical current in the dipole 102 with the port 106 excited and the port 108 passively terminated (as shown as “Magnitude I2 / I1” in the figure). Show). The frequency at which the current ratio (dipole 104 / dipole 102) is maximum corresponds to the 180 ° phase difference frequency between the dipole currents and is slightly higher in frequency than the maximum coupling point shown in FIG. 1D.

  FIG. 1F shows several frequency azimuth gain patterns due to port 106 excitation. These patterns are not uniformly omnidirectional and vary with frequency depending on the magnitude and phase of coupling changes. Due to symmetry, the pattern resulting from the excitation of port 108 is a mirror image of the pattern of port 106. Thus, as the pattern becomes asymmetric from left to right, the pattern becomes more diverse in terms of gain.

  The calculation of the correlation coefficient between patterns reveals the quantitative features of pattern diversity. FIG. 1G shows the calculated correlation between the port 106 antenna pattern and the port 108 antenna pattern. This correlation is significantly lower than that predicted by Clark's model for an ideal dipole. This is due to differences in the patterns introduced by mutual coupling.

  2A-2F illustrate the operation of an exemplary two-port antenna structure in accordance with one or more embodiments of the present invention. The two-port antenna structure 200 includes closely spaced resonant antenna elements 202 and 204 to achieve low pattern correlation and low coupling between the port 206 and the port 208. FIG. 2A schematically shows a two-port antenna structure 200. This structure is similar to the antenna structure 100 including a pair of dipoles shown in FIG. 1B, but further includes horizontal conductive coupling elements 210, 212 between the dipoles on either side of the ports 206, 208. These two ports 206 and 208 are arranged at the same position as the antenna structure of FIG. When one port is excited, this combined structure exhibits a resonance similar to an uncoupled dipole pair, but with a significant decrease in coupling and increased pattern diversity.

A representative model of an antenna structure 200 with dipoles separated by 10 mm is shown in FIG. 2B. This structure has generally the same geometry as the antenna structure 100 shown in FIG. 1C, but further includes two horizontal coupling elements 210, 212 that electrically connect the antenna elements slightly above and below the port. This structure exhibits a strong resonance at the same frequency as the unconnected dipole, but with very different scattering parameters as shown in FIG. 2C. -20
There is a deep coupling drop of less than dB and a shift in input impedance as shown by S11. In this example, the best impedance match (S11 minimum) does not match the minimum coupling (S12 minimum). As shown in FIG. 2D, a matching network can be used to improve input impedance matching and still achieve very small coupling. In this example, a lumped element matching network including a series inductor and a subsequent shunt capacitor is added between each port and the structure.

FIG. 2E shows the ratio of the current in the dipole element 204 to the current in the dipole element 202 (shown as “magnitude I2 / I1” in the figure) due to the excitation of the port 206. This plot shows that the current actually increases in the dipole element 204 below the resonant frequency. Near resonance, the current in dipole element 204 begins to decrease relative to the current in dipole element 202 with increasing frequency. Minimum coupling (in this case 2.44
GHz) occurs near the frequency at which the current magnitudes of both dipole elements are approximately equal. At this frequency, the phase of the current of the dipole element 204 is delayed by approximately 160 degrees from the current of the dipole element 202.

  Unlike the dipole of FIG. 1C that does not have a connection element, the current of the antenna element 204 of the combination antenna structure 200 shown in FIG. 2B is not forced to pass through the termination impedance of the port 208. Instead, as indicated by an arrow in FIG. 2A, a resonance mode is generated in which a current flows downward through the antenna element 204, crosses the connection elements 210 and 212, and flows upward through the antenna element 202. (Note that this current flow represents half of the resonant cycle, and in the other half of the cycle, the direction of the current is reversed.) The resonant mode of this combined structure is characterized by: That is, (1) the current in antenna element 204 bypasses port 208 approximately, thus achieving high isolation between ports 206 and 208, and (2) the current magnitude in antenna elements 202, 204 is approximately equal, As a result, an uncorrelated and uncorrelated gain pattern can be obtained as will be described later.

  Since the magnitude of the current is approximately the same for these antenna elements, a pattern with a much higher directivity is generated compared to the antenna structure 100 shown in FIG. 1C where no dipole is connected (as shown in FIG. 2F). . When the currents are equal, the condition for zeroing the pattern in the x (i.e., phi = 0) direction is that the phase of the current in the dipole 204 is the amount of π-kd from the current in the dipole 202 (where k = 2π / λ , Λ is the effective wavelength). Under this condition, the field propagating from dipole 204 in the phi = 0 direction is 180 degrees out of phase with that of dipole 202 and the combination of the two should be zero in the phi = 0 direction.

In the example model of FIG. 2B, d is 10 mm or the effective electrical length λ / 12. In this case, kd is π / 6 or 30 degrees, and the condition of the directional azimuth radiation pattern where zero toward phi = 0 and maximum gain toward phi = 180 is that the current of dipole 204 is the current of dipole 202 This is the case when it is delayed by 150 degrees. In the resonant state, the current approaches this state (as shown in FIG. 2E), which explains the pattern directivity. In the case of excitation of the dipole 204, the radiation pattern is opposite to that of FIG. 2F, and the maximum gain is in the phi = 0 direction. The difference in antenna patterns originating from the two ports, as shown in Figure 2G, is related to the low predictive envelope correlation (original:
an associated low predicted envelope correlation). The combined antenna structure thus comprises two ports that are separated from each other and generate a gain pattern with low correlation.

  Therefore, the frequency response of this coupling depends on the characteristics of the connecting elements 210 and 212 including impedance and electrical length. In accordance with one or more embodiments of the invention, the frequency or bandwidth at which the desired amount of separation can be maintained is controlled by appropriately configuring the connecting elements. One way of constructing this cross connection is to change the physical length of the connecting elements. An example of this is illustrated by the multimode antenna structure 300 of FIG. 3A, where meanders have been added to the transverse connections of the connecting elements 310, 312. This has the general effect of increasing the electrical length between the two antenna elements 302 and 304 and increasing the impedance of the connection. The performance characteristics of this structure including scattering parameters, current ratio, gain pattern, and pattern correlation are shown in FIGS. 3B, 3C, 3D, and 3E, respectively. In this embodiment, the change in physical length does not significantly change the resonant frequency of the structure, but there is a significant change in S12, with bandwidth and minimum values compared to structures without this meander. It is getting bigger. That is, the separation performance is optimized or improved by changing the electrical characteristics of the connection element.

  Exemplary multimode antenna structures according to various embodiments of the present invention are designed to excite from ground or counterpoise 402 (as indicated by antenna structure 400 of FIG. 4) or as a balanced structure. (As shown by the antenna structure 500 in FIG. 5). In any case, each antenna structure includes two or more antenna elements (402, 404 in FIG. 4 and 502, 504 in FIG. 5) and one or more conductive connection elements (FIG. 4). 406 and 506, 508 in FIG. For ease of illustration, only a two-port structure is shown in the exemplary drawings. However, this structure can be expanded to include more than two ports according to various embodiments of the present invention. Signal connections or ports (418, 412 in FIG. 4 and 510, 512 in FIG. 5) to the antenna structure are provided on each antenna element. Such a connection element electrically connects two antenna elements at a target frequency or frequency range. Although this antenna is a physically and electrically single structure, its operation can be explained when considered as two independent antennas. For an antenna structure that does not include a connecting element, such as antenna structure 100, port 106 of the structure can be considered connected to antenna 102, and port 108 is considered connected to antenna 104. Can do. However, in the case of a combination structure, such as antenna structure 400, port 418 can be considered to be associated with one antenna mode and port 412 is associated with another antenna mode.

  These antenna elements are designed to resonate at the desired operating frequency or frequency range. The lowest order resonance occurs when the antenna element has an electronic length of one quarter of the wavelength. Thus, for an unbalanced configuration, a simple device design is a quarter-wave monopole. Higher order modes can also be used. For example, a structure formed from a quarter-wave monopole exhibits a dual mode antenna function with high isolation at a frequency three times the fundamental frequency. Therefore, a multiband antenna can be manufactured using a higher-order mode. Similarly, in a balanced configuration, the antenna element can be a complementary quarter-wave element, such as in a half-wavelength center-fed dipole. However, the antenna structure can also be made from other types of antenna elements that resonate at a desired frequency or frequency range. Other possible antenna element configurations include, but are not limited to, inductive shunt formats such as spiral coils, broadband planar shapes, chip antennas, meander shapes, loops, and plate-like inverted F antennas (PIFA).

  The multiple antenna elements of the antenna structure according to one or more embodiments of the present invention need not have the same geometric shape or be the same type of antenna element. Each of these antenna elements should resonate at the desired operating frequency or frequency range.

  According to one or more embodiments of the present invention, all antenna elements of one antenna structure have the same geometric shape. This is generally desirable from a design simplicity perspective, especially if the antenna performance requirements are the same for any port connection.

The bandwidth and resonance frequency of the combination antenna structure can be controlled by the bandwidth and resonance frequency of the antenna element. Thus, a wider bandwidth can be achieved with respect to the mode of the combined structure, as shown in, for example, FIGS. 6A, 6B, and 6C, using elements with wider bandwidth. FIG. 6A shows a multi-mode antenna structure 600 that includes two dipoles 602, 604 connected by connecting elements 606, 608. FIG. The dipoles 602 and 604 each have a width (W) and a length (L) and are separated by a distance (d). Figure 6B shows typical dimensions W =
The scattering parameters for a structure with 1 mm, L = 57.2 mm, and d = 10 mm are shown. FIG. 6C shows the scattering parameters for a structure with representative dimensions W = 10 mm, L = 50.4 mm, and d = 10 mm. As shown in the figure, increasing W from 1 mm to 10 mm while keeping the other dimensions substantially the same increases the separation bandwidth and impedance bandwidth of the antenna structure.

  It has also been found that increasing the separation between antenna elements increases the separation bandwidth and impedance bandwidth of the antenna structure.

  In general, the connecting element exists in a large current region of the combination resonance structure. Therefore, it is desirable that the connection element has high conductivity.

  The port is located at the feed point of the antenna element, as if it were used as a separate antenna. Matching elements or structures can be used to match the port impedance to the desired system impedance.

  According to one or more embodiments of the present invention, as shown in FIG. 7, the multimode antenna structure may be, for example, a planar structure incorporated into a printed circuit board. In this example, the antenna structure 700 includes antenna elements 702 and 704 connected by connecting elements 706 at ports 708 and 710. The antenna structure is fabricated on a printed circuit board material 712. The illustrated antenna element is a simple quarter-wave monopole. However, these antenna elements can be of any geometric shape that provides an equivalent effective electrical length.

  According to one or more embodiments of the present invention, an antenna element with a double resonance frequency can be used to make a combined antenna structure with a double resonance frequency and thus a double operating frequency. FIG. 8A shows a representative model of a multi-mode antenna structure 800, where dipole antenna elements 802, 804 are divided into two fingers 806, 808, and 810, 812, which are not equal in length, respectively. These dipole antenna elements have a resonant frequency associated with each of two different finger lengths, thus exhibiting double resonance. Similarly, as shown in FIG. 8B, a multimode antenna structure using a double-resonant dipole arm also exhibits two frequency bands that provide high isolation (ie, small S21).

  According to one or more embodiments of the present invention, the multimode antenna structure 900 shown in FIG. 9 includes variable length antenna elements 902, 904 that form a tunable antenna. This is achieved by changing the effective electrical length of the antenna element with controllable devices such as RF switches 906, 908 in each antenna element 902, 904. In this example, the switch is opened (by operating a controllable device) to shorten the electrical length (for high-frequency operation), or the switch is closed to increase the electrical length (for low-frequency operation). it can. The operating frequency band of the antenna structure 900, including the high isolation feature, can be tuned by co-tuning both antenna elements. This approach can be used in combination with various methods to change the effective electrical length of the antenna element. For this, a controllable dielectric material is used, a variable capacitor such as a MEMS device, varactor, or tunable dielectric capacitor is combined with the antenna element, and the non-excited element is switched on and off.

  According to one or more embodiments of the present invention, the one or more connecting elements electrically connect the antenna elements with an electrical length generally equal to the electrical distance between these elements. In this state and when these connection elements are attached to the port ends of the antenna element, these ports are separated at a frequency near the resonance frequency of the antenna element. This configuration can achieve almost complete separation at a particular frequency.

Alternatively, as described above, the electrical length of the connecting element can be increased to extend the bandwidth where the isolation exceeds a certain value. For example, a linear connection between antenna elements is -25 at a specific frequency.
A bandwidth with a minimum S21 of dB and S21 <-10 dB can be 100 MHz. Increasing the electrical length increases the minimum S21 to -15 dB,
A bandwidth that is <-10 dB gives a new response that can be increased to 150 MHz.

  Various other multimode antenna structures are also possible according to one or more embodiments of the present invention. For example, the connecting elements can have various geometric shapes, or can be made to include components that change the characteristics of the antenna structure. These components can include, for example, active components such as passive inductor and capacitor elements, resonators or filter structures, or phase shifters.

According to one or more embodiments of the present invention, the characteristics of the antenna structure can be adjusted by changing the position of the connecting element along the length of the antenna element. In order to move the frequency band where the port separation occurs to a higher frequency, the point where the connecting element is attached to the antenna element may be moved away from the port toward the distal end of the antenna element. FIGS. 10A and 10B show multi-mode antenna structures 1000, 1002 each having a connection element that is electrically connected to the antenna element. In the antenna structure 1000 of FIG. 10A, the connection element 1004 is arranged so that the gap between the element 1004 and the upper end of the ground plane 1006 is 3 mm. FIG. 10C shows the scattering parameters of this structure and shows that this configuration provides high separation at a frequency of 1.15 GHz. Impedance matching at 1.15 GHz is achieved using a shunt capacitor / series inductor matching network. FIG. 10D shows the scattering parameters of the antenna structure 1002 of FIG. 10B, and the gap between the connecting element 1008 and the upper end 1010 of the ground plane is 19
mm. The antenna structure 1002 of FIG. 10B exhibits an operating band with high isolation at approximately 1.50 GHz.

  FIG. 11 schematically illustrates a multimode antenna structure 1100 according to one or more further embodiments of the present invention. The antenna structure 1100 includes two or more connection elements 1102 and 1104 that are electrically connected to the antenna elements 1106 and 1108, respectively. (For ease of illustration, only two connecting elements are shown. It should be understood that more than two connecting elements are also contemplated.) The connecting elements 1102, 1104 follow the antenna elements 1106, 1108. Are separated from each other. Connection elements 1102 and 1104 include switches 1112 and 1110, respectively. The peak separation frequency can be selected by controlling the switches 1110 and 1112. For example, the frequency f1 can be selected by closing the switch 1110 and opening the switch 1112. The frequency f2 can be selected by closing the switch 1112 and opening the switch 1110.

  FIG. 12 illustrates a multimode antenna structure 1200 according to one or more alternative embodiments of the present invention. The antenna structure 1200 includes a connecting element 1202, to which a filter 1204 is operably coupled. The filter 1204 can be a low pass filter or a band pass filter, and the connection of the connection elements between the antenna elements 1206 and 1208 can be effective only in a desired frequency band such as a high isolation resonance frequency. At higher frequencies, this structure functions as two separate antenna elements that are not joined by a conductive connecting element that becomes an open circuit.

  FIG. 13 shows a multimode antenna structure 1300 according to one or more alternative embodiments of the present invention. The antenna structure 1300 includes two or more connecting elements 1302, 1304, each including filters 1306, 1308. (For ease of illustration, only two connecting elements are shown. It should be understood that more than two connecting elements are also considered.) In one possible embodiment, two frequency bands are high. To create an antenna structure with isolation (i.e., a dual-band structure), the antenna structure 1300 includes a low pass filter 1308 on the connecting element 1304 (closer to the antenna port) and a wide filter 1306 on the connecting element 1302. I have.

  FIG. 14 illustrates a multimode antenna structure 1400 according to one or more alternative embodiments of the present invention. The antenna structure 1400 includes one or more connecting elements 1402 to which a tunable element 1406 is operatively coupled. The antenna structure 1400 also includes antenna elements 1408 and 1410. The tunable element 1406 changes the delay or phase of the electrical connection or changes the reactive impedance of the electrical connection. Since the scattering parameters S21 / S12 and the magnitude of the frequency response are affected by electrical delays or changes in impedance, the antenna structure is adapted or largely optimized for separation at a particular frequency using the tunable element 1406 it can.

FIG. 15 illustrates a multi-mode antenna structure 1500 according to one or more alternative embodiments of the present invention. The multimode antenna structure 1500 can be used, for example, with a WIMAX USB dongle. The antenna structure 1500 is, for example, 2300 to 2700
Can be configured to operate in the MHz WiMAX band.

The antenna structure 1500 includes two antenna elements 1502 and 1504 connected by a conductive connection element 1506. These antenna elements include slots for increasing their electrical length to obtain a desired operating frequency range. In this example, the antenna structure is 2350
Optimized for MHz carrier frequency. To increase the carrier frequency, the slot length can be shortened. The antenna structure is mounted on a printed circuit board assembly 1508. Two component lumped element matching is obtained with each antenna feed.

  The antenna structure 1500 can be manufactured by, for example, metal punching. What is necessary is just to produce this from the copper alloy sheet of thickness 0.2mm, for example. The antenna structure 1500 includes a pick-up mechanism 1510 on the connecting element at its center of mass, which can be used in an automatic pick-and-place assembly process. The antenna structure is also suitable for surface mount reflow assembly.

FIG. 16 illustrates a multi-mode antenna structure 1600 according to one or more alternative embodiments of the present invention. Similar to the multimode antenna structure 1500 of FIG. 15, the antenna structure 1600 can also be used, for example, with a WIMAX USB dongle. This antenna structure is, for example, 2300 to 2700
Can be configured to operate in the MHz WiMAX band.

The antenna structure 1600 includes two antenna elements 1602, 1604 each including a meander monopole. The length of the meander determines the carrier frequency. The representative design shown is 2350
Optimized for MHz carrier frequency. In order to increase the carrier frequency, the length of the meander can be shortened.

  The connection element 1606 electrically connects the antenna elements. Two component lumped element matching is obtained with each antenna feed.

  This antenna structure can be made of copper as a flexible printed circuit (FPC) mounted on a plastic carrier 1608, for example. This antenna structure can be manufactured by a metal-coated portion of FPC. The plastic carrier provides mechanical support and facilitates attachment to the PCB assembly 1610. Alternatively, the antenna structure may be made from a sheet metal.

FIG. 17 shows a multimode antenna structure 1700 according to another embodiment of the invention. This antenna design can be used, for example, for USB, ExpressCard 34, and Express54 data card formats. The representative antenna structure shown is, for example, 2.3-6.
Designed to operate at GHz frequencies. This antenna structure can be manufactured, for example, from a thin metal plate or by FPC covering the plastic carrier 1702.

FIG. 18A shows a multimode antenna structure 1800 according to another embodiment of the invention. This antenna structure 1800 includes a three-mode antenna with three ports. In this structure, the three monopole antenna elements 1802, 1804, and 1806 are connected using a connection element 1808 including a conductive ring that connects adjacent antenna elements. These antenna elements are balanced by a common counterpoise or sleeve 1810, which is a single hollow conductive cylinder. This antenna includes three coaxial cables 1812, 1814, and 1816 for connecting the antenna structure to a communication device. The coaxial cables 1812, 1814, and 1816 pass through the hollow interior of the sleeve 1810. This antenna assembly can be made from a single flexible printed circuit wound in a cylindrical shape and housed in a cylindrical plastic enclosure to provide a single antenna assembly instead of three separate antennas. In one typical configuration, the cylinder diameter is 10mm and the total length of the antenna is 56mm 2.45.
Operates for high isolation between ports in GHz. This antenna structure can be used with multi-antenna wireless systems such as MIMO or 802.11N systems operating in the 2.4-2.5 GHz band, for example. In addition to port-to-port separation, each port advantageously produces a different gain pattern as shown in FIG. 18B. It should be understood that this is one example and that the structure can be scaled to operate at any desired frequency. It is also understood that the methods of tuning, manipulating bandwidth, and creating multiband structures described above in connection with a two port antenna can be applied to the multiport structure.

  Although the above-described embodiment is illustrated as a true cylinder, the same advantage can be obtained by arranging three antenna elements and connection elements in other manners. This includes, but is not limited to, an arrangement with a straight connection so that the connecting elements form a triangle or other polygonal shape. A similar structure may be produced by connecting three separate monopole elements in the same manner instead of the three monopole elements having a common counterpoise. In addition, the symmetrical arrangement of antenna elements has the advantage of producing equivalent operation from each port, that is, the same bandwidth, separation, and impedance matching, but depending on the application, antenna elements may be arranged asymmetrically or unevenly. it can.

  FIG. 19 illustrates an example of the use of a multi-mode antenna structure 1900 in a combiner application according to one or more embodiments of the present invention. As shown, the transmit signal can be applied to both antenna ports of the antenna structure 1900 simultaneously. In this configuration, the multimode antenna can function as an antenna and power amplifier combiner. The high isolation between the antenna ports limits the interaction between the two amplifiers 1902 and 1904. These interactions have been found to have undesirable effects such as signal distortion and reduced efficiency. An optional impedance match at 1906 is obtained at the antenna port.

20A and 20B illustrate a multi-mode antenna structure 2000 according to one or more alternative embodiments of the present invention. The multimode antenna structure 2000 is, for example, WiMAX
It can be used with USB or ExpressCard / 34 devices. This antenna structure can be configured to operate in a WiMAX band of 2300 to 6000 MHz, for example.

The antenna structure 2000 includes two antenna elements 2001 and 2004, each including a wide monopole. The connection element 2002 electrically connects the antenna elements. 5000 with slot (or other notch) 2005
Improve input impedance matching above MHz. The representative design shown is optimized to cover the frequency range of 2300 to 6000 MHz.

  The antenna structure 2000 can be manufactured, for example, by metal stamping. What is necessary is just to produce this from the copper alloy sheet of thickness 0.2mm, for example. The antenna structure 2000 includes a pick-up mechanism 2003 on the connecting element 2002 at approximately its center of mass, which can be used in an automatic pick-and-place assembly process. The antenna structure is also suitable for surface mount reflow assembly. The antenna feed point 2006 serves as a connection point to the radio circuit on the PCB and also supports the structural attachment of the antenna to the PCB. An additional contact 2007 provides structural support.

FIG. 20C shows a test assembly 2010 used to measure antenna 2000 performance. The figure also shows the coordinate reference for the far field pattern. Antenna 2000 30x representing express card / 34 device
Mounted on 88 mm PCB 2011. The grounded part of PCB 2011 is a larger metal plate 2012 (165x254 in this example) that represents the counterpoise size of a typical notebook computer
mm dimensions). PCB 2011 test ports 2014 and 2016 are connected to the antenna via a 50 ohm stripline.

FIG. 20D shows the voltage standing wave ratio measured at test ports 2014, 2016. FIG. 20E shows the measured coupling (S21 or S12) between these test ports. These voltage standing wave ratios and couplings are suitably low over a wide range of frequencies, eg 2300 to 6000 MHz. FIG. 20F shows the measured radiation efficiency from test ports 2014 (port 1), 2016 (port 2). FIG. 20G shows the calculated correlation between the radiation pattern generated by excitation of test port 2014 (port 1) and the radiation pattern generated by excitation of test port 2016 (port 2). The radiation efficiency is preferably high, but the correlation between the patterns is preferably low at the frequency of interest. Figure 20H shows 2500
The far field electromagnetic gain pattern by excitation of the test port 2014 (port 1) or the test port 2016 (port 2) in the frequency of MHz is shown. Figures 20I and 20J show 3500 and 5200, respectively.
The same pattern measurement at MHz frequency is shown. In φ = 0, that is, XZ plane and θ = 90, that is, XY plane, the pattern obtained from the test port 2014 (port 1) is different from and complementary to the pattern obtained from the test port 2016 (port 2).

21A and 21B illustrate a multi-mode antenna structure 2100 according to one or more alternative embodiments of the present invention. The multimode antenna structure 2100 is, for example, WiMAX
Can be used with USB dongle. The antenna structure can be configured to operate in a WiMAX band of 2300 to 2400 MHz, for example.

  The antenna structure 2100 includes two antenna elements 2102, 2104 each including a meander monopole. The length of the meander determines the carrier frequency. Other serpentine shapes such as helical coils and loops can be used to obtain the desired electrical length. The representative design shown is optimized for a carrier frequency of 2350 MHz. A connecting element 2106 (shown in FIG. 21B) electrically connects the antenna elements 2102, 2104. Two component lumped element matching is obtained with each antenna feed.

  The antenna structure can be made of copper as a flexible printed circuit (FPC) 2103 attached to a plastic carrier 2101, for example. This antenna structure can be manufactured by a metal-coated portion of FPC2103. The plastic carrier 2101 includes a mounting pin or pip 2107 for mounting the antenna to a PCB assembly (not shown), and a pip 2105 for fixing the FPC 2103 to the carrier 2101. The metallized portion of 2103 includes exposed portions or pads 2108 for electrically contacting the antenna with circuitry on the PCB.

  In order to increase the carrier frequency, the lengths of the elements 2102 and 2104 can be shortened. 22A and 22B show a multi-mode antenna structure 2200 whose design is optimized for a carrier frequency of 2600 MHz. Since the metal coating at the ends of elements 2202, 2204 has been removed and the width of the element at the feed end has been increased, the electrical length of elements 2202, 2204 is the electrical length of elements 2102, 2104 in FIGS. 21A and 21B. Shorter than target length.

FIG. 23A shows a test assembly 2300 that uses the antenna 2100 of FIGS. 21A and 21B with the coordinate reference of the far field pattern. FIG. 23B shows the voltage standing wave ratio measured at test ports 2302 (port 1) and 2304 (port 2). FIG. 23C shows the coupling (S21 or S12) measured between test ports 2302 (port 1) and 2304 (port 2). Voltage standing wave ratio and coupling is, for example, 2300 to 2400
Preferably low at frequencies of interest in MHz. FIG. 23D shows the measured radiation efficiency from these test ports. FIG. 23E shows the calculated correlation between the radiation pattern generated by excitation of the test port 2302 (port 1) and the radiation pattern generated by excitation of the test port 2304 (port 2). The radiation efficiency is preferably high, but the correlation between the patterns is preferably low at the frequency of interest. FIG. 23F shows a far field electromagnetic gain pattern due to excitation of test port 2302 (port 1) or test port 2304 (port 2) at a frequency of 2400 MHz. The pattern obtained from the test port 2302 (port 1) is different from and complementary to the pattern obtained from the test port 2302 (port 2) in φ = 0, that is, XZ plane and θ = 90, that is, XY plane.

  FIG. 23G shows the voltage standing wave ratio measured at the test port of assembly 2300 using antenna 2200 instead of antenna 2100. FIG. 23H shows the coupling (S21 or S12) measured between these test ports. The voltage standing wave ratio and coupling are preferably low at frequencies of interest, for example 2500-2700 MHz. FIG. 23I shows the measured radiation efficiency from these test ports. FIG. 23J shows the calculated correlation between the radiation pattern generated by excitation of the test port 2302 (port 1) and the radiation pattern generated by excitation of the test port 2304 (port 2). The radiation efficiency is preferably high, but the correlation between the patterns is preferably low at the frequency of interest. FIG. 23K shows a far field electromagnetic gain pattern due to excitation of the test port 2302 (port 1) or the test port 2304 (port 2) at a frequency of 2600 MHz. The pattern obtained from the test port 2302 (port 1) is different from and complementary to the pattern obtained from the test port 2302 (port 2) in φ = 0, that is, XZ plane and θ = 90, that is, XY plane.

  One or more further embodiments of the invention relate to techniques for controlling the beam pattern for null steering or beam pointing purposes. When such a technique is applied to a conventional array antenna (consisting of separate antenna elements separated by a fraction of the wavelength), each element of the array antenna is given a reference signal or a signal with a phase shifted waveform. For a uniform linear antenna array with equal excitation, the generated beam pattern can be described by an array factor F. This factor F depends on the phase of each individual element and the element spacing d.

上記式で、β = 2π/λ、N =素子の総数、α =連続した素子間の移相、θ =アレイ軸からの角度である。

In the above equation, β = 2π / λ, N = total number of elements, α = phase shift between consecutive elements, and θ = angle from the array axis.

By controlling the phase α to the value α _i , the maximum value of F can be adjusted to different directions θ _i and thus the direction in which the maximum signal is transmitted or received can be controlled.

In many cases, the element spacing in the conventional array antenna is approximately 1/4 wavelength, and the antennas are coupled in close proximity and have substantially the same polarization. Since coupling can cause several problems in array antenna design and performance, it is advantageous to reduce coupling between elements. For example, pattern distortion and scanning blindness (Stutzman,
Antenna Theory and Design, Wiley 1998, 122-128, 135-136, and 466-472) and problems such as the reduction in maximum gain obtained with a given number of elements May arise from binding.

  The technique of controlling the beam pattern is suitable for all multimode antenna structures described herein with antenna elements connected by one or more connecting elements (showing high separation between multiple feed points). Applicable. The antenna pattern can be controlled using the phase between the ports in the high isolation antenna structure. It has been discovered that when antennas are used as simple beamforming arrays, higher peak gain can be achieved in a given direction as a result of reduced coupling between feedpoints. Thus, higher gain antenna structures according to various embodiments using phase control of the carrier signal applied to the feed terminal can achieve greater gain in the selected direction.

  In handset applications where the antennas are separated by a distance much shorter than a quarter wavelength, the mutual coupling effects in conventional antennas reduce the radiation efficiency of the array and thus the maximum achievable gain.

By controlling the phase of the carrier signal applied to each feedpoint of the high isolation antenna according to various embodiments, the direction of the maximum gain generated by the antenna pattern can be controlled. The gain advantage of, for example, 3 dB obtained by beam steering is particularly advantageous in portable device applications where the beam pattern is fixed and the device orientation is randomly controlled by the user. As illustrated, for example, in the schematic block diagram of FIG. 24 illustrating the pattern controller 2400 according to various embodiments, the relative phase shift α is a phase shifter 2402 to the RF signal applied to the antenna feeds 2404, 2408, respectively. Brought about by.
These signals are provided to corresponding antenna ports of the antenna structure 2410.

  The phase shifter 2402 can include standard phase shift components such as, for example, an electrically controlled phase shift device or a standard phase shift network.

  FIGS. 25A-25G illustrate antenna patterns generated by a conventional two-dimensional array of closely spaced dipole antennas and antenna patterns generated by a two-dimensional array of high isolation antennas according to various embodiments of the present invention. A comparison is shown for the different phase differences α between the two feeds to the antenna. 25A to 25G show curves related to the antenna pattern at θ = 90 degrees. The solid line in the figure represents the antenna pattern generated by a separate feed single element antenna according to various embodiments, and the dashed line is two separate conventional types separated by a distance equal to the width of this single element separate feed structure. It represents an antenna pattern generated by a monopole antenna. Therefore, these conventional antennas and high isolation antennas are approximately the same size.

  In all cases shown, the peak gain produced by the high isolation antenna structure according to various embodiments yields a larger gain margin compared to the case of two separate conventional dipoles, while the beam pattern orientation Control is also realized. This operation allows this high isolation antenna to be used in transmission or reception applications where additional gain is necessary or desirable in a particular direction. This direction can be controlled by adjusting the relative phase between the drive point signals. This may be particularly advantageous for portable devices that need to direct energy to a receiving point, such as a base station. Compared to two conventional single antenna elements, the combined high isolation antenna of the present invention has significant advantages when adjusted to similar phases.

  As shown in FIG. 25A, combination dipoles according to various embodiments exhibit gain in a uniform orientation pattern (θ = 90) with respect to α = 0 (zero degree phase difference).

  As shown in FIG.25B, the combined dipole according to various embodiments has a larger peak gain (φ = 90 with an asymmetric orientation pattern (θ = 90 plot for α = 30 (30 degree phase difference between feed points)). 0).

  As shown in FIG.25C, the combined dipoles according to various embodiments have a larger peak gain (φ) with a phase shift orientation pattern (θ = 90 plot for α = 60 (60 degree phase difference between feed points)). = 0).

  As shown in FIG.25D, the combined dipoles according to various embodiments have a larger peak gain (θ = 90 plot for α = 90 (90 degree phase difference between feedpoints)) with a phase shift orientation pattern ( (with φ = 0).

  As shown in FIG.25E, the combined dipoles according to various embodiments have a phase shift orientation pattern (θ = 90 plot, larger back lobe (φ = 180) with respect to α = 120 (120 degree phase difference between feedpoints). )) With a larger peak gain (with φ = 0).

  As shown in FIG. 25F, the combined dipoles according to various embodiments have a phase shift orientation pattern (θ = 90 plot) with respect to α = 150 (150 degree phase difference between feed points), a larger back lobe (φ = It shows a larger peak gain (at φ = 0) with 180).

  As shown in FIG.25G, the combined dipole according to various embodiments has a larger peak gain with a double lobe orientation pattern (θ = 90 plot) with respect to α = 180 (180 degree phase difference between feedpoints) (θ = 90 plot). φ = 0 and 180).

  FIG. 26 illustrates the ideal gain merit of a combined high isolation antenna according to one or more embodiments for two separate dipoles as a function of the phase angle difference between the feedpoints of the two feedpoint antenna array. .

  Another embodiment of the invention relates to a multimode antenna structure that increases high isolation between multiband antenna ports operating adjacent to each other in a given frequency range. In these embodiments, a band elimination slot is incorporated into one of the antenna elements of the antenna structure to reduce coupling at the frequency at which the slot is tuned.

  FIG. 27A schematically shows a simple dual-band branch line monopole antenna 2700. The antenna 2700 includes a band elimination slot 2702 that defines two branch resonators 2704, 2706. This antenna is driven by a signal generator 2708. Various current distributions appear in the two branch resonators 2704 and 2706 according to the frequency at which the antenna 2700 is driven.

  The physical dimensions of the slot 2702 are defined by a width Ws and a length Ls as shown in FIG. 27A. The slot mechanism resonates when the excitation frequency satisfies the condition Ls = lo / 4. At this time, as shown in FIG. 27B, the current distribution is concentrated in the vicinity of the short-circuit section of the slot.

  The currents passing through the branch resonators 2704, 2706 are generally equal and face the opposite along the side of the slot 2702. This causes the antenna structure 2700 to operate in a manner similar to the branch band cancellation filter 2720 (shown schematically in FIG. 27C), which translates the antenna input impedance to a level well below the nominal source impedance. As shown in FIG. 27D, this large impedance mismatch causes a very large voltage standing wave ratio, resulting in the desired frequency rejection.

  This band elimination slot technique works with two (or more) antenna elements where one antenna element needs to pass a signal of the desired frequency and the other does not need to pass, and is close to each other It can be applied to an antenna system provided with an antenna element. In one or more embodiments, one of the two antenna elements includes a band elimination slot and the other does not. FIG. 28 schematically shows an antenna structure 2800, which includes a first antenna element 2802, a second antenna element 2804, and a connection element 2806. Antenna structure 2800 includes ports 2808 and 2810 in antenna elements 2802 and 2804, respectively. In this example, a signal generator drives antenna structure 2802 at port 2808 while a meter is coupled to port 2810 for measuring current at port 2810. However, it should be understood that one or both ports may be driven by a signal generator. The antenna 2802 includes a band elimination slot 2818 that outlines the two branch resonators 2814, 2816. In this embodiment, the branch resonator includes the main transmitter of the antenna structure, while the antenna element 2804 includes the diversity receiver of the antenna structure.

  Due to the large mismatch with the band elimination slot 2812 at the port of the antenna element 2802, the mutual coupling between it and the diversity receiving antenna element 2804 (matching the slot resonant frequency) is very small, resulting in a relatively high separation It becomes.

  FIG. 29A shows a perspective view of a multi-mode antenna structure 2900 including a multi-band diversity receive antenna system that uses a band-erasing slot technique in the GPS band according to one or more alternative embodiments of the present invention. (The GPS band is 1575.42 MHz with a bandwidth of 20 MHz.) The antenna structure 2900 is formed on a flex film dielectric substrate 2902, which is a layer on the dielectric carrier 2904. Is formed. The antenna structure 2900 includes a GPS band elimination slot 2906 in the main transmit antenna element 2808 of the structure. Antenna structure 2900 further includes a diversity reception antenna element 2910 and a connection element 2912 that connects diversity reception antenna element 2910 to main transmission antenna element 2808. A GPS receiver (not shown) is connected to the diversity receiving antenna element 2910. In general, in order to minimize antenna coupling from the main transmit antenna element 2808 and to maximize diversity antenna radiation efficiency at these frequencies, the main antenna element 2908 includes a band cancellation slot 2906 and is electrically connected near the center of the GPS band. Tuned to a fraction of a wavelength. Diversity receive antenna element 2910 does not include such a band cancellation slot, but does include a GPS antenna element that is appropriately matched to the main antenna power supply impedance so that approximately maximum power transfer occurs between it and the GPS receiver. ing. Although antenna elements 2908 and 2910 co-exist at close distances, the high voltage standing wave ratio due to slot 2906 in main transmit antenna element 2908 is related to the main antenna element signal source resistance at the frequency at which slot 2906 is tuned. Reduces coupling and thus achieves separation at the GPS frequency between antenna elements 2908 and 2910. As shown in FIGS. 29B and 29C, the mismatch that occurs between the two antenna elements 2908 and 2910 within the GPS band is so great as to reduce the coupling of these antenna elements and meet the isolation requirements of the system design.

  FIG. 30A shows a multi-mode antenna 3000 according to another embodiment of the present invention. This antenna 3000 includes a three-mode antenna with three ports. In this structure, three monopole antenna elements 3002, 3004, and 3006 are connected using a connection element 3008 that forms a conductive ring that connects adjacent antenna elements. These antenna elements are balanced by a common counterpoise or sleeve 3010, which is a single hollow conductive cylinder. This antenna includes three coaxial cables 3012, 3014, and 3016 for connecting the antenna to a communication device. The coaxial cables 3012, 3014 and 3016 pass through the hollow interior of the sleeve 3010. This antenna assembly, including the sleeve, can be made from a single flexible printed circuit made, for example, from a 1-mil thick polyimide material and 1/2 ounce copper. This flexible printed circuit can be rolled into a cylindrical shape (not shown in FIG. 30A for ease of illustration) and housed in a cylindrical plastic enclosure to provide a single antenna assembly that can replace three separate antennas. .

  The separation between the antenna ports depends at least in part on the characteristics of the connecting ring 3008, including the ring width and the amount of meandering. These parameters can be adjusted to optimize antenna performance for a particular application. For simple monopole elements, if the ports are highly isolated, the input impedance at the connection point of the coaxial cable is generally capacitive and is high impedance compared to a 50 ohm system. The input impedance can be converted using a matching network such as a lumped inductor / capacitor element known in the field of the present invention. However, a modified feed shape can also be used to obtain a good match of 50 ohms without using a lumped matching element. This typical device configuration uses an inductive trace 3018 to move the feed point further into the antenna. This technique simultaneously applies series inductance and impedance transformation to the feed and obtains a good match of 50 ohms at the coaxial cable junction.

Using the quarter-wave sleeve 3010 and routing the cable through the center of the sleeve has the effect of decoupling the cable from the antenna. However, some residual coupling between the antenna and the outer surface of the coaxial cable shield
coupling) may occur. Excitation of current at the outer surface of the coaxial cable shield can cause radiation from the shield, thereby affecting the antenna pattern, or this excitation can cause additional loss through the conductive path from the cable. Yes, both are generally undesirable. If leakage current control is important, an additional quarter-wave choke 3020 may be used under the antenna. The choke 3020 is made of a hollow conductive cylinder that opens at the top (end closer to the antenna) and closes at the bottom. As shown in FIG. 30B, the cable shield and the choke are electrically connected at the common point where the cable exits from the bottom of the choke. In this way, the open end of the choke also exhibits a high impedance for common mode or differential (between cables) signals transmitted along the cable.

  FIGS. 30C-30I show a representative configuration of measured motion where the cylinder diameter is 12 mm and the top of the antenna elements 3002, 3004, 3006 to the bottom of the sleeve 3010 is 39 mm. Operates for high isolation between ports at 2.5GHz. In addition to port-to-port separation, each port advantageously has a different gain pattern as shown in FIG. 30G. For reference, port 1 has the same maximum gain, which is the connection point to element 3002 shown in the orientation of FIG. 30A, in the opposite direction of the connection point, while each antenna port is typically the same rotated 120 degrees. Generates a cardioid radiation pattern.

  The plots of FIGS. 30H and 30I compare the radiation patterns from port 1 that occur with and without the additional cable choke 3020. This indicates that the uniformity and pattern smoothness obtained are improved when choke is added.

  FIG. 31A shows a multimode antenna 3100 according to another embodiment of the invention. This antenna 3100 includes a three-mode antenna having three ports. In this structure, monopole antenna elements 3102, 3104, 3106 are connected via connection elements 3108, 3110, 3112. At the bottom of the antenna, coplanar tabs 3114, 3116, 3118 serve as connection points to the antenna and are suitable for solder connection to a printed circuit board (PCB) assembly. Depending on the shape of the antenna, the antenna can be preferably cut and formed from a single metal plate made of, for example, a copper alloy material having a thickness of 0.2 mm.

As shown in FIG. 31B, the antenna tabs 3114, 3116, 3118 may be attached to the PCB 3120 so that the antenna extends from the edge of the PCB. This PCB has at least one RF ground layer (RF
a ground layer), which acts as a counterpoise to the antenna, but may be used to provide a matching circuit for each antenna port or hold other components such as a communication circuit for an electronic device.

FIGS. 31C to 31G show the simulation performance of a representative configuration designed to operate in the frequency range near 2.5 GHz. In this configuration, the length of the antenna from the edge of the PCB is 28mm, the width is 22mm, the size of the PCB is 50mm
x 50mm. Feed tabs 3114, 3116, and 3118 are connected to ports 3, 2, and 1, respectively. Each of these three ports advantageously provides an antenna mode with a low voltage standing wave ratio, low port-to-port coupling, and high radiation efficiency. In addition, each port generates a unique gain pattern that is less correlated with any of the other antenna patterns.

  FIG. 32A shows a multimode antenna 3200 according to another embodiment of the invention. This antenna 3200 includes a multi-mode antenna with three ports. In this structure, three substantially identical antenna elements 3202 are arranged symmetrically on a cylinder. Each antenna element 3202 includes two branches that generate resonance at two different frequencies, with the longer branch associated with a lower resonant frequency and the shorter branch associated with a higher resonant frequency. These antenna elements are connected to each other by interposing a meandering element 3294 between the antenna elements to form a single structure. The antenna elements are balanced by a common counterpoise or sleeve 3206, which is a single hollow conductive cylinder.

  This antenna includes three coaxial cables 3212, 3214, and 3216 for connecting the antenna to a communication device. The shield of the coaxial cable is electrically attached to the sleeve, while the center conductor is attached to the bottom of the antenna at point 3210 on each antenna element. The coaxial cables 3212, 3214, and 3216 pass through the hollow interior of the sleeve 3206. This antenna assembly, including the sleeve, can be made from a single flexible printed circuit 3208 made from, for example, 1 mil thick polyimide material and 1/2 ounce copper. As shown in FIGS. 32B and 32C, respectively, this flexible printed circuit can be rolled into a cylindrical shape and housed in cylindrical plastic enclosures 3218, 3220 to provide a single antenna assembly that replaces three separate antennas.

  FIGS. 32D through 32H show the measured performance of representative configurations designed to operate in the 2.4 to 2.5 GHz and 5.15 to 5.85 GHz frequency bands. Each of these three ports advantageously provides an antenna mode with a low voltage standing wave ratio, low port-to-port coupling, and high radiation efficiency. In addition, each port generates a unique gain pattern that is less correlated with any of the other antenna patterns.

  FIG. 33A shows a multimode antenna 3300 according to another embodiment of the invention. This antenna 3300 includes a multi-mode antenna with four ports. In this structure, four substantially identical antenna elements are arranged symmetrically on a cylinder. These antenna elements 3302, 3304, 3306, 3308 are connected to each other by a spoke-like cross member 3310 to form one structure. These antenna elements are balanced by a common counterpoise or sleeve 3312, which is a single hollow conductive cylinder.

  A structure of the form of FIG. 30A, where adjacent antenna elements are connected by elements along the circumference of the cylinder, but it is possible to use a structure with four monopoles instead of three . However, in this configuration, the separation is different between the case where the antenna ports are adjacent to each other and the case where the antenna ports are arranged facing each other. One reason for this is that the physical distance between the ports in these two cases is different. Another reason is that the connection between the antenna elements may not be a direct path between the antenna elements on opposite sides of the structure. Instead, the connection path is through adjacent antenna elements, and therefore the separation between adjacent ports is not optimal if the shape is optimized for separation between ports on opposite sides. This difference in isolation generally does not allow all four ports to achieve optimal isolation at the same frequency. In order to solve this more preferably, an interconnection member that passes through the central axis of the antenna may be used as the spoke-like cross member 3310 in FIG. 33A between the elements. This structure achieves a more uniform separation response between all ports.

  The antenna of FIG. 33A can be made, for example, from a single flexible printed circuit made from 1 mil thick polyimide material and 1/2 ounce copper. The flexible printed circuit may be wound around a cylinder. With the slots provided in each antenna element 3302, 3304, 3306, 3308, the cross member 3110 can be inserted into a predetermined position from the top and soldered to the antenna. The cross member 3110 may be made of a thin metal plate such as a copper alloy material having a thickness of 0.2 mm. Alternatively, the entire structure can be punched out of a sheet metal and folded into the shape shown. A coaxial cable can be connected to the feed points 3314, 3316, 3318, 3320 inside the antenna to provide a means for connecting the antenna to the communication device. The antenna has a narrow conductor portion at each feed attachment point, which acts to match the input impedance to 50 ohms as described above.

  Figures 33B, 33C, and 33D show the measured performance of a representative configuration designed to operate in the 5.15 to 5.85 GHz frequency band. Each of these ports advantageously provides an antenna mode with a low voltage standing wave ratio, low port-to-port coupling, and high radiation efficiency. In addition, each port generates a unique gain pattern that is less correlated with any of the other antenna patterns.

  Figures 34A, 34B, and 34C illustrate a multi-mode antenna structure 3400 according to one or more alternative embodiments of the present invention. The antenna 3400 includes two antenna elements 3402 and 3404, which are folded from the upper surface as shown in FIG. 34B to the lower surface as shown in FIG. 34C. The lower surface includes a connecting element 3406 and feed points 3408 and 3410. The bottom surface provides electrical connection to the support means and the PCB assembly. The antenna 3400 can be manufactured by, for example, metal punching. What is necessary is just to produce this from the copper alloy sheet of thickness 0.2mm, for example. This antenna does not require additional support or dielectric material. The antenna is also compatible with surface mount reflow assembly.

  The antenna 3400 can be used with, for example, WIMAX USB using a WiMAX band of 2500 to 2700 MHz. A representative assembly of this type was used to test and evaluate antenna 3400 and is shown in FIG. 34D. Test PCB assembly 3420 includes a two component lumped element matching network for each antenna port.

  34E through 34N show the measurement performance of the test assembly 3420. FIG. Each of these ports advantageously provides an antenna mode with a low voltage standing wave ratio, low port-to-port coupling, and high radiation efficiency. In addition, each port generates a unique gain pattern that is less correlated with any of the other antenna patterns.

  FIG. 35A shows a multimode antenna structure 3500 according to one or more alternative embodiments of the present invention. The antenna structure 3500 includes two antenna elements 3502 and 3504 connected by a connecting portion 3506 made of two strips. This antenna has two feed points 3508, 3510 at the bottom. The bottom surface provides electrical connection to the support means and the PCB assembly. The antenna 3500 can be manufactured by, for example, metal punching. What is necessary is just to produce this from the copper alloy sheet of thickness 0.2mm, for example. This antenna does not require additional support or dielectric material. The antenna is also compatible with surface mount reflow assembly.

  The antenna 3500 is designed to be used in two frequency bands. This is realized by the meander structure of the elements 3502 and 3504. As shown in Figure 35B, these elements support a lower frequency resonance through the longer inductive path of the conductor and a second higher frequency resonance along the coupling path across the gap between the meanders. To do. The antenna 3500 can be used in an 802.11a / b / g / n usable device having an operable frequency band of 2400 to 2500 MHz and 4900 to 6000 MHz, for example. A representative assembly of this type was used to test and evaluate antenna 3500 and is shown in FIG. 35C. Test PCB assembly 3520 includes a two component lumped element matching network for each antenna port.

  FIGS. 35D through 35K show the simulation performance of the test assembly 3520. FIG. Each of these ports advantageously provides an antenna mode with a low voltage standing wave ratio, low port-to-port coupling, and high radiation efficiency. In addition, each port generates a unique gain pattern that is less correlated with any of the other antenna patterns.

  FIG. 36A shows a multimode antenna 3600 according to another embodiment of the invention. This antenna 3600 includes a multimode antenna with four ports. In this structure, four substantially identical antenna elements 3602, 3604, 3606, 3608 are arranged substantially symmetrically on a cylinder. Each antenna element 3602 includes two branches that generate resonance at two different frequencies, with the longer branch associated with a lower resonant frequency and the shorter branch associated with a higher resonant frequency. These antenna elements are connected to each other by interposing spoke-like cross members 3612 between the antenna elements to form one structure. These antenna elements are balanced by a common counterpoise or sleeve 3610, which is a single hollow conductive cylinder.

  The antenna shown in FIG. 36A may be manufactured from a single flexible printed circuit wound into a cylindrical shape. The spoke-like cross member 3610 may be electrically connected by soldering to a flexible printed circuit, for example. The cross member may be made of a thin metal plate such as a copper alloy material having a thickness of 0.2 mm. A coaxial cable can be connected to the feed point inside the antenna to provide a means for connecting the antenna to the communication device.

Figures 36B and 36C show simulated voltage standing wave ratio and realized radiation efficiency for a typical configuration designed to operate in the 2.4 to 2.5 GHz and 5.15 to 5.85 GHz frequency bands.
radiation efficiency).

  Each of the antennas shown in FIGS. 30 to 36 includes two, three, or four antenna elements, but each antenna structure can be configured to include any number of antenna elements connected by connecting elements. Should be understood.

  Further, the antennas shown in FIGS. 30, 32, 33, and 36 can have a cylindrical or polyhedral shape (ie, comprising a plurality of flat surfaces).

  In the antennas described herein according to various embodiments of the present invention, the antenna elements and connecting elements are not separate radiating structures, but preferably form a single, integral radiating structure. The signal given to the port is also radiated as a whole by exciting the entire antenna. Thus, the techniques described herein provide for antenna port isolation without using decoupling circuitry at the antenna feedpoint.

  Although the present invention has been described with reference to particular embodiments, it is to be understood that the above-described embodiments are intended to be illustrative only and are not intended to limit or limit the scope of the invention.

  Various other embodiments, including but not limited to the following description, are within the scope of the claims. For example, the elements or components of the various multimode antennas described herein may be further divided or combined into additional components that perform the same function, and fewer configurations that perform the same function. It may be an element.

  While preferred embodiments of the invention have been described, it should be apparent that modifications can be made without departing from the spirit and scope of the invention.

  The claims are as follows.

Claims (30)

A multimode antenna structure for transmitting and receiving electromagnetic signals in a communication device, wherein the communication device includes a circuit for processing signals transmitted and received by the antenna structure,
A plurality of antenna ports coupled to the circuit;
A plurality of antenna elements each operably coupled to a different one of the antenna ports;
A plurality of connecting elements, each electrically connecting adjacent antenna elements, wherein the antenna element and the connecting element are disposed along a periphery of the antenna structure, and a single radiating structure is formed Forming a plurality of connecting elements;
The antenna element comprises an inductive trace extending between a feed point connected to the antenna port and a point on each antenna element;
The current on one antenna element flows to the connected adjacent antenna element and bypasses the antenna port coupled to said adjacent antenna element, and the antenna mode excited by one antenna port is given the desired A multi-mode antenna structure that is electrically separated from a mode excited by another antenna port in the signal frequency range, and wherein the antenna structure generates a diversity antenna pattern.   2. The multimode antenna structure according to claim 1, wherein the plurality of antenna elements include three antenna elements, and the plurality of connection elements include three connection elements.   The multimode antenna structure according to claim 1, wherein the antenna elements are balanced by a common counterpoise.   4. The multimode antenna structure according to claim 3, wherein the common counterpoise includes a hollow conductive cylinder. 5. The multimode antenna structure of claim 4, wherein each of the feed points is coupled to a cable extending through the cylinder.   6. The multimode antenna structure according to claim 5, further comprising a choke with a hollow conductive cylinder through which each of the cables passes.   7. The multi-mode antenna structure according to claim 6, wherein each of the cables is a coaxial cable with a cable shield electrically connected to the choke at a common point.   2. The multimode antenna structure according to claim 1, wherein each of the plurality of connection elements has a meandering shape so as to have a given electrical length.   The multi-mode antenna structure according to claim 1, wherein the multi-mode antenna structure is made of a flexible printed circuit.   10. The multimode antenna structure according to claim 9, wherein the flexible printed circuit is wound in a cylindrical shape and accommodated in a cylindrical plastic enclosure.   The antenna structure is formed from a thin metal plate and includes a plurality of coplanar tabs each connected to an antenna element, the coplanar tabs configured to be secured to an edge of a printed circuit board assembly. 2. The multimode antenna structure according to claim 1, wherein:   The multimode antenna structure according to claim 1, wherein each antenna element includes two branches having different lengths that generate resonance at two different frequencies. A multimode antenna structure for transmitting and receiving electromagnetic signals in a communication device, wherein the communication device includes a circuit for processing signals transmitted and received by the antenna structure,
A plurality of antenna ports coupled to the circuit;
A plurality of antenna elements each operably coupled to a different one of the antenna ports, the plurality of antenna elements disposed along a periphery of the antenna structure;
A connecting element that electrically connects the antenna elements to a common point to form a single radiating structure;
The antenna element comprises an inductive trace extending between a feed point connected to the antenna port and a point on each antenna element;
The current on one antenna element flows to another antenna element and bypasses the antenna port coupled to the other antenna element, and the antenna mode excited by one antenna port is within a given desired signal frequency range. A multi-mode antenna structure, wherein the antenna structure generates a diversity antenna pattern that is electrically separated from a mode excited by another antenna port. The multi-mode antenna structure according to claim 13 , wherein the plurality of antenna elements include four antenna elements. The multimode antenna structure of claim 13 , wherein the antenna elements are balanced by a common counterpoise. The multimode antenna structure of claim 15 , wherein the common counterpoise comprises a hollow conductive cylinder. The multi-mode antenna structure of claim 16 , wherein each feed point is coupled to a cable extending through the cylinder. The multimode antenna structure of claim 17 , further comprising a choke including a hollow conductive cylinder through which each of the cables passes. The multimode antenna structure of claim 18 , wherein each of the cables is a coaxial cable with a cable shield electrically connected to the choke at a common point. The multimode antenna structure according to claim 13 , wherein the antenna structure is formed of a thin metal plate. The multimode antenna structure of claim 13 , wherein the monopole antenna element is fabricated from a flexible printed circuit. The multi-mode antenna structure according to claim 21 , wherein the connection element is soldered to the flexible printed circuit. The multi-mode antenna structure according to claim 13 , wherein each antenna element includes two branches having different lengths that generate resonance at two different frequencies. The multi-mode antenna structure according to claim 13 , wherein the connection element has a spoke shape. A multimode antenna structure for transmitting and receiving electromagnetic signals in a communication device, wherein the communication device includes a circuit for processing signals transmitted and received by the antenna structure,
A plurality of antenna ports coupled to the circuit;
A plurality of antenna elements each operably coupled to a different one of the antenna ports, each parallel and spaced apart upper and lower planar portions and side portions connecting the upper and lower planar portions A plurality of antenna elements, including
One or more connecting elements, each electrically connecting adjacent antenna elements at one of the planar portions, wherein the antenna elements form a single radiating structure And connecting elements of
The antenna element comprises an inductive trace extending between a feed point connected to the antenna port and a point on each antenna element;
A current on one antenna element flows to one connected adjacent antenna element, bypasses the antenna port coupled to the adjacent antenna element, and flows through the one antenna element and the adjacent antenna element The currents are equal in magnitude, and in a given desired signal frequency range, an antenna mode excited by one antenna port is electrically separated from a mode excited by another antenna port, and the antenna structure is A multimode antenna structure that generates a diversity antenna pattern. The multi-mode antenna structure according to claim 25 , wherein each of the plurality of connection elements has a meandering shape so as to provide a given electrical length. 26. The multi-mode antenna structure of claim 25 , wherein each of the plurality of antenna elements has a serpentine shape to provide a given electrical length. The multimode antenna structure according to claim 25 , wherein the multimode antenna structure is made of a thin metal plate. The multi-mode antenna structure according to claim 25 , wherein two connection elements electrically connect adjacent antenna elements. 26. The multimode antenna structure of claim 25 , wherein the lower planar portion of each antenna element is connected to a printed circuit board assembly.

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