201108595 六、發明說明： 【發明所屬之技術領域】 本發明係則卜健·流馬達_方法，侧是關於磁場路徑具凸 極特性的凸極無刷直流馬達驅動方法。 【先前技術】 . 在_節能減碳觀念_時’提高各式機具的能源轉換效率是非常關鍵 的議題之，尤其眾多工商產業而耗能機器使用的電動馬達更是影響能源 籲轉換效率之關鍵要角。近年來又由於無刷直流馬達製造技術不斷突破，經 發展後已具備有高功率密度、高效率、免維護，體積小、重量輕以及結構 堅固等諸多優點’是以逐漸取代傳統有刷直流馬達以及部份感應馬達而佔 據市場。 -般習用磁路對稱分佈的無刷直流馬達，其驅動方法係採用導通區 間波寬調變法，使反流器開關以固定順序切換，同時間馬達繞組依此順序 進行2相繞組激磁’以期達到相電流與相反電勢同相位的最佳功率輸出。 籲就如第九圖顯示磁路對稱無刷直流馬達之理想A相反電勢電壓〜波形及 反流器6個功率開關習用的切換模式示意圖’理想磁路對稱無刷直流馬達 •中，其A相繞組電流正半週期介於第九圖的3〇。至⑼。之間，負半週期繞組 電流介於第九_训。至加之間，使其能在額定轉速以下驅動馬達運轉； 再參照第十圖，顯示-理想八相反電勢電壓‘波形及反流器6個功率開關 的弱磁切換模式，此圖裡為了說明方便，做定切換模式超前第九圖約^ 角第十圖所示’理想無刷直流馬達之A相繞組電流正半週期介於第十 圖約15至135之間’負半週期繞組電流介於第十圖的⑽至奶。之間，使 201108595 之能在額定轉速以上驅動馬達高速運轉。 然而，無刷直流馬達固定轉子位置產生驅動信號的驅動方法，並無法在 各種轉速下產生最佳轉矩輸出，因而有如我國專利證號483231之「依轉速 調整換相時機之無刷直流馬達控制方法」所採的相位超前弱磁驅動方法之 先前技術被提出，其係根據轉速資訊採用120°換向相位超前法驅動技術， 這種切換方法可使馬達轉速高於額定轉速，但此方法不適用於凸極無刷直 流馬達運轉於低速高轉矩的情況。另一先前技術，如我國專利證號1259648 之「電動車直流無刷馬達之控制方法及裝置」所採用的相位超前弱磁驅動 亦是用120°驅動技術，惟，其需要額外的感測器量測直流電壓，以及需要 將馬達轉矩命令再結合轉速才能共同決定適當驅動時機進行電動車無刷直 流馬達運轉。 近來，有一種磁場路徑具&極特性的無刷直流馬達，其轉矩成分除電磁 轉矩外，還增加一磁阻轉矩，進而能提升輸出轉矩能力，另因其轉子磁石 可以欲入轉子内部’使得磁;5不易因高速旋轉飛脫，故目前已廣泛應用於 各式電動車以及電動機具領域，此種凸極無刷直流馬達驅動方法基本上與 前述習用磁路對稱分佈的無刷直流馬達相近，但仍與學理所證明之最大能 源轉換效率有頗大差距存在。 因此，本發明在磁場路徑具凸極特性的無刷直流馬達的驅動中提 供-種創新的凸極無刷直流馬達_方法，以達成發揮其較佳性 能目的。 【發明内容】 本發明之主要目的在提供一種凸極無刷直流馬達驅動方法， 201108595 在低轉速高轉矩的重載到弱磁高轉速時，都可發揮學理的高能源 轉換效率。 為達到上述之目的，本發明之凸極無刷直流馬達驅動方法乃 . 根據基本學理分析獲得的知識發現，可藉馬達實際轉速與反流器 ， 功率開關的工作週期，結合轉子位置資訊後決定驅動反流器的電 壓大小及相位，再透過軟體程式轉換為驅動反流器的控制信號， 以驅動凸極無刷直流馬達達到所需的轉速及轉矩。故，首先提供 # 一反流器電連一凸極無刷直流馬達，以用反流器之切換開關控制 輸出馬達之電壓大小與相位，接著提供一數位化反流器控制裝置 電性連接反流器以及凸極無刷直流馬達，於數位化反流器控制裝 置中設定一指定轉速，再擷取凸極無刷直流馬達之一轉速，以利 用該轉速與該指定轉速計算反流器之切換開關一工作週期以及一 最大工作週期；於上述步驟結束之後，再取該凸極無刷直流馬達 之一額定轉速以及一最大超前驅動電壓相角予該數位化反流器控 • 制裝置，藉以配合該工作週期及該最大工作週期計算一超前驅動 電壓相角，接著擷取凸極無刷直流馬達之一轉子區域以及一轉子 • 角度，由數位化反流器控制裝置根據轉子區域、轉子角度以及該 - 超前驅動電壓相角間之關係調配一控制訊號，以由反流器接收該 控制訊號而驅動凸極無刷直流馬達達到所需的轉速以及轉矩。 本發明之目的或其他目的對於此技藝之通常知識者而言，閱 讀以下實施例之詳細内容後係顯而易知的。先前的概述與接下來 的詳細敘述都是範例，以便能進一步解釋本發明之專利請求項。 201108595 【實施方式】 針對凸極永磁同步馬達驅動方法，在學理上許多文獻已有研究報告， 例如刊登於 2008 年元月 IEEE Transactions on Industrial Electronics 國際期 刊上的論文“Voltage Constraint Tracking Based Field Weakening Control 〇f IPM Synchronous Motor Drives”已經闡述相關理論基礎，能使凸極永磁同步 馬達在高速到低速，小轉矩到大轉矩，同時發揮電磁轉矩及磁阻轉矩（一 般磁路對稱同步馬達不具有磁阻轉矩成分），並且達到更高電能轉換效率， 然，一般凸極無刷直流馬達的磁路分佈呈現非弦波，在數學上的理論分析 有非線性特性’在工程上’其驅動的基本原_與磁路弦波分佈的凸極永 磁同步馬達種，故’本紐之研發動機源於要驅動凸極無刷直流馬達而 提出’其精神在選擇反流H 6個功糊_切換模式與時機須考量馬達在 不同運轉需求情況下的轉速資訊及功率開關工作週期資訊。 請參照第-圖，係藉讀現本發明之凸極無織流馬達驅動方法 之-系統示意圖。如圖示之系統100，其係包括一凸極無刷直流馬達n， -反流器12祕凸極無刷直流馬達u再外接_電壓源i3，以接收一直流 電屋並將其轉成-交流電壓輸出至凸極無刷直流馬達ι卜其中反流器以系 一三相全橋反流n ’㈣六偏率關S2、S3、s4、沾及％，就如 第二圖所示’每—功率開關S1、S2、S3、S4、s5、％包括—功率電晶體與 一她接的背接二極體，故反流器12透過關S1至開關％可控制輸出至 凸極無刷直流馬達u的電壓大小與相位…轉子位置細器μ _凸極 無刷直流馬達u，該轉子位置偵測器14可由三個霍爾感測器組成，用以横 測馬達U之轉子磁石與定子繞組的相對位置而輪出三個位置信號㈣、W。 201108595 一負載15耗接馬達11與轉子位置偵測器14。 -數位化反流n控織置16_轉子位置細^ 14，且數位化反流器 控制裝置1(5為實現本發明方法的中心單元，其包括一轉子位置區域偵測單 το I6卜-轉速計算單元162、一速度控制及電流限制單元⑹及一控制信 號單元164 ’其中’轉子位置區域偵測單元161 _接轉子位置偵測器14以 接收轉子位置雜u、v、w，經辨識及制後再輸出—轉子區域及一轉子 角度;而轉速計算單元162 —樣祕轉子位置細器丨4而接收三個位置 信號u v w ’藉以叶算馬達u之轉速叫；另提供數位化反流器控制裝置 16-心疋轉迷％，並將馬達u的轉速％以及指定轉速<與轉速叫間之一 誤差同時_至速紐做電祕解元163 t，此時，速雜制及電流 限制單元163將根據轉速叫與該誤差之關係判斷輸出反流器12切換開關的 工作週期D與-最大卫作週期I。控輸鮮元164祕轉子位置區 域偵測單元⑹、轉速計算單元162、速度控制及電流限制單元⑹，以接 收轉子區域、轉子角度^轉速叫、最大工作週期D眶與工作週期d•之參 數，再獲得控制反流器12的六個功率開關控制信號。且該數位化反流器控 制裝置16是透過數位化方式實現，可採任何市售的單晶片微處理器、數位 信號處理器、可規劃邏輯陣列、各種形式電腦或其他相似功能的數位化處 理器等。 此外，就純無刷纽馬達運轉之動作原理而言，大略可劃分成如下 幾種狀I低速讎矩時，較佳的繞組電流猶反電勢的㈣相對較小， 故’低速低轉矩時控制反流器電壓所需超前相角可只由轉速大小決定。而 低速高轉矩時’轉關玉作·增加，較佳的繞組電流超歧電勢的相 201108595 角相對較大，故，控制反流器電壓所需超前相角也需加大，此時是由功率 開關工作週期配合轉速的大小來決定所需的超前相角。以及，於高速弱磁 運轉時’較佳的繞組電流超前反電勢的相角相對較大，功率開關工作週期 呈增加趨勢’控制反流器電壓所需超前相角也必須加大，此時反由轉速配 合功率開關工作週期決定所需的超前相角。 根據前述系統100結構說明，以及凸極無刷直流馬達動作原理，本發 明進行凸極無刷直流馬達驅動方法時，主要是當開關工作週期D·低於一臨 界工作週期Dth時，就只由轉速大小決定控制反流器12電壓所需超前驅動 電壓相角thadv。因此就如第一圖所示般，數位化反流器控制裝置16控制 反流器12輸A至凸極觸錢馬達u的電壓大小與相位，所以本發明之 驅動方法首先如第三圖之步娜所示，提供馬達u之指定轉速<參數予 數位化反流器控制裝置16，以由控制信號單元164藉由馬達u轉速历以及 該轉速％與該指定轉速<之誤差參數制反流器U切換開_工作週期 D以及最大工作週期Dmx，由於所有參數，係包括：轉子區域、轉子角度0、 轉速叫、最大工作週期D贿與工作週期D·終將進入控制信號單幻64中， 故’再配合第四圖本發明控制信號單幻64之邏輯作動流程示意圖所示， 結束步驟si之後叙步驟％時，係先提供—臨界工作猶μ控制信號 單兀164 ’並進行步驟S2卜由控制信號單元164判斷開關工作週期〇·是否 低於臨界工作週期’若是，進行步驟％2,利用馬達u之—額定轉速必 參數以及-最大超前驅動電_角thadv麵參數計算馬達u之超前驅動; 廢相角thadv ’其公式（1)如下列所示· 201108595 thadv = fthadv„201108595 VI. Description of the Invention: [Technical Field] The present invention is a method of driving a salient pole brushless DC motor with a salient characteristic of a magnetic field path. [Prior technology] In the _ energy-saving and carbon-reduction concept _ when 'increasing the energy conversion efficiency of various types of equipment is a key issue, especially the industrial and industrial industries and the electric motor used by energy-consuming machines is the key to affect the efficiency of energy conversion The corner. In recent years, due to the continuous breakthrough in the manufacturing technology of brushless DC motor, it has been developed with high power density, high efficiency, maintenance-free, small size, light weight and strong structure. It is gradually replacing the traditional brushed DC motor. And some induction motors occupy the market. Generally, a brushless DC motor with a magnetic circuit symmetric distribution is used, and the driving method adopts a conduction interval wave width modulation method to switch the inverter switch in a fixed order, and at the same time, the motor winding performs the excitation of the 2-phase winding in this order. The optimum power output of the phase current in phase with the opposite potential. As shown in the ninth figure, the ideal phase A reverse potential voltage of the magnetic circuit symmetrical brushless DC motor ~ waveform and the switching mode of the six power switches of the inverter are used in the schematic diagram of the ideal magnetic circuit symmetrical brushless DC motor. The positive half cycle of the winding current is between 3〇 in the ninth figure. To (9). Between the negative half cycle winding current is between the ninth and the training. Between the two, it can drive the motor to run below the rated speed; refer to the tenth figure, showing the - ideal eight opposite potential voltage 'waveform and the weak magnetic switching mode of the six power switches of the inverter. , the fixed switching mode is ahead of the ninth figure. The eleventh figure shows that the positive half cycle of the A phase winding current of the ideal brushless DC motor is between the 15th and the 135th of the tenth figure. (10) to milk in the tenth figure. Between the 201108595, the motor can be driven at a high speed above the rated speed. However, the brushless DC motor fixes the rotor position to generate the driving signal driving method, and can not produce the optimal torque output at various speeds. Therefore, as in China Patent No. 483231, the brushless DC motor control according to the speed adjustment commutation timing The prior art of the phase advance weak magnetic drive method adopted by the method is proposed, which adopts a 120° commutation phase advance driving technique according to the rotational speed information. This switching method can make the motor speed higher than the rated speed, but this method does not It is suitable for the case where the salient pole brushless DC motor operates at low speed and high torque. Another prior art, such as the "Phase Control Method and Device for Electric Vehicle DC Brushless Motor" of China Patent No. 1259648, uses a phase-leading field-weakening drive that uses 120° drive technology, but requires an additional sensor. The DC voltage is measured, and the motor torque command needs to be combined with the rotational speed to jointly determine the appropriate driving timing for the electric vehicle brushless DC motor. Recently, there is a brushless DC motor with a magnetic field path and an extreme characteristic. The torque component adds a reluctance torque in addition to the electromagnetic torque, thereby increasing the output torque capability, and the rotor magnet can be used for the rotor magnet. The inside of the rotor makes magnetic; 5 is not easy to fly off due to high-speed rotation, so it has been widely used in various electric vehicles and electric motors. The salient-pole brushless DC motor driving method is basically symmetrically distributed with the conventional magnetic circuit. The brushless DC motor is similar, but it still has a large gap with the maximum energy conversion efficiency proved by the theory. Accordingly, the present invention provides an innovative salient brushless DC motor method for driving a brushless DC motor having a salient pole characteristic of a magnetic field path for the purpose of achieving its superior performance. SUMMARY OF THE INVENTION The main object of the present invention is to provide a salient-pole brushless DC motor driving method, and the high-energy conversion efficiency can be achieved in the low-speed high-torque heavy-duty to weak magnetic high-speed. In order to achieve the above object, the salient pole brushless DC motor driving method of the present invention is based on the knowledge obtained by the basic theoretical analysis, and can be determined by combining the actual rotational speed of the motor with the inverter, the duty cycle of the power switch, and the rotor position information. The voltage magnitude and phase of the driving inverter are converted into a control signal for driving the inverter through the software program to drive the salient pole brushless DC motor to achieve the required speed and torque. Therefore, firstly, a #-inverter is connected to a salient-pole brushless DC motor to control the voltage magnitude and phase of the output motor by using a switch of the inverter, and then a digital inverter control device is electrically connected. a flow brush and a salient pole brushless DC motor, set a specified rotational speed in the digital inverter control device, and then take a rotational speed of one of the salient pole brushless DC motors to calculate the reflux device by using the rotational speed and the specified rotational speed Switching the switch to a duty cycle and a maximum duty cycle; after the end of the step, taking a nominal speed of the salient brushless DC motor and a maximum lead drive voltage phase angle to the digital inverter control device, Calculating a leading drive voltage phase angle in accordance with the duty cycle and the maximum duty cycle, and then drawing a rotor region of the salient-pole brushless DC motor and a rotor angle, the digital inverter control device according to the rotor region and the rotor The angle and the relationship between the phase angles of the leading drive voltages are matched with a control signal to receive the control signal by the inverter to drive the salient pole without brush Flow motor up to speed and the torque. The objectives and other objects of the present invention will become apparent to those skilled in the <RTIgt; The previous summary and the following detailed description are examples to further explain the patent claims of the present invention. 201108595 [Embodiment] For the salient-pole permanent magnet synchronous motor driving method, many studies have been published in the literature, such as the paper published in the 2008 IEEE IEEE on the International Journal of International Electronics "Voltage Constraint Tracking Based Field Weakening Control 〇f IPM Synchronous Motor Drives has already explained the relevant theoretical basis, which can make salient-pole permanent magnet synchronous motor at high speed to low speed, small torque to large torque, and at the same time exert electromagnetic torque and reluctance torque (general magnetic circuit symmetry synchronization The motor does not have a reluctance torque component), and achieves higher power conversion efficiency. However, the magnetic circuit distribution of a general salient brushless DC motor exhibits a non-chord wave, and the mathematical analysis has nonlinear characteristics. 'The basic original of its drive _ and the salient-pole permanent magnet synchronous motor of the magnetic path sine wave distribution, so 'the engine of this New York originates from the drive of the salient pole brushless DC motor and puts its spirit in choosing the reverse flow H 6 A little work _ switching mode and timing must consider the motor speed information and power switcher under different operating requirements Cycle information. Please refer to the first figure, which is a system schematic diagram of the salient pole non-woven motor driving method of the present invention. The system 100 as shown includes a salient pole brushless DC motor n, a reflector 12, a salient brushless DC motor u, and an external _voltage source i3 to receive the all-current electricity house and convert it into - The AC voltage is output to the salient pole brushless DC motor. The inverter is connected to a three-phase full-bridge reverse flow n '(four) six-bias ratio off S2, S3, s4, and %, as shown in the second figure' Each of the power switches S1, S2, S3, S4, s5, % includes a power transistor and a back-connected diode connected to the other, so that the inverter 12 can control the output to the salient pole without brushing through the switch S1 to the switch %. The voltage magnitude and phase of the DC motor u...the rotor position closer μ _ salient brushless DC motor u, the rotor position detector 14 can be composed of three Hall sensors for transversely measuring the rotor magnet of the motor U Three position signals (4), W are rotated out of the relative positions of the stator windings. 201108595 A load 15 consumes the motor 11 and the rotor position detector 14. - digitally inverted n-controlled woven 16_rotor position fine 14, and digitally configurable control device 1 (5 is a central unit for implementing the method of the present invention, which includes a rotor position area detection unit το I6 - The rotational speed calculation unit 162, a speed control and current limiting unit (6) and a control signal unit 164' wherein the 'rotor position region detecting unit 161_ is connected to the rotor position detector 14 to receive the rotor position mis-u, v, w, and is identified And after the system is output - the rotor area and a rotor angle; and the rotational speed calculation unit 162 - the sample rotor position 丨 4 receives the three position signals uvw ' by the speed of the leaf motor u; and provides digital reversal The device control device 16-heart 疋 疋 %, and the motor u's speed % and the specified speed < one of the errors between the speed and the speed _ to the speed of the new secret 163 t, at this time, the speed of miscellaneous The current limiting unit 163 determines the duty cycle D of the output inverter 12 switching switch and the maximum guarding period I according to the relationship between the rotational speed and the error. The control unit 164 the rotor rotor position detecting unit (6) and the rotational speed calculating unit 162, speed control The current limiting unit (6) receives the parameters of the rotor region, the rotor angle ^the rotational speed, the maximum duty cycle D眶 and the duty cycle d•, and then obtains six power switch control signals for controlling the inverter 12, and the digital backflow The device control device 16 is implemented by digitalization, and can be used in any commercially available single-chip microprocessor, digital signal processor, programmable logic array, various forms of computers or other similarly functioning digital processors. In the principle of operation of the pure brushless New Motor, it can be roughly divided into the following types of low-speed moments. The preferred winding current is relatively small (4), so the control is reversed at low speed and low torque. The leading phase angle required for the voltage of the device can be determined only by the magnitude of the rotational speed. When the low speed and high torque are used, the phase of the better winding current super-distribution potential is relatively large, so the control inverter is relatively large. The leading phase angle required for the voltage also needs to be increased. At this time, the required phase angle is determined by the power switch duty cycle and the speed. And, during high-speed field weakening operation, 'better The phase angle of the group current leading back EMF is relatively large, and the duty cycle of the power switch is increasing. 'The leading phase angle required to control the inverter voltage must also be increased. At this time, the speed required to match the power switch duty cycle determines the required advance. According to the structure description of the foregoing system 100, and the principle of operation of the salient pole brushless DC motor, when the salient pole brushless DC motor driving method of the present invention is performed, mainly when the switching duty cycle D· is lower than a critical duty cycle Dth, The leading drive voltage phase angle thadv required to control the voltage of the inverter 12 is determined only by the magnitude of the rotational speed. Therefore, as shown in the first figure, the digital inverter control device 16 controls the inverter 12 to transmit the A to the salient pole contact. The voltage magnitude and phase of the money motor u, so the driving method of the present invention first provides the specified rotational speed of the motor u < parameter to the digitizing inverter control device 16 to be controlled by the signal unit as shown in the step of the third figure. 164 by the motor u tachometer and the rotation speed % and the specified speed < error parameter system inverter U switch open _ work cycle D and maximum duty cycle Dmx, due to all parameters The system includes: rotor area, rotor angle 0, speed call, maximum duty cycle D bribe and work cycle D· will enter the control signal single fantasy 64, so 'further cooperate with the fourth figure, the logic signal of the control signal of the invention is 64 As shown in the flow diagram, when step % is finished and step % is followed, the critical operation is first provided to control signal unit 兀 164 ' and step S2 is performed. Control signal unit 164 determines whether the switching duty cycle 〇 · is lower than the critical duty cycle. 'If yes, proceed to step %2, use the motor u's - rated speed must parameter and - maximum lead drive electric _ angle thadv surface parameter to calculate the motor u's advanced drive; waste phase angle thadv 'the formula (1) as shown below 201108595 thadv = fthadv„
-H (1); 倘若步驟S21答案為否’此時超前驅動電壓相角thadv仍為未知者，因 此進行步驟S23以及步驟S24 ’分別計算二暫時超前驅動電壓相角thadv丨、 thadv2 ’再計算該超前驅動電壓相角thadv，其_步驟S23係先計算第一個 暂時超前驅動電壓相角thadw，其如公式(2)所示： thadv, ,thadvra =(—— (2)；-H (1); If the answer in step S21 is no 'At this time, the lead drive voltage phase angle thadv is still unknown, so step S23 and step S24' are respectively performed to calculate the two temporary lead drive voltage phase angles thadv丨, thadv2' and then calculate The lead drive voltage phase angle thadv, the step S23 is to first calculate the first temporary lead drive voltage phase angle thadw, as shown in the formula (2): thadv, , thadvra = (- (2);
以及計算第二個暫時超前驅動電壓相角thadV2，其數學公式如下： (3)；And calculating the second temporary lead drive voltage phase angle thadV2, the mathematical formula is as follows: (3);
thadv2 -i^adv_-thadVl)〇y-D,h) (D.nax-DJ 接著步驟S24,如公式(4)所示，再將兩個暫時超前驅動電壓相角ώα(1νι 與thadv2相加’即可得到超前驅動電塵相角thadv，其中公式(4)係如下所列： thadv = thadv, + thadv2 (Δ,. W ’ 回到第二圖示，不論是由步驟S22或者是由步驟S23與S24得到馬達 φ 11的超則驅動電壓相角，還都必須經過步驟S3與最大超前驅動電壓 相角thadvmax比較的判斷步驟；若在步驟S3，超前驅動電壓相肖祕乂大於 最大超刖驅動電壓相角thadVmax ’則選擇進入步驟S4，將超前驅動電壓相角 thadv限制為最大超前㈣電壓相角thadvmax，並接著進行步驟S5 ;反之， 若在步驟S3巾翁驅㈣齡肖thadv^、域雜最大麟驅動電壓相 角thadVinax，則由步驟％直接進入步驟％ ;再步驟％中根據先前棟取到 的轉子區域及轉子肖度^與轉S1至挪&計算制的超前驅動電壓相 角thadv之關係選擇功率開關控制信號，再經數位化反流器控制裝置16的 201108595 硬體介面電料出-_錢，轉觀絲12鶴凸極無刷直流馬達 1卜上述實紐餘料止是―個職結束，特下—個職關始時再 重返步驟S1 ’藉由此流程週社循環而達聰健動凸極無職流馬達的 效能。依此方法驅動之凸極無刷直流馬達特性例如第五圖所示，圖中表示 -凸極無刷直流馬達U之A相反電勢電壓〜、B相反電勢電壓〜浦子 位置偵測器14所輸出2轉子位置信號u，v波形相對位置。 在說明完本發明揭示之凸極無刷直流馬達驅動控制方法之後，更以一 實際實施麻代紐實驗結級證本發明之優點，係紐第—圖建立一凸 極無刷直流馬達速度驅練s。設定_四極凸極無刷直流馬達丨丨額定電愿 是48V’額定功率是800w，額定轉速45〇〇rev/mh，電壓源13是術〜55v， 數位化反顏控職置16採贿練錄號處㈣，實驗設定臨界工作週 期Dth=0.8，超前的驅動電壓相角thadv=3〇e。 根據前述餅進行實測，再請參照第六圖之實驗縣，顯示馬達u低 速3780rev/min且重載時，依據本發明之驅動方法實測之響應波形圖第六 圖中之信號u波形與八相反電勢電壓原為同相，而圖中所見A相電流波形 ~已超前信號U波形，3，®中同時顯示出電1源13的電流zDe平均值為 12.52A ’以及心的峰對峰值是13 8A ;為比較性能，再提供習用一驅動無 刷直流馬達方法之實驗結果進行比較，如第七圖習用驅動無刷直流馬達方 法之實驗結果所示，相似地顯示馬達在低速3720rev/min ’且與上述第六圖 實驗承受相同重載時之實測響應波形圖，第七圖中之信號u波形仍與a相 反電勢電壓為同相，而圖中所見A相電流波形L並無超前信號u波形，另， 圖中顯不電壓源13的電流z_dc平均值為13.85A，以及!DC的峰對峰值是2〇 〇 201108595 A;顯見第六圖所示電流Zdc平均值低於第七圖所示電‘平均值差有μ 大小，意即本發明方法較習用方法消耗更低功率，同時再比較第六圖所示 電流峰對峰值低於第七圖所示電流峰對峰值約有6 Α，意即本發明方法 •較習用方法將使馬達產生更小脈動轉矩。另外，第八圖係顯示採用本發明 方歧馬達運轉於最高轉速6480 rev/min時的實驗波形，在，之信號u 波形仍與A相反電勢電壓為同相，而圖中所見A相電流波形0顯超前信 號u波形，且圖中同時顯示電壓源13的電流^平均值約為2ia，以及^的 # ♦對峰值約是14 A。 由以上實驗絲更加峨證本發明方法的性能遠高於㈣鶴法的性 能。由上述3個實驗結果可知，透過本發明所揭示的驅動方法，在低速重 載時’較習財法有較小的/〇c平均值與峰對峰值，故驗證本方法不但能提 尚驅動器效率最大達到約6%，即減少約娜功率，並相當程度降低了馬 達的脈動轉矩’而同時保持在高速弱磁進行高效率運轉，故，以本發明之 驅動方法H動之·無職流財，其麟魏鮮理更為相近。 综上所述之實施例僅係為說明本發明之技術思想及特點，其目的在使 熟習此項技藝之人士能夠瞭解本發明之内容並據以實施，當不能以之限定 本發明之專利範圍’即大凡依本發明所揭示之精神所作之均等變化或修 飾’仍應涵蓋在本發明之專利範圍内。 【圖式簡單說明】 第一圖為用以實現本發明驅動方法之一系統示意圖❶ 第二圖為本發明使用之反流器結構示意圖。 第二圖為本發明驅動方法之方法流程圖。 201108595 第四圖為使用本發明計算超前驅動電壓相角之流程圖。 第五圖為獅本發魏動方法之凸極無職流馬達其A減電勢電壓、b 相反電勢電壓與轉子位置信號u，v波形相對位置實測波形圖。 第六圖為本發明-實賴，凸極無織流馬達驅動方法在低速(3徽ev/min) 高轉矩時產生相位超前電流實驗波形圖。 第七圖為習用無刷直流馬達驅動法在低速(3720rev/min)高轉矩時無法產生 相位超前電流實驗波形圖。 第八圖為本發明另一實施例，凸極無刷直流馬達驅動方法在高速 鲁 (6480rev/min)時能產生相位超前電流實驗波形圖。 第九圖為習知反流器6個功率開關切換模式以及理想A相反電勢電壓波 形之示意圖。 第十圖為習知反流器6個功率開關弱磁切換模式與理想A相反電勢電壓 波形之示意圖。 、 【主要元件符號說明】 100系統 0 11 凸極無刷直流馬達 12 反流器 13 電壓源 14 轉子位置偵測器 15負載 16 數位化反流器控制裝置 161轉子位置區域偵測單元 12 201108595 162 轉速計算單元 163 速度控制及電流限制單元 164控制信號單元Thadv2 -i^adv_-thadVl)〇yD,h) (D.nax-DJ Next, in step S24, as shown in equation (4), the two temporary lead drive voltage phase angles ώα (1νι and thadv2 are added together) The leading drive electric dust phase angle thadv can be obtained, where the formula (4) is as follows: thadv = thadv, + thadv2 (Δ, W ' Back to the second illustration, either by step S22 or by step S23 S24 obtains the super-drive voltage phase angle of the motor φ11, and must also go through the judging step of comparing step S3 with the maximum lead drive voltage phase angle thadvmax; if in step S3, the lead drive voltage phase is greater than the maximum super-turn drive voltage The phase angle thadVmax ' then proceeds to step S4 to limit the leading drive voltage phase angle thadv to the maximum leading (four) voltage phase angle thadvmax, and then proceeds to step S5; conversely, if in step S3, the towel is driven (four) age shadv^, domain miscellaneous The maximum lining drive voltage phase angle thadVinax, then step % directly enters step %; in step %, according to the rotor region and rotor oscillography obtained by the previous ridge ^ and the S1 to move & calculation system, the leading drive voltage phase angle thadv Relationship selection power switch control letter No., 201108595 hardware interface electrical material output by the digital inverter control device 16 - _ money, turn to the wire 12 crane salient brushless DC motor 1 Bu above the real surplus material is "the end of the job, special At the beginning of the next job, then return to step S1 'by the process of the weekly cycle of the company, and the performance of the smart and unobtrusive motor. The salient pole brushless DC motor driven by this method, for example, the fifth figure In the figure, the opposite potential voltage of the salient-pole brushless DC motor U, the opposite potential voltage of B, and the relative position of the two rotor position signals u, v of the output position of the sub-position detector 14 are shown. After the salient pole brushless DC motor drive control method, the advantage of the present invention is realized by an actual implementation of the Maide New Experiment junction level certificate, and the system is built to establish a salient pole brushless DC motor speed s s. setting _ quadrupole Salient pole brushless DC motor 丨丨 rated power is 48V' rated power is 800w, rated speed 45〇〇rev / mh, voltage source 13 is surgery ~ 55v, digital anti-yan control position 16 bribery practice record (D), the experiment sets the critical duty cycle Dth = 0.8, ahead of the drive The pressure phase angle thadv=3〇e. According to the above-mentioned cake, the actual measurement of the cake, please refer to the experimental county of the sixth figure, showing the motor u low speed 3780rev/min and heavy load, the response waveform diagram measured according to the driving method of the present invention is sixth. The signal u waveform in the figure is in phase with the eight opposite potential voltages, and the A phase current waveform seen in the figure has advanced the U waveform, and the current zDe average of the output source 13 is 12.52A. And the peak-to-peak value of the heart is 13 8A; for comparative performance, the experimental results of the conventional brushless DC motor method are compared, as shown in the experimental results of the seventh-pattern driving brushless DC motor method, similarly displayed The measured response waveform of the motor at the low speed of 3720 rev/min ' and the same heavy load as the experiment in the sixth figure above, the signal u waveform in the seventh figure is still in phase with the opposite potential voltage, and the A phase current waveform seen in the figure L has no leading signal u waveform. In addition, the current z_dc average of the voltage source 13 is 13.85A, and the peak-to-peak value of !DC is 2〇〇201108595 A. The current Zdc average shown in the sixth figure is shown. Lower than seventh The electric mean value difference shown in the figure has a μ size, which means that the method of the present invention consumes lower power than the conventional method, and at the same time, the current peak to peak value shown in the sixth figure is lower than the current peak to peak value shown in the seventh figure. 6 Α, meaning that the method of the invention • a conventional method will result in a smaller pulsating torque of the motor. In addition, the eighth figure shows the experimental waveform when the square motor is operated at the maximum speed of 6480 rev/min, and the signal u waveform is still in phase with the opposite potential voltage of A, and the phase A current waveform seen in the figure is 0. The signal u waveform is advanced, and the current value of the voltage source 13 is also shown to be about 2ia, and the #♦ of the ^ is about 14 A. From the above experimental yarns, it is further proved that the performance of the method of the present invention is much higher than that of (4) the performance of the crane method. From the above three experimental results, it can be seen that the driving method disclosed by the present invention has a smaller average value and a peak-to-peak value at the time of low speed and heavy load, so that the method can not only improve the driver. The efficiency is up to about 6%, that is, the power of the genus is reduced, and the pulsating torque of the motor is considerably reduced while maintaining the high-speed weak magnetic field for high-efficiency operation. Therefore, the driving method of the present invention is driven by the non-professional flow. Cai, its Lin Wei Xianli is more similar. The embodiments described above are merely illustrative of the technical spirit and characteristics of the present invention, and the purpose of the present invention is to enable those skilled in the art to understand the contents of the present invention and to implement the present invention. 'Equivalent variations or modifications made by the spirit of the invention as disclosed herein are still to be included in the scope of the invention. BRIEF DESCRIPTION OF THE DRAWINGS The first figure is a schematic diagram of a system for implementing the driving method of the present invention. The second drawing is a schematic diagram of the structure of the inverter used in the present invention. The second figure is a flow chart of the method of the driving method of the present invention. 201108595 The fourth figure is a flow chart for calculating the phase angle of the advanced driving voltage using the present invention. The fifth picture shows the measured waveform of the relative position of the A-reduction potential voltage, the b-side potential voltage and the rotor position signal u, v waveforms of the spurs of the lion. The sixth figure is the waveform diagram of the phase lead current generated by the salient-poleless motorless driving method at the low speed (3 ev/min) high torque. The seventh figure shows the experimental waveform of the phase lead current not generated by the conventional brushless DC motor drive method at low speed (3720 rev/min). The eighth figure is another embodiment of the present invention. The salient pole brushless DC motor driving method can generate a phase lead current experimental waveform diagram at a high speed Lu (6480 rev/min). The ninth figure is a schematic diagram of six power switch switching modes of the conventional inverter and an ideal A opposite potential voltage waveform. The tenth figure is a schematic diagram of the waveforms of the six power switch weak magnetic switching modes and the ideal A opposite potential voltages of the conventional inverter. [Main component symbol description] 100 system 0 11 salient pole brushless DC motor 12 inverter 13 voltage source 14 rotor position detector 15 load 16 digital inverter control device 161 rotor position region detecting unit 12 201108595 162 Speed calculation unit 163 speed control and current limiting unit 164 control signal unit