200838114 九、發明說明： 【發明所屬之技術領域】 ' 本發明係有關一種切換式(switching-mode)電源轉換器及脈寬 調變(pulse width modulation; PWM)控制電路，特別是關於一種應 用於一次側(primary-side)迴授控制的切換式電源轉換器及脈寬調 f 變控制電路。 【先前技術】 『第1圖』所示係習知的反馳式轉換器(flyback i 〇〇 ，來自交流電壓源101提供的交流電壓VAC經電磁干擾(Electro-Magnetic Interference; EMI)濾波器 102 及橋式 整流器 （bridge 、 reCtifier)104濾波及整流後得到具有漣波的直流輸入電壓％，輕合 至變壓器ΤΧ的一次侧繞組(primary-side winding)Lp，x力率開關sw 與變壓器TX的一次侧繞組Lp串聯，控制器1〇6輪出脈寬調變信 號Vpwm切換功率開關SW，以轉換輪入電壓Vi至變壓器τχ的 二次侧(secondary_side)產生輸出電壓Vo。電阻R3與變壓哭τχ的 一次侧繞組Lp串聯，用以偵測一次侧繞組電流Ιρ。為了調節 5 200838114 (regulate)輸出電壓v 籍 %也周整器11〇及光輕合器⑽— coupler)108 從變壓哭 τχ 的-Α η 白勺—次側取一與輪出電壓成正比的迴授 仏號Vfb給控制器1〇6，控制 106根據迴授信號Vfb以及一次 侧繞組電流Ip來控制脈寬調變 又乜唬VpWm的責任週期(duty cycle) ，因而穩定輪出電壓γ0。 反馳式轉換器 100可以提供精準的輸出電 壓調節，但需要二 次侧迴授相關的電路，控制系統的成本較高。 【發明内容】 本發明的目的在於提出—種制於—次側迴授控制的切換式 電源轉換觀絲靖控制電路，其具有低成本的優勢。 本發明之脈寬靖㈣，應祕—次_授控制的切換 式電源轉換ϋ，該切換式電轉換器包含—變_、—功率開關 、一電流感測(current sensing)電阻以及一脈寬調變控制電路。該變 t為具有-次側繞組、二次側繞組以及_輔助繞組，其中該辅助 繞組之電壓經由整流後輸出一直流的迴授電壓；與該一次側繞組 串4之該功率開關X該脈ϋ艘控制II電路產生之脈寬調變信號 200838114 次 切換，該電流感測電阻配置於該功率開關與接地之間，將該 侧繞組之電流轉換為一感測電壓。 本發明第一實施例的脈寬調變控制電路包含一取樣保抄 (sample andhold)!^ > (transconductor)f } 器根據該誤差信號決定 放大器以及-脈寬調變產生器。其中該取樣保持電路對該感測電 壓作祕取樣後’再將_電壓之I值雜在輪出端；該轉導電 路減至該取樣保持電路之輪出端，將感測電壓之秦值轉" 電流訊號；該誤差放大器具有二輸人端，其-輸人端_該電流 訊號及該迴授電壓’另—輪人端墟至―岐參考電屋， 放大產生-#差^§號；該脈寬調變產生 該脈寬調變信號的責任週期。 本發明第二實施例·寬調變控機 弟取樣保持 電路、一弟二取樣保持雷 、 叫、—運算處理單元、一轉導電路、一 誤呈放大器以及一脈寬調 斗抑#^ 、 中該第—取樣保持 該感測電壓作峯值取樣狳 、 再將_麵之峯值鱗在輪出端. 該第二取樣縣電路對 ’ •屋作初始值取樣後，再將感_ 7 200838114 i之初始值保持在輪出端；該運算處理單元將該感測電壓之桊值 減去該初始值，產生一感測電壓變化量；該轉導電路耦接至該運 算處理單元之輪出端，將該感測電壓變化量轉換為一電流訊號； 該誤差放大器具有二輪入端，其一輸入端耦接該電流訊號及該迴 授電壓，另一輪入端耦接至一固定參考電壓，該誤差放大器產生 一誤差信號；該脈寬調變產生器根據該誤差信號決定該脈寬調變 信號的責任週期。 【實施方式】 『第2(a)圖』顯示本發明第一實施例的切換式電源轉換器20 ，來自交流電壓源202提供的交流電壓VAC經EMI濾波器204及 橋式整流器206產生直流輸入電壓Vi，耦合至變壓器TX的一次 側繞組Lp，功率開關SW與變壓器TX的一次侧繞組Lp串聯， PWM產生器214產生脈寬調變信號Vpwm來切換功率開關SW， 將輸入電壓Vi轉換為輸出電壓V〇。 『第2(b)圖』顯示『第2(a)圖』中脈寬調變信號Vpwm、一次 侧繞組電流Ip及二次侧繞組電流Is在不連續導通模式 200838114 (Discontinuous Conduction Mode ; DCM)的波形圖。 在『第2(b)圖』中波形200b表示脈寬調變信號Vpwm，波形 202b表示一次侧繞組電流Ip，波形204b表示二次侧繞組電流Is 。在脈寬調變信號Vpwm的責任期間(on-dutyperiod)Ton，功率開 關SW導通，一次侧繞組電流ip從〇逐漸增加至秦值^peak value)IpK ，一次侧繞組Lp因此儲存能量200838114 IX. Description of the invention: [Technical field to which the invention pertains] The present invention relates to a switching-mode power converter and a pulse width modulation (PWM) control circuit, particularly to one application. A primary-side feedback controlled switching power converter and a pulse width modulated variable control circuit. [Prior Art] The first embodiment shows a conventional flyback converter (flyback i 〇〇, an alternating current voltage VAC supplied from an alternating voltage source 101 via an electromagnetic interference (EMI) filter 102 And the bridge rectifier (bridge, reCtifier) 104 filters and rectifies to obtain a chopper DC input voltage %, which is lightly coupled to the transformer's primary-side winding Lp, x force rate switch sw and transformer TX The primary side winding Lp is connected in series, and the controller 1〇6 turns out the pulse width modulation signal Vpwm to switch the power switch SW to convert the wheeling voltage Vi to the secondary side of the transformer τχ to generate an output voltage Vo. The resistor R3 and the transformer The first side winding Lp of the crying χ is connected in series to detect the primary winding current Ιρ. In order to adjust 5 200838114 (regulate) the output voltage v is also the peripheral 11 〇 and the optical coupling (10) — coupler) 108 from the transformer Crying τχ-Α η - The secondary side takes a feedback nickname Vfb proportional to the wheel-out voltage to the controller 1〇6, and the control 106 controls the pulse width modulation according to the feedback signal Vfb and the primary side winding current Ip. Change and 乜唬V The duty cycle of pWm, thus stabilizing the turn-off voltage γ0. The flyback converter 100 can provide accurate output voltage regulation, but requires secondary side feedback related circuits, and the cost of the control system is high. SUMMARY OF THE INVENTION An object of the present invention is to provide a switching power supply switching control circuit that is controlled by a secondary side feedback control, which has the advantage of low cost. The pulse width (four) of the present invention is a switched-type power converter that is controlled by a sub-control system, and the switching type electric converter includes a variable-current, a power switch, a current sensing resistor, and a pulse width. Modulation control circuit. The variable t is a secondary-side winding, a secondary-side winding, and an _ auxiliary winding, wherein a voltage of the auxiliary winding is outputted through a rectified output current; and the power switch X of the primary winding series 4 The pulse width modulation signal generated by the control circuit II is switched to 200838114 times. The current sensing resistor is disposed between the power switch and the ground, and converts the current of the side winding into a sensing voltage. The pulse width modulation control circuit of the first embodiment of the present invention includes a sample and hold device (^) (transconductor), which determines an amplifier and a pulse width modulation generator based on the error signal. Wherein the sampling and holding circuit secretly samples the sensing voltage, and then the I value of the voltage is mixed at the wheel end; the transducing circuit is reduced to the rounding end of the sampling and holding circuit, and the voltage value of the sensing voltage is Turn " current signal; the error amplifier has two input terminals, its - input terminal _ the current signal and the feedback voltage 'other - round people end to the 岐 reference electric house, amplification generated - #差^§ No.; the pulse width modulation produces a duty cycle of the pulse width modulation signal. The second embodiment of the present invention is a sample-and-hold circuit for a wide-band variable-control machine, a sample-and-hold sample, a lightning, a call, an operation processing unit, a transduction circuit, a mis-amplifier, and a pulse width adjustment control. The first sample-sampling maintains the sensing voltage for peak sampling, and then the peak of the _ surface is at the round end. The second sampling county circuit samples the initial value of the house, and then feels _ 7 200838114 i The initial value is maintained at the rounding end; the arithmetic processing unit subtracts the threshold value of the sensing voltage from the initial value to generate a sensing voltage change amount; the transducing circuit is coupled to the rounding end of the arithmetic processing unit Converting the sensed voltage change into a current signal; the error amplifier has a second wheel end, one input end coupled to the current signal and the feedback voltage, and the other wheel end coupled to a fixed reference voltage, The error amplifier generates an error signal; the pulse width modulation generator determines a duty cycle of the pulse width modulation signal according to the error signal. [Embodiment] FIG. 2(a) shows a switching power converter 20 according to a first embodiment of the present invention, and an AC voltage VAC supplied from an AC voltage source 202 generates a DC input via an EMI filter 204 and a bridge rectifier 206. The voltage Vi is coupled to the primary side winding Lp of the transformer TX, and the power switch SW is connected in series with the primary side winding Lp of the transformer TX. The PWM generator 214 generates a pulse width modulation signal Vpwm to switch the power switch SW to convert the input voltage Vi into an output. Voltage V〇. "2nd (b)" shows the pulse width modulation signal Vpwm, the primary winding current Ip, and the secondary winding current Is in the discontinuous conduction mode 200838114 (Discontinuous Conduction Mode; DCM) Waveform. In the "second (b) diagram", the waveform 200b indicates the pulse width modulation signal Vpwm, the waveform 202b indicates the primary side winding current Ip, and the waveform 204b indicates the secondary side winding current Is. During the duty period of the pulse width modulation signal Vpwm (on-dutyperiod) Ton, the power switch SW is turned on, the primary side winding current ip is gradually increased from 〇 to the peak value IpK, and the primary side winding Lp stores energy.
：LpxIpk-|lp(^xW VI」 Γ χι~χΤοηζ Lp [公式1] 其中VI為一次侧繞組Lp上的跨壓。在脈寬調變信號Vpwm的非 責任期間(off-dutyperiod)T〇ff，功率開關SW關閉，儲存在一次側 繞級Lp的能量傳遞到二次侧繞組Ls，因此二次側繞組電流^從 峰值^逐漸減少到〇。與二次側繞M連接的二極體m用來整流 ’其順向偏壓(f™i讀age)為Vf，R〇為二次側的輪出阻抗， 1〇為一次側繞組電流Is的平均值，路左仏山& 推~在輪出電容C。的電壓為輸 出龟壓Vo。在非責任期間T〇g·,變墨哭τ π〇ΤΧ的二次側消耗能量為:LpxIpk-|lp(^xW VI" Γ χι~χΤοηζ Lp [Formula 1] where VI is the voltage across the primary winding Lp. During the non-responsible period of the pulse width modulation signal Vpwm (off-dutyperiod) T〇ff The power switch SW is turned off, and the energy stored in the primary side winding stage Lp is transferred to the secondary side winding Ls, so that the secondary side winding current is gradually reduced from the peak value to the 〇. The diode connected to the secondary side winding M is m Used to rectify 'the forward bias (fTMi read age) is Vf, R 〇 is the secondary side of the wheel-out impedance, 1 〇 is the average value of the primary side winding current Is, the left-hand side of the mountain & push ~ The voltage of the output capacitor C is the output turtle pressure Vo. In the non-responsible period T〇g·, the energy consumption of the secondary side of the ink 哭τ π〇ΤΧ is
VoxIoxTof f+I〇2VoxIoxTof f+I〇2
X off [公式2] R〇xToff+I〇xYfx^ 200838114 貫際上一次側繞組Lp儲存的能量係傳遞給二次側繞組^及 一次側的輔助繞組Laux，但是辅助繞組Laux消耗的能量非常小 ，故可以忽略不計。公式1等於公式2，均除以可得 V〇xl〇+I〇2 xR〇+i〇xVf」XZLX Ton2 2 Lp Toff [公式3] 又因為二次側繞組Ls上的跨壓 V2=V o+I〇xR〇+Vf [公式4] 公式3除以1〇，代入公式4可得 V2=Vo+I〇xR〇+Vf-ixIf vTon2 1 2 LpX^fX[ [公式 5] 由以：> 可知’ §一次側平均電流1〇因為負載增加而增加時，二 人侧、、几組Ls上的％壓V2及輸出電壓v。都將下降，反之則上升。 辅助繞社祖在變壓器TX的一次侧，其繞組電流Iaux經二 極體D2整流’對電容C1充電產生直流之迴授電壓vfb，輔助繞 組Laux上的跨壓 [公式6] V3-Vf2+Vfb 其中’電壓Vf2為二極體D2的順向偏壓。假設繞組Lp、Ls及 10 2〇〇838li4 的阻數比為nl : n2 ·· n3，則二次侧繞組電壓 • V2==(n2/n3)xV3=NxV3 [公式 7] 其中，N為二次侧繞組Ls對輔助繞組Laux的匝數比。 電阻R1及R2串聯，可視為電阻分壓器，跨在電阻R2的電 壓 Vfbi，vfbl=vfbx[R2/(R1+R2)] 電流感測電阻Res與一次侧繞組Lp串聯，可將一次侧繞組電 流Ip轉換為感測電壓Vcs。 脈見調變控制電路200包含一取樣保持電路220、一轉導電路 -40、一誤差放大器212以及一脈寬調變產生器ΜΑ。取樣保持電 路220包含M〇s電晶體221及224、電容222及您、一缓衝器 (buffer)223 以及一反相器 226。 在脈覓調變信號Vpwm為邏輯高準位時(如『第2(b)圖』之To幻 ’由於電晶體221導通，感測電壓％傳遞至電容222充電，當 脈寬調變信號Vpwm轉為邏輯低準位時(如『第2(b)圖』之T〇ff) ，電晶體221酬，使得電容222白勺取樣電壓為一次侧繞組電流 的峯值IPK (如『第2(b)圖』所示)乘上電流感測電阻〜，即感測 11 200838114 電壓vcs之峯值VpK，此時 版一24 ¥通，充電至電容222之感 測電壓峯值乂视經由緩衝器223傳 寻邈至毛谷22:>亚保持住，感測 電壓峯值VPK經由一電壓轅拖雷、六 胬換包概的轉導電路240後，可得到一 電流lx， [公式8] ^ = VPK/Rx 誤差放大器2U具有二輸入端，其反相輸入端輕接該感測電 流lx以及電隨與R2的串接點，該串接點观正比於迴 授電麼vfb;誤差放大器212的非反相輪入端雛至—固定參考電 壓 Vref。 由『第2⑻圖』可看出，流經電阻R1之電流n為流經電阻 幻之電流12及流經電阻以之電流ίχ之總和，即 Ι1=Ι2+Ιχ [公式 9] 迴杈電壓Vfb扣除電流流經電阻R1的壓降即為誤差放大器 312的反相輪入端電壓Vfbl，X off [Formula 2] R〇xToff+I〇xYfx^ 200838114 The energy stored in the last primary winding Lp is transmitted to the secondary winding ^ and the auxiliary winding Laux on the primary side, but the auxiliary winding Laux consumes very little energy. Therefore, it can be ignored. Equation 1 is equal to Equation 2, and is divided by V 〇 xl 〇 + I 〇 2 x R 〇 + i 〇 x Vf "XZLX Ton2 2 Lp Toff [Equation 3] and because of the cross-pressure V2 = V o on the secondary winding Ls +I〇xR〇+Vf [Formula 4] Divide Equation 3 by 1〇 and substitute Equation 4 to get V2=Vo+I〇xR〇+Vf-ixIf vTon2 1 2 LpX^fX[ [Formula 5] by: > ; It can be seen that § The average current of the primary side is increased by 1 〇 because of the increase in load, the two-person side, the % voltage V2 on the Ls and the output voltage v. Both will fall, and vice versa. On the primary side of the transformer TX, the winding current Iaux is rectified by the diode D2' to charge the capacitor C1 to generate the DC feedback voltage vfb, and the voltage across the auxiliary winding Laux [Equation 6] V3-Vf2+Vfb Where 'voltage Vf2 is the forward bias of diode D2. Assuming that the resistance ratios of the windings Lp, Ls and 10 2〇〇838li4 are nl : n2 ·· n3, the secondary winding voltage • V2==(n2/n3)xV3=NxV3 [Equation 7] where N is two The turns ratio of the secondary winding Ls to the auxiliary winding Laux. The resistors R1 and R2 are connected in series and can be regarded as a resistor divider. The voltage across the resistor R2 is Vfbi, vfbl=vfbx[R2/(R1+R2)]. The current sense resistor Res is connected in series with the primary winding Lp to enable the primary winding. The current Ip is converted into a sensing voltage Vcs. The pulse modulation control circuit 200 includes a sample and hold circuit 220, a transducing circuit -40, an error amplifier 212, and a pulse width modulation generator. The sample and hold circuit 220 includes M〇s transistors 221 and 224, a capacitor 222 and you, a buffer 223, and an inverter 226. When the pulse modulation signal Vpwm is at a logic high level (such as "To 2" in the 2nd (b) diagram), since the transistor 221 is turned on, the sense voltage is transmitted to the capacitor 222 for charging, and when the pulse width modulation signal is Vpwm When turning to logic low level (such as T〇ff in Figure 2(b)), the transistor 221 is paid so that the sampling voltage of the capacitor 222 is the peak IPK of the primary winding current (eg, 2nd (b) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) ) )邈到毛谷22:> sub-hold, the sense voltage peak VPK can be obtained by a voltage 辕 、 , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , , The Rx error amplifier 2U has two input terminals, and the inverting input terminal is connected to the sensing current lx and the series connection point of the electric and R2. The series connection point is proportional to the feedback power vfb; the non-reverse of the error amplifier 212 The phase wheel enters the end to the fixed reference voltage Vref. It can be seen from the 2nd (8) diagram that the current n flowing through the resistor R1 is the current flowing through the resistor. 12 and the sum of the currents flowing through the resistors, that is, Ι1=Ι2+Ιχ [Equation 9] The voltage drop of the return voltage Vfb minus the current flowing through the resistor R1 is the inverted wheel terminal voltage Vfbl of the error amplifier 312,
Vfb—Il*Ri：=vfbl [公式 10] 由於整個迴授系統穩定後，誤差放大器212的反相輸入端電 12 200838114 壓VfM將等於誤差放大器212的非反相輪入端電壓w，即 Vfbl = Vref [公式 1η 將公式11及公式9代入公式10，又因12 =勒體，可得到 Vfb = \/^产[1+(111船)]+ ix*Rl 公式 η 假設二極體m及D2具有相同的順向偏壓，即vf=vQ，根 據公式4、公式6及公式7可得 :VfbxN+(N-l)Vf_I〇xR〇 [公式13] 由一八可推知，當一二又侧平均電流因為負載增加而突增 加時，輸出錢Vo將下降；由公式5可知，當二次側平均電流& 囚為負载增加而突增加時，二次側繞組Ls上的跨壓％將下降， 气6及&式7可推導出迴授電墨.也隨之下降，再由公式 0可推導出誤魏大器犯的反相輸入端電壓观也隨之下降， 使件决差放大$212的輪出端之誤差信號％有誤差值綠 產生，脈寬調變產生器214偵測該誤差值綠後，則提高該脈寬 ’信號外麵的責任週期，此時-次側繞組Lp上的電流Ip也 感測vGSH^升高，電流Ιχ也升高，由公式η可得知 13 200838114Vfb_Il*Ri:=vfbl [Equation 10] Since the entire feedback system is stabilized, the inverting input terminal 12 of the error amplifier 212 200838114 voltage VfM will be equal to the non-inverting wheel terminal voltage w of the error amplifier 212, that is, Vfbl = Vref [Formula 1η Substitutes Equation 11 and Equation 9 into Equation 10, and because of 12 = Lex, Vfb = \/^ can be produced [1+(111 ships)]+ ix*Rl Formula η Hypothetical diode m and D2 has the same forward bias, ie vf=vQ, according to formula 4, formula 6 and formula 7: VfbxN+(Nl)Vf_I〇xR〇[Equation 13] can be inferred from one to eight When the current suddenly increases due to the increase of the load, the output money Vo will decrease; from Equation 5, when the secondary side average current & Prison increases the load, the cross-pressure % on the secondary winding Ls will decrease. Gas 6 and & 7 can be used to derive the feedback ink. It also decreases. From Equation 0, it can be deduced that the voltage of the inverting input terminal of the false Wei instrument is also reduced, so that the wheel of the difference is enlarged by $212. The error signal % at the out end has an error value of green, and after the pulse width modulation generator 214 detects the error value, the pulse width is increased. Duty cycle outside, at this time - the current Ip on the primary winding Lp also sense vGSH ^ increases, current Ιχ also increased, can be learned by the formula η 13200838114
Vfb也升高。 再由公式13可推導出輸出電壓v〇也將隨娜升高而上升， 可以抵消S二次側平均電流！。增加所造成的下降。 『第3⑻圖』顯示本發明第二實施例的切換式電源轉換器％ ，『第3(b)圖』顯示『第3⑷圖』中脈寬調變信號VpWm、一次侧 繞組電流及二摘繞組f流Is棘料雜式(C⑽ Conduct· Mode ; CCM)的波形圖。在『第吻圖』中波形遍 表示脈寬調變信號Vp職，波形3〇2b表示_次側缓组電·，波 形304b表示二次侧繞組電流Is。在脈寬調變信號外魏的責任期 間^，功率開關SW導通，一次側繞組電流Ip從初始值&逐漸 增加至峯值IPK，在脈寬勒讓Vpwm的非責任_ .功率 開關SW賴，儲存在—次侧繞組Lp的能量傳遞到二次側繞组Vfb also rises. From Equation 13, it can be deduced that the output voltage v〇 will also rise with the increase of Na, which can offset the average current of the secondary side of S! . Increase the resulting decline. "3rd (8)" shows a switching power converter % according to a second embodiment of the present invention, and "3 (b) diagram" shows a pulse width modulation signal VpWm, a primary winding current, and a second winding in "3(4)" Waveform of f-flow Is spurs (C(10) Conduct·Mode; CCM). In the "Kin kiss diagram", the waveform traverse indicates the pulse width modulation signal Vp, the waveform 3 〇 2b indicates the _ secondary side slow group power, and the waveform 304b indicates the secondary side winding current Is. During the duty period of the pulse width modulation signal, the power switch SW is turned on, and the primary side winding current Ip gradually increases from the initial value & to the peak IPK, and the pulse width allows Vpwm to be non-responsible. The energy stored in the secondary winding Lp is transferred to the secondary winding
Ls，因此二次侧繞組電流Is從峯值Isk逐漸減少到。 ”本lx㈣f 婦彳相比較’第二實施例的脈寬調變控制 電路增加-取樣保持電路33Q及―運算處理單元挪取樣保 持電路330與樣保持電路220相比較，取樣保持電路330增加— 14 200838114 單發(⑽㈣脈衝產生電路331;在該脈寬調變信號Vp赠由邏 輯低準位轉為邏輯高準位時，脈衝產生器別製造—短的脈衝， 取騎$路；3:0细魏脈衝對感戦壓V。作初始值取樣後 :感、!U[之初始值vp。鱗在輸出端，此感測電壓初始值Vp〇 正比於『第聊』的初始電流Ip0;取樣保持電路220與本發明 的第一實施例一樣，在輪出端輪出感測電壓VCS之峯值VpK;運算 處理單心50將輸_之峯值Vp谢軸始值〜，產生一 感測電壓變化量；轉導電路_ _ 杜罘一A %例中耦接至該運算處 理單元说之輸出端，將該感測電壓變化量轉換為電心。’ 在本發明的第二實施 .連續導通模式時，更能達到精確 守^感洌電壓變化量轉換為電流lx ，對於切換式電源轉換器操作在 控制的效果。 雖然本發明以前述之實施例揭露如上， 明。在不脫離本發明 χ θ之精神和範圍内， 本發明之專娜護範圍。關 附之申請專利範圍。 然其並非用以限定本發 所為之更動與潤飾，均屬 方、本發明所界定之_範圍請參考所 15 200838114 【圖式簡單說明】 第1圖係習知的反馳式轉換器； 第2(a)圖顯示本發明的第一實施例的切換式電源轉換器； 第2(b)圖顯示第2⑻圖中脈寬調變信號、—次側繞組電流及二 次側繞組電流的波形圖； 第3(a)圖顯示本發明的第二實施例 號、一次侧繞組電流及， 切換式電源轉換器；及 第3(b)圖顯示第3(a)圖中脈寬調變作號 次侧繞組電流的波形圖。 【主要元件符號說明】 100 反馳轉換器 101 交流電壓源 102 EMI濾波器 104 橋式整流器 106 控制器 108 光耦合器 110 穩壓調整器 16 200838114 200b 脈寬調變信號的波形 202b 一次侧繞組電流的波形 204b 二次侧繞組電流的波形 20 切換式電源轉換器 200 脈寬調變控制電路 202 交流電壓源 204 EMI濾波器 206 橋式整流器 212 誤差放大器 214 產生器 220 取樣保持電路 221、 224 電晶體 222 \ 225 電容 223 緩衝器 240 轉導電路 30 切換式電源轉換器 17 200838114 300 脈寬調變控制電路 300b 脈寬調變信號的波形 302b 一次侧繞組電流的波形 304b 二次侧繞組電流的波形 330 取樣保持電路 33! 單發脈衝產生電路 350 運算處理單元 18Ls, therefore the secondary winding current Is is gradually reduced from the peak Isk. Compared with the sample-and-hold circuit 33Q and the operation processing unit, the sample-and-hold circuit 330 is compared with the sample-and-hold circuit 220, and the sample-and-hold circuit 330 is increased as compared with the sample-and-hold circuit 33 of the second embodiment. 200838114 Single-shot ((10) (four) pulse generation circuit 331; when the pulse width modulation signal Vp is converted from the logic low level to the logic high level, the pulse generator is not manufactured - short pulse, take the $ road; 3:0 After the initial value is sampled: the sense, !U[the initial value vp. The scale is at the output, the initial value of the sensed voltage Vp〇 is proportional to the initial current Ip0 of the "chat"; sampling The holding circuit 220 rotates the peak value VpK of the sensing voltage VCS at the wheel end as in the first embodiment of the present invention; the arithmetic processing unit 50 outputs a peak value of the peak value Vp of the input voltage to generate a sensing voltage change. The transducing circuit _ _ 罘 A A A A A A A A A A A A A A A A A A A A A A A A A A A A A A A A A A A A A A A A A A A A When it is more accurate, the amount of voltage change is converted to Flow lx, the effect of the operation of the switching power converter operation. Although the present invention has been disclosed in the foregoing embodiments, the scope of the present invention is not limited to the spirit and scope of the present invention. Included in the scope of patent application. However, it is not intended to limit the changes and refinements of this publication. All of them are defined by the scope of the invention. Please refer to the section 15 200838114 [Simple description of the diagram] Figure 1 is a counter-intuitive 2(a) shows a switched-mode power converter according to a first embodiment of the present invention; FIG. 2(b) shows a pulse width modulation signal, a secondary side winding current and a second in the second (8) diagram Waveform diagram of the secondary winding current; Fig. 3(a) shows the second embodiment of the present invention, the primary winding current and the switching power converter; and Fig. 3(b) shows the 3(a) diagram The waveform of the mid-pulse width modulation is changed to the waveform of the secondary winding current. [Main component symbol description] 100 Reverse converter 101 AC voltage source 102 EMI filter 104 Bridge rectifier 106 Controller 108 Photocoupler 110 Regulator regulator 16 200838114 200b Pulse width Waveform 202b of variable signal Waveform 204b of primary side winding current Waveform of secondary side winding current 20 Switched power converter 200 Pulse width modulation control circuit 202 AC voltage source 204 EMI filter 206 Bridge rectifier 212 Error amplifier 214 Generator 220 sample and hold circuit 221, 224 transistor 222 \ 225 capacitor 223 buffer 240 transfer circuit 30 switching power converter 17 200838114 300 pulse width modulation control circuit 300b pulse width modulation signal waveform 302b primary side winding current waveform 304b Secondary side winding current waveform 330 Sample hold circuit 33! Single shot generation circuit 350 Operation processing unit 18