SU1749843A2 - Device for measuring average rate of frequency and linearity of modulation characteristics of frequency-modulated generators - Google Patents

Abstract

The invention relates to a radio metering technique and can be used to measure the rate of change of frequency and linearity, the modulation characteristics of frequency modulated generators. The purpose of the invention is to extend the functionality by measuring the parameters of linear-frequency-modulated (chirp) signals of short duration. The purpose of the invention is achieved by extending the length of the input chirp signal using a dispersion delay line 23. For this purpose, a device for measuring the average rate of change of the frequency of frequency-modulated oscillators containing mixer 2. phase shifters 1 and 7,

Description

2

Yu

00 4 S

hch

counter 6, intermediate frequency amplifier 3, delay lines 14 and 24, controlled oscillator 13, frequency detector 8. summing amplifier 9, amplitude detector 25. phase detectors 17 and 18, shaper 21 pulses, strobe-cascade 10, memory block 11 these frequencies, phase shifter on nil 15, subtracting

block 22, video amplifier 28, integrators 14 and 27, electron-counting frequency meter 16 and oscilloscope 26, additionally introduced delay line 23 delay mixers 5 and 29. tunable oscillator 20, band-pass filters 12 and 30, limit amplifiers 19 and 31 and switches 32 - 35. 9 Il.

The invention relates to a radio metering technique and can be used to measure the speed of measuring the frequency and linearity of the modulation characteristics of frequency-modulated oscillators and is an improvement of the invention. No. 1499259,

The aim of the invention is to extend the functionality by measuring the rate of change of frequency and its separation from the linear law of linear frequency-modulated (chirp) signals of short duration.

FIG. 1 shows a block diagram of the device; Fig 2 is the same as the frequency memory block; in fig. 3 - the same, bandpass filter block; in fig. 4-9 plots showing the operation of the device.

The device contains (Fig. 1) phase shifter I, mixer 2, amplifier 3 with intermediate frequency, delay line 4. mixer 5, counter 6, phase shifter 7, frequency detector 8, summing amplifier 9. strobe stage 10, frequency memory block 11, band-pass filter, 12, controlled oscillator 13, integrator 14, phase shifter on nil 15, electronic counting frequency meter 16, phase detector 17, phase detector t8, limiter amplifier 19, tunable oscillator 20, pulse generator 21, subtractive unit 22, dispersion delay line DLL, 23 delay line 24, amplitude detector 25, oscilloscope 26. integrator 27 , video amplifier 28, mixer 29, block 30 bandpass filters, power limiter 31, switches 32-35.

At the same time, the device is connected in series to the second mixer 5, the first band-pass filter 12, the first switch 32, the first amplifier-limiter 19, the delay delay line 23, the second switch 33, the third mixer 29, the band-pass filter unit 30, the third switch 34, the second limiting amplifier 31, fourth switch 35, second delay line 24, first phase shifter 1, first mixer 2, intermediate frequency amplifier 3, strobe stage 10, frequency memory block 11, natg / 2 phase shifter 15, second phase detector 18, subtracting bl to 22, video amplifier 28, a first integrator 27, the oscilloscope 26. The second inputs of the second 5 and the third mixer 29 connected to the output of the tunable oscillator 20. A first input of the second mixer 5 is connected to the second inputs of the first 32 and fourth switch 35 and is an input device. The second output of the second switch 33 is connected to the second input of the third switch 34. The input of the delay line 24 is connected to the inputs of the second phase shifter 7 and the amplitude detector 25. The amplitude detector 25 and the driver 21 of the pulses are connected in series, the output of which is connected to the synchronized inputs of the strobe stage 10 and the oscilloscope 26

Frequency detector 8, summing amplifier 9, second integrator 14, controlled oscillator 13, counter 6, the first and second outputs of which are connected to the second inputs of the first 1 and second 7 phase shifters, the output of which is connected to the second input of the first mixer 2, are connected in series. generator 13 is connected to the input of the electron-counting frequency meter 16, the second inputs of the phase detectors 17 and 18 through the first delay line connected to the output of the intermediate frequency amplifier 3 and the input of the detector 8. The first input of the phase detector Pa 17 is connected to the output of frequency memory unit 11, and the output is connected to the second inputs of summing amplifier 9 and subtractive unit 22. Frequency memory unit 11 contains

an adder 36, a delay line 37 and a synchronized generator 38 (FIG. 2).

In block 11, the frequency memory is connected to a sequentially synchronized generator 38, a third delay line 37 and an adder 36, the output of the adder 36 is connected to the input of the synchronized generator 38, the input of the adder 36 is the input of the frequency memory block, and the output of the synchronized generator 38 is frequency memory output.

Block 30 of bandpass filters contains N-bandpass filters 39 and switches 40 N-inputs (Fig. 3). The inputs of the N-bandpass filters are interconnected and are the input of the bandpass filter unit 30, and the N-outputs of the bandpass filters 39 are connected to the N-inputs of the switch 40, the output of which is the output of the band filter unit 30.

The device works as follows.

Depending on the position of the switches 32-35, the device may operate in one of three measurement modes. Measurement mode for chirp signals of more than a few microseconds. The measurement mode of the chirp signals with a duration equal to or less than a few microseconds, having a frequency spectrum and bandwidth of the dispersion delay line 23, with the sign of the rate of frequency modulation of the chirp signal coincides with the slope of the dispersion characteristic of the SLD. The measurement mode of chirp signals with a duration equal to or less than a few microseconds with a frequency spectrum outside the passband of the DLS, or the sign of the frequency modulation of the chirp signal that does not coincide with the slope of the dispersion characteristic of the DLS.

In the measurement mode of chirp signals with a duration of more than a few microseconds, switch 35 connects the input of delay line 24 to the input bus of the device. In this mode, the device operates as follows. The measured chirp signal is applied via switch 35 to the input of delay line 24. The controlled oscillator 13 generates clock pulses with a following frequency, 5 RSpd n, where Psdv is the signal frequency offset, and n is the number of approximation steps of the sawtooth phase modulation. RSdv is chosen so that it is significantly more than the estimated deviations of the instantaneous frequency of the studied chirp signals from the linear law of the frequency change over time and is 15-20

MHz. When using known discrete phase modulators with a number of control digital code bits, the number of approximation stages can be 5, where n is a bit binary counter of 6 clock pulses, working with overflow reset, counts the clock pulses of the controlled oscillator 13. Output

0 direct and inverse codes of the binary counter 6 are supplied as control, respectively, to discrete phase shifters 1 and 7. To the high-frequency inputs of discrete phase shifters 1 and 7 are fed

5, respectively, delayed on line 24 of the delay and undelayed research of the chirp signal. As a result of stepwise sawtooth phase modulation of the chirp signal on discrete phase shifters 1 and 7

0 there is a shift of the delayed and not delayed investigated signals in frequency, respectively, up and down by the value of Pcm / 2. The use of discrete phase shifters 1 and 7 allows to reduce

5 the effect of their inertia on the measurement accuracy due to a two-fold decrease in the repetition rate of the control code.

Separated in frequency by the amount of FCM delayed and undelayed chirp

0 are multiplied by mixer 2. At the output of mixer 2, combination frequency components are formed, from which the difference frequency component is separated by amplifier of Frequency Frequency 3 and through delay line 4 is fed to inputs of phase detector 17 and phase detector 18. Reference oscillation for phase detector 18 is generated by a synchronized generator 38 of block 11

0 memory frequency, storing the frequency and the initial phase of the signal received at its input from the output of the amplifier 3 intermediate frequency through the strobe cascade 10 and the adder 36 during the enabling pulse with

5 of the pulse former 21, which opens the strobe cascade 10. The duration of the enabling pulse is chosen such that during this time in block 11 of the frequency memory a steady state is established.

0 Maintaining a constant frequency of the synchronized generator 38 for the duration of the chirp of the radio pulse is carried out using a self-synchronization loop, and the line 37 is included in the pattern.

5 delays and an adder 36. In order not to lose information about the measured parameters at the beginning of the pulse, the signal from the output of the amplifier 3 of the intermediate frequency is fed to the second input of the phase detector 17 through a line of 4 delays, the delay time of which

equal to the total transient time in the mixer 2 and the synchronized generator 38. The signal from the frequency memory block 11 is fed to the second input of the phase detector 17. The output voltage of the phase detector 17 is proportional to the instantaneous phase difference of the oscillations of the synchronized generator 38 and the signal from the output of the delay line 4.

Due to the inertia of the phase detector 17, the pulse voltage at its output has spikes at the beginning and end of the pulses (Fig. 4a). These transients, resulting in loss of information, as well as distortion of the signal at the output of phase detector 17, are compensated at subtractive unit 22. To do this, parallel to phase detector 17, a similar phase detector 18 is turned on, to which the reference signal from frequency memory unit 11 is supplied through the phase shifter 15 per l / 2, the operating point of the characteristic of the phase shifter 15 from position I moves to position II (Fig. 5). In this case, the phase detector 18 becomes insensitive to the phase difference of the input and reference signal, and its output signal characterizes the amplitude distortion of the input signal and its own transients. This signal (Fig. 46) is subtracted at the subtractive unit 22 from the output signal of the phase detector 17. The differential signal (Fig. 4c), which characterizes the frequency distortion of the analyzed chirp signal, through the video amplifier 28, the integrator 27 is fed to the observation and measurement on the oscilloscope 26. The oscilloscope 26 and the strobe cascade 10 are synchronized by pulses from a generator of 21 pulses triggered by the leading edge of the envelope of the analyzed chirped radio pulse selected by the amplitude detector 25.

Since the frequency of the synchronized generator 38 can vary within limited limits and in the steady state only take discrete values, and the difference frequency at the output of mixer 2 at a wide range of rates of change in the frequency of the analyzed chirp signals varies widely, a difference occurs in different frequencies the inputs of the phase detector 17. This leads to errors in the measurement of both the frequency measurement rate and the linearity of the modulation characteristics of frequency modulated (FM) oscillators.

Changing the difference frequency and phase of the signal at the output of the mixer 2 when analyzing chirp signals with different rates of change of frequency is compensated by the inertial scheme of the frequency-phase auto-tuning of the controlled oscillator 13 formed by the frequency detector 8, summing amplifier 9 and integrator 14. At the same time, the change in the frequency of the signal by The output of the intermediate frequency amplifier 3 causes a change in the pulse frequency of the controlled oscillator 13 fn. This will change the repetition rate of the control code by

the output of counter 6 is equal to the frequency of the chirp signal offset at each of the discrete phase shifters 1 and 7 I RSdV11: IFcAB2l 0.5FcAB fk, and the difference in the difference frequency RSdv is also compensated by

therefore, the difference in the frequencies of the signals at different inputs of the phase detector 17 that occurs when the rate of change of the frequency of the analyzed chirp signal changes. At the same time, the compensation of the difference between the oscillation frequencies at the various inputs of the phase detector 17 increases when the signals from frequency detector 8 are added to the summing amplifier 9 and phase detector

17, which ensures the frequency-phase auto-tuning of the frequency of the controlled oscillator 13. The frequency of the controlled oscillator 13 is proportional to the rate of change of the frequency of the analyzed chirp signal, which can be determined by measuring the electron-counting frequency meter 16.

In the measurement mode chirp signals of short duration, having a frequency spectrum in

the bandwidth of the dispersion delay line 23 and the coincidence sign of the frequency modulation frequency of the chirp signal with the slope of the dispersion characteristic of line 23, the switch 32 is transferred to the second position and connects the device input to the input of the first limiting amplifier, the switches 33 and 34 are switched to the second position and connect the output of the dispersed delay line 23 to the input of the second

the amplifier limiter 31, the switch 35 connects the output of the second amplifier limiter 31 to the inputs of the second delay line 24. In this mode, the device operates as follows.

The measured chirp signal of short duration through the switch 32 is fed to the input of the first amplifier-limiter 19 (Fig. BA). The bandwidth of the amplifier 19 is chosen equal to the bandwidth of the dispersed delay line 23. Strengthened and limited in amplitude in order to eliminate the influence of amplitude modulation, the chirp signal from the output of amplifier 19 is fed to the input of DLS 23 (Fig.

66). Since the sign of the speed of the frequency modulation of the LFM signal and the slope of the dispersion characteristic of the DLS 23 coincide (FIG. 7), at the output of the DLC 23 there appears a chirp signal with a frequency deviation equal to the deviation of the input chirp signal, but longer than the TC the input chirp signal (Fig. 6c) and delayed by time to be determined by the design features of the specific SLS and the initial frequency of the chirp signal (Fig. 7) .f

The delay time to is determined by the time to (the delay to inherent in a particular SLD, and determines the delay of the LFM response of the SLF to the influence of the delta function 5 (t) (Fig. 8) and the delay of the input chirp signal due to the dispersion of the LFD, which depends on the initial frequency fH chirp signal (Fig. 7):

to to + t / yi3 (fn),

where to is a constant delay for all frequencies determined by the LLL design; ) is the delay at the frequency fH determined by the dispersion characteristic of the SL (Fig. 7).

In this case, using FIG. 7 to determine Gvyh:

Afc

Your TS + GDLZ Tc + TDLZ,

where Afc is the frequency deviation of the input chirp signal; tdlz - DLZ 23 bandwidth; tdlz duration of the output chirp response DLZ 23 when applying to its input (t) - pulse (Fig. 8). Thus, the chirp signal at the output of the DLS 23 is increased in duration by the value of tdlz (Afc / A f for dz) tdlz

Further, the chirp extended for the duration of the chirp through the switches 33 and 34 is fed to the input of the second amplifier 31, which serves to amplify the chirped DLZ 23 chirped radio pulse and amplitude limiting in order to eliminate the influence of parasitic amplitude modulation (introduced due to non-uniformity amplitude-frequency characteristic DLZ 23) on the measurement results and thereby increase the measurement accuracy. The amplified and amplified chirped radio pulse (Fig. 6d) through the switch 35 enters the inputs of the second delay line 24, the second phase generator 7 and the amplitude detector 25, In the rest, the operation in this mode is similar to the device in measuring the parameters of chirp signals of longer duration a few microseconds.

Let us determine the average rate of the change in the frequency of the chirp signal at the output of the DLS 23. The average rate of the change in the frequency of the input chirp signal is determined by the formula

y - afc

YC TS

similarly

R

where Udlz - the slope of the dispersion characteristics of DLZ 23 (Fig. 7) dotted, from Fig. 7 follows Y fBWx Afc, therefore:

 f afc

Tdlz + Tc

one

 tdlz 4-ts

one . one

UDLZ Us

OR

from here

Us

UDLZ ALARM UDLZ - ALARM

where OO is the measured value s with a scatter measurement 16 for measuring the frequency of the chirp signal (Phys. 7).

Thus, the duration of the analyzed chirp signal after DL 23 becomes sufficient for normal operation of the frequency memory block 11, the influence of the inertia of phase shifters 1 and 7 on the measurement accuracy is reduced by decreasing the frequency of chromic frequency modulation at the output of the DRL 23. Moreover, The loss of information due to the delay in line 24 is reduced. Consequently, the introduction of DL 23 allows one to measure short-pulse chirp signals with a large frequency deviation.

In the measurement mode of short-duration chirp signals with a frequency spectrum outside the passband of the DLS 23, and also if the sign of the frequency modulation of the analyzed chirp signal does not match the slope of the dispersion characteristic of the DLS 23, switch 32 connects the output of the band-pass filter 12 to the input the amplifier limiter 19, the switch 33 connects the output of the DLZ 23 to the input of the mixer 29, the switch

34 is switched to the first position and connects the output of the bandpass filter block 30 to the input of the amplifier-limiter 31, the position of the switch

35 remains the same as in the previous case.

In this mode, the device operates as follows. The input chirp signal analyzed is input to the mixer 5, to the second input of which a continuous monochromatic signal fr is applied from the tunable generator 20. The generator frequency 20 fr is set so that the difference frequency fc-fr at the output of the mixer 5 if the sign of the frequency modulation rate coincides The chirp signal and the slope of the dispersion characteristic of the DLS 23, was within the passband of the DRL 23 (Fig. 9a). Further, the transformed chirp signal with the center frequency fc-fr is fed to a band-pass filter 12, the passband of which is chosen equal to the passband of the SLR 23.

A lazy filter 12 serves to suppress the conversion byproducts at the output of mixer 5, which affect the phase structure of the chirp signal at fc-fr due to the amplitude-phase conversion in amplifier 19, as they generate combinational components that are included in the chrm spectrum at frequency fc -fr. In the case of a mismatch of the sign of the frequency modulation rate of the input chirp signal and the slope of the dispersion characteristic of the DLS 23, the frequency of the generator 20 is set so that the frequency fr-fc is separated at the output of the band-pass filter 12.

Thus, in mixer 5, the spectrum is inverted, which means that the frequency of the frequency modulation of the chirp signal is reversed to the opposite (Fig. 96) and coincides with the slope of the dispersion characteristic of the DLC 23. The chirp signal converted by frequency through switch 32 and limiting amplifier 19 enters at the entrance DLZ 23, where the expansion of its duration. The extended chirp signal with the center frequency fc-fr or fr-fc through the switch 33 enters the input of the mixer 29, where the frequency fr of the generator 20 is subtracted. This is necessary to eliminate the influence of the location of the generator of the frequency of the generator fr 20 on the measurement accuracy. Using the bandpass filter block 30, the frequency fc of the original chirp signal is extracted.

The bandwidth of the filters 39 is chosen equal to the bandwidth DLZ 23,

and the total number is such as to cover the operating frequency range of the input chirp signals. Switching the filters 39 to extract the chirp signal with the center frequency fc is performed by the switch 40 in the bandpass filter bank 30 (Fig. 3). The chirp signal from the output of the bandpass filter unit 30 through the switch 34, the amplifier-limiter 31, the switch 35 is fed to

the inputs of the delay line 24 of the phase shifter 7 and the amplitude detector 25. Otherwise, the operation of the device is similar to operating in the measurement mode of the parameters of the chirp signals with a duration of more than several microseconds.

Thus, the goal is achieved by extending the input analyzed chirp signal in duration using the dispersion delay line

23, which expands the functionality of the device for measuring short-pulse chirped radio pulses with a large frequency deviation.

Claims (1)

Invention Formula A device for measuring the average rate of change of frequency and the linearity of the modulation characteristics of frequency-modulated oscillators according to the author. No. 1499259, I differ from the fact that, in order to expand the functionality by measuring the parameters of short-duration modulated short-duration signals, the second mixer, the first band-pass filter, the first switch, the first amplifier-limiter, the dispersion line are introduced into it delay, second switch, third mixer, the bandpass filter unit, the third switch, the second amplifier-limiter, the fourth switch, the output of which is connected to the input of the second delay line, the tunable generator, the output of which is connected to the second inputs of the second and third mixers, the second output of the second switch is connected to the second input of the third switch, the second the input of the fourth switch is connected to the second the input of the first switch, with the first input of the second mixer and is the input of the device. but Ifv TO to t / a JT Ј I d corner five Y / L.41