RU2782244C1 - Receiver - Google Patents

Abstract

FIELD: sound waves receiving technology.

SUBSTANCE: invention relates to a technology for receiving sound waves in water. The receiving device contains M receivers, M FIR filters, a combining device and a filter coefficient calculation unit. M FIR filters have a tap length shorter than the delay spread, i.e. possible range between the arrival time of the direct wave and the arrival time of the delayed wave from sound waves. M receivers are configured to receive signals based on sound waves propagating in water. The M FIR filters are configured to perform a waveform operation on the signals received by the receivers. The merging device is configured to combine the output signals of the M FIR filters. The filter coefficient calculation unit is configured to calculate the tap coefficient M of the FIR filters so as to reduce the error of the output signals

EFFECT: reducing the effect of waveform distortion caused by delay spread.

5 cl, 5 dwg

Description

The technical field to which the invention belongs

[0001] The present invention relates to technology for receiving sound waves in water.

State of the art

[0002] In water (especially in the sea), the absorption and attenuation of electromagnetic waves is very large. For this reason, sound waves are typically used as carrier waves for wireless transmission. However, sound waves have a very slow propagation speed compared to electromagnetic waves. Therefore, the possible delay time range between delayed waves that arrive after reflection from the sea surface, bottom, underwater structure, or the like, and direct waves that arrive without reflection (hereinafter referred to as "delay spread") is very big. Severe waveform distortion occurs when signals with large delay times overlap. In addition, low propagation speed increases the Doppler frequency difference (hereinafter referred to as "Doppler spread") between direct waves and delayed waves due to the influence of sea currents and waves. Since the phase difference between the direct wave and the delayed wave changes, the waveform distortion pattern (hereinafter referred to as "propagation path characteristic") changes with time. Both the amount of delay spread and the amount of Doppler spread are proportional to the reciprocal of the propagation velocity. Accordingly, the delay spread and Doppler spread of sound waves used in water are about 200,000 times greater than the delay spread and Doppler spread of electromagnetic waves used in air under the same conditions. For this reason, in order to be able to transmit information using sound waves, continuous compensation of waveform distortion, which changes rapidly with time, is required.

[0003] As a method for solving such a problem inherent in underwater acoustic communications, an adaptive compensation method has been proposed that adaptively calculates and uses an inverse FIR filter to characterize the propagation path between transmitters and receivers (eg, NPL 1).

[0004] FIG. 5 is a diagram showing a specific example of a receiver 90 that uses the adaptive equalization method described in NPL 1. The receiver 10 includes a receiver 901, a transform block 902, a finite impulse response (FIR) filter 903, a symbol estimator 904 and block 905 calculating the coefficient of the filter. The 901 receiver converts the sound waves in the water into electrical signals. Converter 902 samples the signals received by receiver 901. FIR filter 903 performs a waveform operation on the sampled signal. A symbol estimator 904 estimates the transmitted symbols based on the signal that has passed through the FIR filter 903. The filter coefficient calculator 905 adaptively calculates the tap gain of the FIR filter 903 so that the error between the signal passing through the FIR filter 903 and the estimated symbols is minimized. . The FIR filter 903 has a tap length that is long enough to cover the delay spread. That is, when the delay spread is σ_T seconds and the sampling rate of the FIR filter 903 is f_s Hertz, the number of taps N_tap is expressed by expression (1) below.

[0005] Mat. one

[0006] The N_tap multi-tap design is described on pages 33 and 80 of NPL 1 described above.

[0007] As described above, the FIR filter 903, which spans a time interval longer than the delay spread, is adaptively optimized. This optimization allows the FIR filter 903 to compensate for waveform distortion created by the superposition of the direct wave and the delayed wave. That is, it is possible to dynamically configure an equalization filter having an inverse characteristic for the propagation path characteristic. This processing compensates for the strong waveform distortion that changes over time.

[0008] As described in NPL 1, a configuration using multiple receivers 901 and multiple FIR filters 903 coupled to receivers 901 has also been proposed. In this configuration, the number of taps of the FIR filters 903 is also designed to satisfy expression (1) as in the above configuration. Each FIR filter 903 operates to compensate for waveform distortion created between the respective transmitter and receiver. That is, each FIR filter 903 operates to form an inverse cancellation filter for propagation path response, and a plurality of receivers 901 are used to obtain a diversity effect.

List of bibliographic references

Non-Patent Literature

[0009] [NPL 1]

Hiroshi Ochi, "Research on Underwater High-Speed Acoustic Transmission of Digital Data Using Wideband Transducers", The University of Electro-Communications Doctoral thesis, March 2009

The essence of the invention

Technical task

[0010] In principle, the time required to estimate the coefficient of a filter having an inverse characteristic for the propagation path characteristic is larger than the delay spread. On the other hand, the propagation path characteristic changes with time in proportion to the reciprocal of the Doppler spread. Thus, when the Doppler spread and the delay spread are large, the propagation path characteristic is changed while the coefficient of a filter having an inverse characteristic of the propagation path characteristic is estimated. Compensation for waveform distortion is therefore difficult to achieve with the conventional method.

[0011] The amount of Doppler spread is proportional to the frequency and speed of movement that are used. Therefore, when the high frequency band is used for high-speed communication, or when the transmitter or receiver moves at high speed, it is difficult to sufficiently compensate for waveform distortion by the conventional method. As a result, information cannot be transmitted. For this reason, the communication speed of traditional underwater acoustic devices has been limited to a few tens of kbps. Therefore, it has been difficult to communicate with underwater acoustic devices in a high-speed moving environment.

[0012] In view of the above circumstances, it is an object of the present invention to provide a method that can reduce the effect of waveform distortion caused by delay spread.

Troubleshooter

[0013] One aspect of the present invention is a receiver including: M receivers configured to receive signals based on sound waves propagating in water; M FIR filters configured to perform a waveform operation on the signals received by the receivers; a combining device configured to combine the outputs of the M FIR filters; and a filter coefficient calculator configured to calculate a tap coefficient M of the FIR filters so as to reduce the error of the output signals combined by the combiner. The M FIR filters have a tap length that is shorter than the delay spread, i.e., the possible range between the time of arrival of the direct wave and the time of arrival of the delayed wave from the sound waves.

Benefits of the Invention

[0014] The present invention can reduce the effect of waveform distortion caused by delay spread.

Brief description of the drawings

[0015] FIG. 1 is a schematic block diagram showing the basic principles of the functional configuration of the receiver 10 of the present invention.

Fig. 2 is a diagram showing the operation principle of the receiver 10.

Fig. 3 is a table showing the operating conditions for the simulations of the conventional receiver and the receiver 10 of the present embodiment.

Fig. 4 is a graph showing the results of simulations performed under the operating conditions shown in FIG. 3.

Fig. 5 is a diagram showing a specific example of a receiver 90 that uses the adaptive frequency response equalization method described in NPL 1.

Detailed description of embodiments

[0016] Referring to the drawings, an embodiment of the receiver of the present invention is now described in detail.

The essence of the invention

Fig. 1 is a schematic block diagram showing the basic principles of the functional configuration of the receiver 10 of the present invention. The receiver 10 of the present invention includes a plurality of FIR filters 103 having a tap length sufficiently short compared to the expected delay spread σ_T. The tap length is a value obtained by multiplying the sampling time interval by the number of taps. Since the FIR filters 103 have such a tap length, the FIR filters 103 cannot form filters with an inverse waveform distortion characteristic. However, the receiver 10 of the present invention is adapted to spatially remove the delayed wave, which is the root cause of waveform distortion. Thus, it is possible to reduce the effect of waveform distortion caused by a large delay spread. The details of the receiving device 10 of the present invention are now described.

[0017] Details

The receiver 10 includes M receivers 101 (101_1 to 101_M), M transform units 102 (102_1 to 102_M), M FIR filters 103 (103_1 to 103_M), a combiner 104, a symbol estimator 105, and a filter coefficient calculator 106 . M is an integer greater than or equal to 2. It is desirable that the same number of receivers 101, transform units 102 and FIR filters 103 be provided. In the following description, when a configuration common to M components of the same name is described, each component is indicated without using the reference "_1", for example. For example, the receiver is listed as "receiver 101" instead of "receiver 101_1".

[0018] The receiver 101 receives sound waves propagating in water and converts the received sound waves into electrical signals.

[0019] The conversion unit 102 samples the electrical signal converted by the receiver 101. In particular, this is done as follows. The conversion section 102 performs analog-to-digital conversion on the electrical signal converted by the receiver 101. Then, the conversion section 102 performs frequency conversion on the digital signal obtained by the analog-to-digital conversion. Other configurations (modifications) may be used for the conversion unit 102, and modifications will be described below.

[0020] The FIR filter 103 performs a waveform operation on the signal sampled by the transform block 102 . The number of taps of the FIR filter 103 is set such that the tap length can be quite short compared to the expected delay spread σ_T. That is, the number of taps of the FIR filter 103 is set based on the sampling time interval and the estimated delay spread σ_T.

[0021] The combiner 104 combines the M signals on which the waveform operations are performed by the FIR filters 103.

[0022] The symbol estimator 105 evaluates the symbols included in the sound waves that have propagated in the water based on the signals combined by the combiner 104. In other words, it evaluates the symbols included in the received sound wave at the time the sound wave was sent. .

[0023] The filter coefficient calculator 106 adaptively calculates the tap coefficient M of the FIR filters 103. The tap coefficient is adaptively calculated such that an error is minimized between the symbols indicated by the output of the combiner 104 and the symbols estimated by the symbol estimator 105 .

[0024] For example, when the distance which is the largest among the distances between the pairs of receivers 101 is d [m], and the speed of sound input is c [m/s], the number of taps Ntap of each of the FIR filters 103_1 to 103_M is given by 2 below. Since d is the largest distance among the distances between each pair of receivers 101 as described above, if all distances between each pair of receivers 101 are uniform, the uniform distance is the value of d.

[0025] Mat. 2

[0026] That is, a plurality of taps is designed such that the tap length is sufficiently short compared to the possible delay time range between the forward wave and the delayed wave (delay spread) and longer than the arrival time difference between the receivers 101 This number of taps may be common to all M FIR filters 103. A length that is "sufficiently shorter than the delay spread (σ_T)" may be, for example, one tenth of the delay spread or one hundredth of the delay spread. For example, the tap length can be defined as being sufficiently short compared to the delay spread when it is closer to the d/c value than to the median between the delay spread and the d/c value.

[0027] The plurality of taps are now described in detail. In the receiver 10 of the present embodiment, a plurality of receiver systems (combinations each including a receiver 101, a transform unit 102, and a FIR filter 103) are provided. The tap length of each FIR filter 103 is set sufficiently short compared to the delay spread as described above. The filter coefficient calculator 106 adaptively optimizes the FIR filters 103 of these receiver systems. Because the tap length of the FIR filters 103_1 to 103_M is sufficiently short compared to the delay spread, these filters do not form filters with an inverse waveform distortion characteristic. On the other hand, the receiver 10 of the present embodiment uses a waveform equalization approach, which is completely different from the conventional approach, and spatially eliminates the delayed wave, which is the root cause of waveform distortion.

[0028] Details are given below. Fig. 2 is a diagram showing the principle of operation of the receiving device 10. In the example of FIG. 2, it is assumed that the transmitted signal is a pulse signal, the number of receivers 101 is two (101_a and 101_b), and the number of delayed waves is one, for the sake of intuition. Also, for convenience, a configuration is described in which the additional FIR filter 103_c is connected in the subsequent stage of the combiner 104. In this configuration, each of the FIR filters 103_1 to 103_M is divided into two FIR filters 103a and 103b, and placed in the stage following the merging device 104. That is, the configuration shown in FIG. 2 can perform an operation equivalent to that of the receiver 10.

[0029] As shown in FIG. 2, the FIR filter 103_a and the FIR filter 103_b in the previous step perform, as a waveform operation, a time shift and a phase inversion so that the two delayed waves have the same time point and opposite phases. This waveform operation removes the delayed wave component in the combined signals. As a result, only the waveform distortion caused by the superposition of the direct waves received by the two receivers 101_a and 101_b remains. Then, the FIR filter 101_c in a subsequent step compensates for waveform distortion caused by superposition of direct waves. The alignment of the frequency response of the waveform is completed by the above processing.

[0030] Among 101 receivers, the difference between direct waves in arrival time and the difference between delayed waves in arrival time depend only on the distances of transmitted and received waves and are thus much shorter compared to the time difference between direct wave and delayed wave ( delay spread). As such, the waveform operation described with reference to FIG. 2 can be implemented with a FIR filter 103 having a very short tap length. Additionally, among the receivers 101, the arrival time difference and phase difference between direct waves and the arrival time difference and phase difference between delayed waves are constant as long as the directions of arrival of direct waves and delayed waves do not change. Thus, they are more stable compared to the time difference and phase difference between the direct wave and the delayed wave. For this reason, unlike a conventional receiver which estimates the reciprocal response for the propagation path, which depends on the time difference and phase difference between the direct wave and the delayed wave, waveform equalization can be carried out in a largely stable manner even when the path propagation has a large Doppler spread.

[0031] The filter coefficient calculator 106 does not need to use the new configuration to perform the operation explicitly described with reference to FIG. 2, and it is sufficient that the filter coefficient calculator 106 has the traditional function of the filter coefficient calculator. That is, the tap length of the FIR filter 103 is deliberately set to be sufficiently short compared to the delay spread, and also long compared to the arrival time difference between the receivers 101, resulting in the FIR filter 103 which performs the operation, described with reference to FIG. 2. The reason is that such a configuration can theoretically perform the operation described with reference to FIG. 2, and the error of the signal output by performing the operation is minimized.

[0032] As described with reference to FIG. 2, an operation equivalent to that of the configuration in FIG. 1 could theoretically be done with a configuration in which the FIR filter 103_c is further connected in a subsequent stage of the combiner 104. As such, either the configuration in FIG. 1 or the configuration in FIG. 2 can be used. However, in the cascaded configuration of the FIR filter 103, as shown in FIG. 2, the error characteristic surface implies a 4th order function so that the configuration has a local minimum solution. Therefore, an adaptive algorithm such as LMS or RLS may cause convergence to an erroneous filter coefficient. This is also described in the following reference literature.

[0033] References: Kazunori Hayashi, Shinsuke Hara, "A Spatio-Temporal Equalization Method with Cascade Configuration of an Adaptive Antenna Array and a Decision Feedback Equalizer", The transactions of the Institute of Electronics, Information and Communication Engineers. B, Vol.J85-B, No.6, pp.900-909

[0034] For this reason, in practice, it is desirable to have a configuration that does not involve cascading the FIR filter 103 in a subsequent stage, as shown in FIG. 1. Each FIR filter 103 in the configuration of FIG. 1 has the functions of the FIR filter 103 in the previous stage (103_a or 103_b) and the FIR filter 103 in the subsequent stage (103_c) in FIG. 2.

[0035] The receiver 10 of the present embodiment has the ability to remove delayed waves that are one less than the M receivers 101. As such, the natural number "M" for the number of receivers 101 and FIR filters 103 (the number of receiver systems) is preferably a positive integer greater than the number of delayed waves expected to arrive at the receivers 101 (delayed waves to be processed for removal).

[0036] FIG. 3 is a table showing the operating conditions for the simulations of the conventional receiver and the receiver 10 of the present embodiment. Fig. 4 is a graph showing the results of simulations performed under the operating conditions shown in FIG. 3. In particular, FIG. 4 shows the bit error rate after character determination versus moving speed, as a result of the simulations. As is evident from FIG. 4, in 400 kbps (200 kbaud, QPSK) high-speed communication, the conventional receiver cannot update the filter at the time when the moving speed exceeds about 0.001 m/s, and the bit error rate starts to be allowed. Accordingly, it is impossible to follow small changes in the propagation path, such as waves and tidal currents, indicating that communication is difficult to use in practice even in a stationary environment. On the contrary, the receiver 10 according to the present embodiment can maintain a bit error rate of 0.1% or less even in an ultra-high-speed moving environment exceeding 10 m/s. The simulation results suggest that the conventional receiver has better bit error rate performance when the moving speed is very low. The reason is that a conventional receiver can perform perfect waveform equalization by compensating with a filter having an inverse propagation path response as long as it can update the filter in a timely manner.

[0037] As described above, in the receiver 10 of the present embodiment, a tap length sufficiently short compared to the delay spread is used in FIR filters 103 connected to a plurality of receivers 101. Also, a configuration is applied in which a filter coefficient for The FIR filters 103 are adaptively optimized. Such a configuration can spatially remove the delayed wave, which is the root cause of waveform distortion. Spatial removal of the delayed wave enables high-speed communication in an underwater high-speed moving environment.

[0038] Modifications

In the present embodiment, conversion units 102 may be configured to perform only A/D conversion. Converters 102 may be configured to perform frequency conversion on an analog signal and then perform analog to digital conversion.

[0039] The physical intervals M of the receivers 101 may all be equal or may differ from each other.

[0040] In addition, a configuration can be used in which a feedback filter for removing the residual error of the FIR filter (feedforward filter) as described in NPL 1 is connected in a subsequent step.

[0041] The symbol estimator 105 and the filter coefficient calculator 106 of the receiver 10 are configured with a processor such as a CPU and a memory. The symbol evaluator 105 and the filter coefficient calculator 106 operate when the processor reads and executes the program stored in the memory. The program may be recorded on a computer-readable recording medium. A computer-readable recording medium is a non-volatile storage medium, which may be a removable medium such as a floppy disk, magnetic optical disk, ROM, or CD-ROM, or a storage device, etc., such as a hard disk built into a computer system. The program may be transmitted over a telecommunications line. Some or all of the operations of the symbol estimator 105 and the filter coefficient calculator 106 may be implemented in hardware including an electronic circuit using an LSI, ASIK, PLD, FPGA, or the like.

[0042] Some embodiments of the present invention are described above in detail with reference to the drawings, but specific configurations are not limited to these embodiments and include designs and the like. in a range that does not deviate from the essence of the present invention.

Industrial Applicability

[0043] The present invention is applicable to communication using sound waves in water.

List of reference marks

[0044] 10 Receiver

101 receiver

102 Conversion block

103 FIR filter

104 Uniting device

105 Symbol evaluation block

106 Filter coefficient calculation block

Claims (10)

1. A receiving device, comprising: M receivers configured to receive signals based on sound waves propagating in water; M FIR filters configured to perform a waveform operation on the signals received by the receivers; a combining device configured to combine the outputs of the M FIR filters; and a filter coefficient calculation unit configured to calculate the tap coefficient M of the FIR filters so as to reduce the error of the output signals combined by the combiner, wherein the M FIR filters have a tap length that is shorter than the delay spread, i.e., the possible range between the time of arrival of the direct wave and the time of arrival of the delayed wave from the sound waves. 2. The receiver according to claim 1, wherein the length of the tap is longer than the time obtained by dividing the distance, which is the largest among the distances between each pair of M receivers, by the speed of sound in water. 3. The receiving device according to claim 1 or 2, wherein the value of M is a positive integer greater than the number of delayed waves to be processed. 4. The receiver of claim 1, further comprising M conversion units configured to convert the sound waves received by the receivers into digital signals. 5. The receiver of claim 4, wherein the conversion units are configured to further perform frequency conversion on the sound wave signals.

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