RU2770889C1 - Fully digital flowmeter based on measurement of transit time, in which time-reversed acoustics is used - Google Patents

Abstract

FIELD: physics.

SUBSTANCE: invention relates to a microcontroller and a method for determining flow rate using an electronic processing unit of an ultrasonic flow meter, based on measurement of transit time, with signals of arbitrary shape. Electronic processing unit comprises receiving and transmitting leads, a signal processing unit and a signal generating unit which is configured to generate an oscillatory electrical output signal with a time-dependent amplitude, wherein the time-dependent amplitude varies in accordance with the stored signal parameters. Main feature of the invention is the use of digital filtering using a time-reversed acoustic signal, which eliminates the need for a DAC with an analogue part, which can be replaced by a pulse generator.

EFFECT: simplification of electronics with increased accuracy and reduced power consumption.

14 cl, 22 dwg

Description

The present invention relates to flow meters, and more specifically to ultrasonic flow meters based on transit time measurement.

Various types of flow meters are currently used to measure the volumetric flow rate of a fluid, such as liquid or gas, through a pipe. Ultrasonic flowmeters are either Doppler flowmeters, which use the acoustic Doppler effect, or transittime-based flowmeters, sometimes also called signal-based flowmeters, which use the difference in propagation time caused by the relative motion of the source and medium. Transit time is also called time of flight or transfer time.

The transit time-based ultrasonic flow meter evaluates the difference in propagation time of ultrasonic pulses propagating in the direction of flow and against it. Ultrasonic flowmeters are provided as inline flowmeters, also known as submersible or fluid-filled flowmeters, or as fixed flowmeters, also known as non-immersible flowmeters. Other types of flowmeters include Venturi channels, weirs, radar flowmeters, Coriolis flowmeters, differential pressure flowmeters, magnetic inductive flowmeters, and other types of flowmeters.

In the presence of non-standard flow profiles or open channels, more than one propagation path may be needed to determine the average flow rate. Among other things, multipath propagation procedures are described in hydrometric standards such as IEC 41 or EN ISO 6416. As an additional application, ultrasonic flow meters are also used to determine flow profiles, for example with an acoustic Doppler flow meter (ADCP). The ADCP is also suitable for measuring the velocity and flow of water into rivers and open waters.

White paper SLAA814-December 2017 "Matched-Filter Ultrasonic Sensing: Theory and Implementation" by Zhaohong Zhang describes the operation principle of the matched filter ultrasonic location technology and the single-chip platform for its implementation using the microcontroller (TI) MSP430FR6047.

It is an object of the present invention to provide an improved transfer time based flow meter and a corresponding computer-based method for measuring the average flow rate or flow profile of a fluid in general, and specifically liquids such as water and/or gases.

The flow measurement device of the present invention uses sonic transducers, for example in the form of piezoelectric elements, also known as piezoelectric transducers, to generate and receive a measurement signal.

Alternative sound transmitters include lasers that vibrate a metal membrane or other light-absorbing surface, or electrodynamic loudspeakers. In other embodiments, the flowmeter generates compression waves in other ways, such as with a MEMS device, using a piezoelectric membrane, etc. The receiving side may also be provided by other means that are different from piezoelectric transducers but detect ultrasonic waves.

Although the term "piezoelectric transducer" is often used in this specification, it also refers to other sound wave transducers that produce or detect ultrasonic waves.

The measurement signal according to the present invention can be generated by a matched filter. When the word "signal" is used with reference to the signal control step, it may specifically refer to the representation of the signal in a computer-readable memory.

In particular, the signal representation can be defined by a pair of digitized amplitude values and the associated discrete time.

The ultrasonic flow meter of the present invention can provide the desired focusing power properties by using an arbitrary waveform to obtain the ultrasonic flow meter with a signal with the desired properties in the receiving transmitter or by post-receiving ADC calculations.

For example, the frequency of sound waves that are used in the flow meter according to the present invention may be from 20 kHz to 2 MHz, which corresponds to an oscillation period of 0.5 microseconds (µs), but may even be up to 800 MHz.

In one aspect, the present invention describes a microcontroller for determining the flow rate using an ultrasonic transit time-based flow meter using a time-reversed digitized signal. In particular, this flowmeter may be "fully digital" in the sense that the evaluation of the signals is done by digital signal processing and the electrical signals supplied to the ultrasonic transducers are step signals, which can also be considered digital signals. In other words, these electrical signals comprise a sequence of discrete voltage levels that are constant for predetermined sampling times. This eliminates the need for a digital-to-analogue converter (DAC) with an analog part, and the DAC can be replaced by a pulse generator.

The microcontroller is configured to generate arbitrary pulse signals. In particular, the microcontroller may be configured to output pulse trains of arbitrary frequency and duration.

The microcontroller of the present invention may include a plurality of buffer output pins, such as four or eight output pins instead of one. To this end, the microcontroller may comprise a plurality of buffers, eg four or eight buffers, in a parallel arrangement, which makes it possible to time-multiplex output all signals to all respective channels or buffers.

In other words, a transmitter path, or a signal generation path in a chip or FPGA, or a path connected to such a chip or FPGA, is configured to generate arbitrary pulse trains with an arbitrarily selected pulse width. As used herein, "arbitrary" means arbitrary within a predetermined range and precision, such as within a predetermined range between the minimum and maximum frequency and within a predetermined precision.

The microcontroller contains the first receiving terminal for connection with the first ultrasonic measuring transducer, the first transmitting terminal for connecting with the first ultrasonic measuring transducer, the second receiving terminal for connecting with the second ultrasonic measuring transducer, the second transmitting terminal for connecting with the second ultrasonic measuring transducer. Receive pins and transmit pins are also called "connectors".

In one embodiment, the first receive terminal and the first transmit terminal match the second receive terminal and the second transmit terminal. This can be especially useful when there is no overlapping send and receive.

The first receiving terminal and the second receiving terminal are connected to a signal processing unit or signal processing means which are provided for evaluating the signals received from the measuring transducers. The signal processing block contains an analog-to-digital converter (ADC) and an evaluation circuit. Unlike a simple timing scheme, ADC can provide various options for calculating the waveform.

A predicted waveform may be used to reduce the sampling rate and/or amplitude resolution of the ADC. In particular, the waveform of the present invention may find application in energy saving applications. To further reduce power consumption, the sampling rate and/or amplitude of the ADC amplitude digitization accuracy may also be non-uniform and time dependent, and this time dependence may depend on the predicted waveform. In addition, the ADC can only be turned on after a predetermined time after sending the measuring signal and turned off again after receiving the response signal.

The first transmission output and the second transmission output are connected to the signal generation unit. The signal generation unit contains a memory device for storing signal parameters and a buffer.

The signal generation unit and the signal processing unit are configured to send an oscillatory electrical output signal to the first transmit terminal, and to receive a response signal at the second receive terminal, and to obtain a fluid flow rate based on at least the received response signal.

In one embodiment, the signal generating block buffer is connected to a pulse width modulation (PWM) block, which is connected to a low pass filter. Pulse width modulation can provide a simple and reliable way to generate an analog electrical signal based on a digital signal, which is then applied to an ultrasonic transducer, such as a piezoelectric element, a loudspeaker piezomembrane or membrane, an MEMS element, or another type of transducer. Through the use of power electronics, the output signal of the pulse width modulation can be made strong enough that additional signal amplification is not required in many cases. PWM can provide sufficient quality of the resulting analog signal.

To generate a stepped or step-shaped output signal, the microcontroller's PWM can also be used with little or no low-pass filtering after buffer or PWM output. To this end, the low-pass filter can be provided as an adjustable or switchable low-pass filter.

PWM signals store amplitude information in signal timing information. The resolution of the PWM signal is affected by the granularity of the temporal resolution. For this reason, the PWM signal generator can be configured to provide a higher, eg 10 times higher frequency than a comparable ADC. The amplitude of the PWM signal generator may be equal to or similar to the pulse generator.

In a further embodiment, the microcontroller's signal generation unit comprises a user programmable gate array (FPGA) that is connected to a digital input/output connector. The signal generating unit is configured to obtain an oscillatory signal based on the FPGA output signal, wherein the FPGA output signal is modified by delivering software commands to the digital input/output connector.

In particular, the FPGA may contain a logic device or circuit layout to obtain the required output signal based on the provided signal parameters. Among other things, these signal parameters can be read from a memory device, pre-calculated or calculated in real time. In one embodiment, the calculation of the signal parameters from which the output signal is obtained depends on the received signal.

In a further embodiment, the signal generating unit is configured to generate a signal sequence, the signal sequence comprising waveform portions separated by predefined silence periods in which the output voltage is constant, in particular during the silence period, the output voltage may be zero. In one embodiment, portions of the waveform are repeated at regular intervals. In another embodiment, the duration of the parts of the waveform and the time intervals between them is determined by calculation. In particular, the calculation of these time periods may include a random component, or "jitter", within a predetermined range of amplitudes, whereby parts of the waveform have a randomly varying distance in time from one another.

According to a further embodiment, generating a waveform signal includes applying pulse width modulation and applying a low pass filter to the output pulse width modulation signal.

In yet another further embodiment, generating a waveform signal includes generating a waveform sequence, the waveform sequence comprising portions of the waveform separated by predefined periods of silence, in which the output voltage is constant, where the constant voltage level may be zero.

During the measurement of the flow rate, the microcontroller applies a second measurement signal to one of the measurement transducers, receives a response signal to the second measurement signal in the other of the measurement transducers, and processes the response signal to obtain the flow rate.

According to one aspect, the present invention describes a computer-executable method for determining the flow rate of a fluid in a fluid conduit or conduit, in particular a pipe or tube, using an ultrasonic flow meter based on transmission time measurement. In a preferred embodiment, "computer-executed" refers to execution on fine-grained electronic components such as microprocessors, ASICs, FPGAs, and the like, which can be used in portable or compact stationary digital signal processing devices that are typically smaller larger than workstations or base computers, and which can be placed at the desired location along the fluid conduit.

Further, the terms "channel", "pipeline", "passage", etc. are used as synonyms. The object of the present invention can be applied to all types of fluid conduits, regardless of their respective shape and regardless of whether they are open or closed. The object of the present invention can also be applied to all types of fluids or gases, whether they are gases or liquids, or a mixture of both.

Throughout this application, the term "computer" is often used. Although a computer refers to devices such as a laptop or desktop computer, the transmission and reception of signals can also be performed by microcontrollers, application specific integrated circuits (ASICs), FPGAs, etc.

In addition, the connecting line between the transducers may be offset from the center of the fluid conduit to obtain a flow velocity in a predetermined layer, and there may be more than one pair of transducers. In addition, the measurement signal may be provided by more than one measurement transducer and/or the response signal to the measurement signal may be measured by more than one measurement transducer.

Furthermore, according to the present invention, a flow measuring device is disclosed with a first piezoelectric transducer that is connected to a first connector and a second ultrasonic transducer, such as a piezoelectric transducer, that is connected to a second connector. In particular, ultrasonic transducers, such as piezoelectric transducers, may be provided with attachment means, such as a clamping mechanism, for attaching them to the pipe.

In addition, according to the present invention, a flow measurement device with a pipe part is disclosed. A first ultrasonic transducer, such as a piezoelectric transducer, is installed on the pipe part at a first location, and a second ultrasonic transducer, such as a piezoelectric transducer, is installed on the pipe part at a second location. In particular, the measuring transducers can be fixed to a part of the pipe. Providing the device with a pipe part can provide advantages when the device is pre-adjusted with respect to the pipe part.

The device can be made compact and portable. The portable device of the present invention, which is equipped with surface-mountable transducers, such as mechanically attached transducers, can be used to test a pipe at any available location. In general, the device may be fixed or portable. Preferably, the device is compact enough to be placed in the desired location and sufficiently protected from environmental conditions such as moisture, heat and corrosive substances.

In particular, dedicated electronic components may be provided with electronic components comprising the aforementioned computer-readable memory. In other embodiments, the application specific electronic components are provided with hardwired or configurable circuitry such as an application specific integrated circuit (ASIC) or a user programmable gate array (FPGA).

The training procedure for signal training may be carried out as described below. In this document, the signal generated by the training procedure may refer to a digital filter or other discrete time sequence. This signal may be used to generate an audio signal, or it may be used for calculations in processing a digitized version of the received audio signal.

Training is performed by capturing the step response of the channel under controlled conditions, for example, in zero flow mode. It can be done using one or more pulses as a transmit signal that is transmitted over a channel. As used herein, "channel" refers to a communication channel containing a fluid.

The stepped response is then processed as follows.

In the case of a generated PWM signal, the step response is time-reversed and digitized such that the transmitted PWM signal corresponds to the time-reversed step response.

In the case of input as a pulse train, the step response of the channel after analog-to-digital conversion is stored in the digital receive filter.

The measurement phase can then be carried out as follows.

In the case of using a PWM signal, the time-reversed channel impulse response obtained by the learning process is used as a transmitted signal by delivering the time-reversed digital signal as an input to the PWM device and connecting the output of the PWM device to an ultrasonic transducer.

In the case of using a pulse train, the same signal as used in the teach phase can be used as the transmitted signal or measurement signal in the measurement phase. For example, a step signal, or a single pulse, or a train of pulses can be used as the transmitted signal or as the measuring signal. In one example, the pulse train contains 10-30 pulses. Alternatively, the measurement signal may be longer or shorter than in the learning phase.

In the case of using a PWM signal as the transmitted signal, the transmitted signal is convolved by the channel impulse response and a pronounced peak is received at the ADC input. In other words, in the LTI (linear stationary system) approximation, the received response signal is a convolution of the transmitted signal with a predetermined channel impulse response.

The received signal is digitized and the peak of the signal is used, for example, to obtain transit time information. This can be done either by a high resolution sampling process or alternatively by interpolation of the received signal. Said signal peak detection belongs to PWM, with peaks appearing only immediately after ADC if the correlation process is performed on the channel.

The peak can also be computed after the ADC if the stored inverse filter is correlated with the receive filter.

In the case of using a pulse train as a transmitted signal, signal processing, as explained, can be performed by the DTRAF or DTRAC method according to the present invention. The DTRAC method can provide greater processing power efficiency compared to the DTRAF method. One reason for the greater efficiency is that the DTRAF method filters all received signals with FIR filters. On the other hand, in the DTRAC method, this processing-intensive operation is not performed, and instead, the correlation of the digitized measurement signals is performed directly.

In the case of DTRAF, the received signal is filtered using the reciprocal of the channel response that has been stored previously, typically in a Finite Impulse Response (FIR) filter after the training phase. As a result of this filtering operation, a pronounced peak is obtained. This operation can be performed on both upstream and downstream signals. The resulting signals are then subjected to cross-correlation and interpolation.

Second, in the case of DTRAC, the measurement process ADC signals are cross-correlated with the training signals and then interpolated. This can also be done for upstream and downstream signals, so that the travel time difference ΔT between the upstream and downstream signals can be calculated.

The object of the present invention is explained in more detail below with reference to several figures, in which:

in fig. 1 shows a first arrangement of a flow meter with two piezoelectric elements;

in fig. 2 shows the layout of the flowmeter of FIG. 1 with one direct signal and two scattered signals;

in fig. 3 shows the layout of the flowmeter of FIG. 1 when viewed in the direction of flow;

in fig. 4 shows a second flowmeter arrangement with four piezoelectric elements and four direct signals;

in fig. 5 shows the layout of the flow meter of FIG. 4 when viewed in the direction of flow;

in fig. 6 shows an arrangement of many-to-one flow sensors according to the present invention;

in fig. 7 shows the arrangement of one-to-many flow sensors according to the present invention;

in fig. 8 shows a one-to-one sensor arrangement for layer flow measurement according to the present invention;

in fig. 9 shows an arrangement with multiple sensors for measuring flow in several layers according to the present invention;

in fig. 10 shows a device for measuring the flow rate according to the present invention;

in fig. 11 depicts an iterative procedure for obtaining a waveform for use in the flowmeter of FIG. ten;

in fig. 12 depicts an additional procedure for obtaining a waveform for use in the flowmeter of FIG. ten;

in fig. 13 depicts an additional procedure for obtaining a waveform for use in the flowmeter of FIG. ten;

in fig. 14 shows a comparison of a PWM signal approximating a time-reversed TRA signal (time-reversed acoustic signal) with a time-reversed signal;

in fig. 15 shows a comparison of the response signal to the PWM signal of FIG. 15 and the time-reversed response signal of FIG. fifteen;

in fig. 16 shows simulation results for PWM-based correlated TRA;

in fig. 17 shows simulation results for signals generated by the digital correlation method using TRA shown in FIG. 21;

in fig. 18 shows simulation results for signals generated by the TRA digital filtering method shown in FIG. nineteen;

in fig. 19 shows a visualization of the DTRAF method in the form of a block diagram;

in fig. 20 shows an image of the overall interpolation of the peaks to determine the maximum value of the correlation function;

in fig. 21 shows a visualization of the DTRAC method in the form of a block diagram; and

in fig. 22 shows a further embodiment of the proposed flow rate measuring device.

The following description provides details for describing embodiments of the present invention. However, it should be obvious to one skilled in the art that the embodiments may be practiced without such details.

Some parts of the embodiments that are shown in the figures have similar parts. Like parts have the same name or like part numbers with a main character or with an alphabetic character. The description of such similar parts is also applicable by reference to other similar parts where appropriate, thereby reducing repetition of text without limiting the present invention.

In FIG. 1 shows a first arrangement 10 of a flow meter. In the flow meter arrangement, the first piezoelectric element 11 is located on the outer wall of the pipe 12, which is also called the tube 12. The second piezoelectric element 13 is located on the opposite side of the pipe 12 so that a straight line between the piezoelectric element 11 and the downstream piezoelectric element 13 is oriented at an angle β to the direction 14 of the mean flow, which at the same time is also the direction of the axis of symmetry of the pipe 12. The angle β is chosen to be approximately 45 degrees in the example of FIG. 1, but it may also be larger, such as 60 degrees, for example, or smaller, such as 30 degrees, for example.

A piezoelectric element, such as the piezoelectric elements 11, 13 of FIG. 1 can generally operate as an acoustic transmitter and as an acoustic sensor. The acoustic transmitter and the acoustic sensor may be provided by the same piezoelectric element or different portions of the same piezoelectric element. In this case, the piezoelectric element or transducer is also called a piezoelectric transmitter when it works as a transmitter or sound source, and it is also called an acoustic sensor or receiver when it works as an acoustic sensor.

When the flow direction is as shown in FIG. 1, the first piezoelectric element 11 is also referred to as the "upstream" piezoelectric element, and the second piezoelectric element 13 is also referred to as the "downstream" piezoelectric element. The flow meter according to the present invention operates in both flow directions essentially the same, and the flow direction of FIG. 1 is provided as an example only.

In FIG. 1 shows the electrical signal flow of FIG. 1 for a configuration in which the upstream piezoelectric element 11 operates as a piezoelectric transducer and the downstream piezoelectric element 13 operates as an acoustic sensor. For clarity, the device operates upstream and downstream, i.e. the positions of the piezoelectric elements can be interchanged.

The first calculation unit 15 is connected to the upstream piezoelectric element 11, and the second calculation unit 16 is connected to the downstream piezoelectric element 13. The first calculation unit 15 includes a first digital signal processor, a first digital buffer, and a first analog-to-digital converter (ADC). ). Similarly, the second calculation unit 16 includes a second digital signal processor, a second digital buffer, and a second analog-to-digital converter (ADC). The first calculation unit 15 is connected to the second calculation unit 16 .

The arrangement with two calculation units 15, 16 shown in FIG. 1 is provided by way of example only. Other embodiments may have different numbers and layouts of calculation units. For example, there may be only one central calculus, or there may be two ADCs or buffers and one central calculus, or there may be two small transmitter calculators and one larger central calculus.

The calculation unit or units may be provided with, for example, microcontrollers or application specific integrated circuits (ASICs) or user programmable gate arrays (FPGAs).

The method for executing the measurement process according to the present invention includes the following steps.

The predetermined digital measurement signal is generated by synthesizing the electrical signal with the digital signal processor of the first calculation unit 15 . This electrical signal is sent from the first calculation unit 15 to the piezoelectric measuring transducer 11 via the signal path 17 . The piezoelectric transducer 11 generates a corresponding ultrasonic test signal. Blocks 15 and 16 may also be provided in one separate block.

The measuring signal is provided as an arbitrary waveform. For example, an arbitrary waveform can be represented by a pulse width modulated wobble at a fundamental frequency in the MHz range, such as a 1 MHz wobble. The arbitrary waveform may also represent a single pulse.

The ultrasonic test signal is passed through a fluid (eg liquid) in conduit 12 to a piezoelectric transducer 13. FIG. 1, the forward path of the ultrasonic signal is indicated by an arrow 18. Similarly, the direct path of the ultrasonic signal in the reverse direction is indicated by an arrow 19. The response signal received by the piezoelectric sensor 13 is sent to the second calculation unit 16 along the signal path 20 and is digitized by the second block 16 calculation.

In a further step, a digital measurement signal is obtained based on the digitized response signal by means of signal processing. According to additional embodiments, obtaining the measurement signal includes additional processing steps.

In a flow meter according to one embodiment of the present invention, the same measurement signal is used for both downstream and upstream directions 18, 19, providing a simple and efficient arrangement. In other embodiments, different measurement signals are used for both directions. In particular, the measurement signal may be applied to the initial receiver of the test signal. Such arrangements can provide advantages for asymmetrical pipe positions and shapes.

The operating principle of the flowmeter is explained in more detail below with reference to FIG. one.

The ultrasonic measurement signal passes through the liquid in the pipe 12 to the piezoelectric sensor 13. The response signal is received by the piezoelectric sensor 13, sent to the second calculation unit 16 along the signal path 20, and digitized by the second calculation unit 16.

A similar process is carried out for the signal passing in the reverse direction 19, namely, the aforementioned measuring signal is applied to the downstream piezoelectric element 13, and the response signal is measured by the upstream piezoelectric element 11 to obtain the upstream time of flight TOF _up in the reverse direction. 19. The first calculation unit 15 determines the flow rate, for example, according to the formula:

where L is the length of the direct path between the piezoelectric elements 11, 13, β is the angle of inclination of the direct path between the piezoelectric elements 11, 13 and the direction of the average flow, and c is the speed of sound in the liquid at a given pressure and temperature conditions.

The time of flight upstream has the form:

and the time of flight downstream has the form:

which leads to the formula:

Using this formula, it is not necessary to determine the temperature or pressure, which in turn determine the density of the fluid and the speed of sound, or directly measure the speed of sound or the density of the fluid. Conversely, the first error component is not excluded for only one direction of measurement.

Instead of using the factor L∙cos(β), a flow-dependent value can be derived from calibration measurements with a known flow rate. These calibration values take into account additional effects such as flow profiles and contributions from sound waves that are scattered and not traveling in a straight line.

According to another embodiment, the measurement signal to be applied to the transmitting piezoelectric element is synthesized using an arbitrary waveform.

According to a further embodiment of the present invention, the time-of-flight of a signal is estimated using a cross-correlation method. In particular, the corresponding time offsets can be estimated by cross-correlating a signal received downstream or upstream with a signal received in zero flow mode, according to the formula:

- временные переменные, Sig_Flow представляет сигнал выше по потоку или ниже по потоку в условиях измерения, когда имеется поток текучей среды через трубу, и где Sig_NoFlow представляет сигнал в условиях калибровки в режиме нулевого расхода. Пределы бесконечной суммы представляют достаточно большое временное окно [-T1, +T2] от момента времени T1 до момента времени T2. В общем, -T1 и +T2 не должны быть одинаковыми, и с практической точки зрения это может быть преимущественным для расходомера.where t and

- time variables, Sig _Flow represents the signal upstream or downstream under measurement conditions when there is fluid flow through the pipe, and where Sig _NoFlow represents the signal under calibration conditions in zero flow mode. The infinite sum limits represent a sufficiently large time window [-T1, +T2] from time T1 to time T2. In general, -T1 and +T2 do not have to be the same, and from a practical point of view this can be advantageous for the flowmeter.

The time offset TOF _up - TOF _down is then obtained by comparing the time of maximum value of the upstream correlation function with the time of maximum value of the correlation function downstream. The envelope of the correlation function can be used to more accurately determine the location of the maximum.

In yet another embodiment, a separate estimator is provided between the first estimator 15 and the second estimator 16, which evaluates the signal arrival time and flow rate values.

In general, the measured acoustic sensor signal is obtained based on the superimposition of the scattered signals and the direct signal. Scattered signals are reflected from the inner and outer walls of the pipe one or more times, including the processes of additional scattering in the pipe wall. This is shown as an example in FIG. 2.

The arrival time can also be determined using a matched filter technique. According to a simple model based on the assumptions of rectilinear signal propagation and specular reflections from the pipeline walls, the received response signal can be modeled as:

x(t)= A*s(t-TOA) + n(t),

where t is the time variable, A is the attenuation factor, s is the emitted signal time-shifted by an unknown TOA arrival time, and n(t) is the noise component. The TOA arrival time is then obtained by correlating the received signal x(t) with the time-shifted measurement signal s(t) according to the following:

y(t)= Integral(-inf, inf, x(τ)s(t- τ)),

where -inf, inf are the limits of integration from minus infinity to plus infinity, and the convolution x(τ)s(- τ) is the function to be integrated over the time variable τ.

The arrival time is the time value or function argument for which the correlation becomes maximum:

TOA = argmax[y(t)].

The above correlation can also be expressed as a convolution using a "matched filter" h(t) taking the form:

where a is the normalization factor and s(t) is the measurement signal. This procedure can be generalized for the case of multiple receivers as follows. A set of phase shifts is determined that maximizes the sum of the response signal amplitudes of the individual receiving transducers. The arrival time is determined by applying a matched filter to a function of the sum of the individual received response signals offset by a predetermined set of phase shifts.

The transducer configuration of FIG. 1 is straight. Other arrangements that use reflections on the opposite side of the pipe are also possible, such as V-shaped and W-shaped configurations. V-shaped and W-shaped configurations operate on the basis of reflections on the pipe wall, which induce more scattering than a straight-line configuration. The object of the present invention is advantageous with these configurations, provided that the paths are understood correctly.

In a V-configuration, two transmitters are mounted on the same side of the pipe. To record the reflection at a 45 degree angle, they are placed at a distance approximately equal to the diameter of the pipe, in the direction of flow. The W-shaped configuration uses three reflections. Similar to the V-configuration, two transmitters are mounted on the same side of the pipe. To record the signal after two reflections at an angle of 45 degrees, they are placed at a distance equal to two pipe diameters in the direction of flow.

In FIG. 2, by way of example, a first acoustic signal "1" is shown, which passes directly from the piezoelectric element 11 to the piezoelectric element 13, and a second acoustic signal "2", which is scattered twice around the circumference of the pipe 12.

For simplicity, scattering cases are shown in Fig. 2-5 as reflections, but in reality the scattering process can be more complex. In particular, scattering, which is of the greatest importance, usually occurs on the pipe wall or on the material that is installed in front of the piezoelectric transducers. In FIG. 3 is a view of FIG. 2 in the direction of flow in the direction of observation A-A.

In FIG. 4 and 5 show a second sensor arrangement in which the additional piezoelectric element 22 is positioned at a 45 degree angle to the piezoelectric element 11 and the additional piezoelectric element 23 is positioned at a 45 degree angle to the piezoelectric element 13.

In addition, in FIG. 4 and 5 show the acoustic signal paths as straight or even lines for a situation in which the piezoelectric elements 11, 22 act as piezo transducers and the piezoelectric elements 13, 23 act as acoustic sensors. The piezoelectric element 23, which is located behind the tube 12 in the form of FIG. 4 is shown by a dotted line in FIG. 4.

In FIG. 6-9 show by way of example various mounting piezoelectric transducer arrangements for which flow measurement according to the present invention can be used. By delivering arbitrary waveform signals to a plurality of transducers, an improved signal, such as improved signal beamforming properties, can be obtained at the receiving transducer. By providing a plurality of receiving measurement transducers, the received measurement signal can be evaluated more efficiently and/or greater design freedom can be achieved for arbitrary waveform measurement signals.

Fig. 6-9 are aligned so that the force of gravity acting on the liquid in pipe 12 is directed downward. However, arrangements that are rotated relative to the arrangements shown in FIGS. 6-9. The direction of observation in Fig. 6-9 extends along the longitudinal axis of pipe 12. The position upstream or downstream of the transducer in FIG. 6-9 is not specified.

In the arrangement of FIG. 6, an array of five piezoelectric elements 31-35 is provided at a first location, and an additional piezoelectric element 36 is located upstream or downstream of the first location. If an array of five elements 31-35 is used as a transmitter and an additional element 36 is used as a receiver, the array of piezoelectric elements 31-35 can be used to obtain a predetermined wave front and to achieve improved focusing of the acoustic wave in a predetermined direction.

In the arrangement of FIG. 7, one piezoelectric element 37 is provided at the first location, and an array of five piezoelectric elements 38-42 is located upstream or downstream of the first location. An array of piezoelectric elements 38-42 can be used to obtain an improved wavefront recording of the response signal. This improved recording can then be used to produce an improved flow measurement signal, which is then applied to the single piezoelectric element 37.

In FIG. 8 shows an arrangement of two piezoelectric elements 43, 44 with one element downstream of the other. The distance d from the connection line between the piezoelectric elements 43, 44 to the axis of symmetry of the pipe 12 is approximately half the radius of the pipe 12, so that the flow layer can be measured at a distance d to the central axis of the pipe 12.

Especially for fixed transducers such as the piezoelectric elements 43, 44 shown in FIG. 8, the flow measurement according to the present invention provides an improved signal at the receiving piezoelectric element 44, 43 due to beamforming.

In FIG. 9 shows an arrangement of eight piezoelectric elements 45-52 that are 45 degrees apart from each other. With respect to upstream and downstream arrangements, several arrangements are possible.

In one arrangement, sensor locations alternate around the perimeter between upstream and downstream placement, for example, 45, 47, 49, 51 upstream and 46, 48, 50, 52 downstream.

In another arrangement, the first four in-order perimeter elements, such as 45-48, are located upstream or downstream of the other four elements, such as 49-52. In yet another arrangement with 16 piezoelectric elements, all of the piezoelectric elements 45-52 of FIG. 9 are in the same plane, and the arrangement of FIG. 9 is repeated in the upstream or downstream direction.

In the arrangements of FIG. 6-9, the receiving transducer is offset from the sending transducer with respect to the pipeline longitudinal or flow direction. In particular, the transducer 36 of FIG. 6 is offset from the transducers 31-35, the transducer 37 of FIG. 7 is offset from transducers 38-42, transducer 44 of FIG. 8 is offset from transducer 43, and in FIG. 9, opposite measuring transducers are offset relative to each other in the longitudinal direction of the pipeline. For example, transducer 51 is offset from transducer 47 and transducer 52 is offset from transducer 46.

In FIG. 10 shows, by way of example, a flow measurement device 60 for measuring flow in the arrangement of FIG. 1 or other arrangements according to the present invention. In the arrangement of FIG. 1, the flow measurement device 60 is provided with first and second calculation units 15 and 16 (not shown in FIG. 10).

The first connector 61 of the flow measuring device 60 is connected to the first piezoelectric transducer 11 in the fluid conduit 12, and the second connector 62 of the flow measuring device is connected to the second piezoelectric transducer 13 in the fluid conduit 12.

Inside the flow measurement device 60, a first connector 61 is connected to an A/D converter 64 via a multiplexer 63 and a first amplifier 74. A second connector 62 is connected to a digital buffer 67 via a second amplifier 75 and a demultiplexer 66.

Buffer 67 is connected to a waveform generator 69 which is connected to a waveform database 70 . The waveform database 70 is connected to the ADC 64 via a matching module 68, whereby the matching module 68 is provided to match the parameters to the specific conditions of the pipeline 12 under test. device 72 for results.

During the waveform generation phase, the waveform generator 69 extracts the waveform parameters from the waveform database 70, obtains an electrical signal based on the extracted waveform parameters, and sends the waveform to the digital buffer 67.

More specifically, the flow measurement device 60 may be provided with a processing electronics 53 in the form of a microcontroller or FPGA, which includes, among other things, several high resolution ADCs 64, a digital correlator (not shown in FIG. 10), and a module supporting signal envelope detection. . In the transmission direction, the flow measurement device 60 generates digital signals with a variable frequency. In the case of a square wave signal, the signal duty cycle can be changed by modulating the pulse width of the square wave signal, also known as PWM.

The ADC 64, the matching module 68, the waveform database 70, the waveform generator 69 and the buffer 67 are provided, for example, in the electronic processing unit 53, for example, a microcontroller or a microprocessor. Among other things, the electronic processing unit 53 includes a connector pin 54 for connecting to a first amplifier 74, a connector pin 55 for connecting to a second amplifier 75, a connector pin (not shown) for connecting to a power battery (not shown), and a connector pin (not shown) for connection to earth potential (not shown).

Additional components may be provided between the battery and the connector pin, not shown in FIG. 10, such as the power converter.

In an alternative embodiment, it is possible to measure the "current fingerprint" of the electronic processing unit 53 while executing the proposed methods. It has been found that the computational part of the methods is not in the field of view (which could be a pronounced current fingerprint), but rather in flat areas following the field of view.

The flow measurement device 60 may transmit signals at a predetermined frequency and amplitude. These signals are output by digital buffer 67. Alternatively, the output may be provided by a component that converts a predetermined output signal to a PWM signal.

The device according to Fig. 10 is shown for illustration purposes. The device for performing flow measurement according to the present invention may contain more or fewer components than shown in FIG. 10. In particular, the digital signals according to the DTRAF and DTRAC methods mentioned below can be generated by the variable pulse generator unit 76 and do not require a matching module 68 and/or a waveform database 70 and/or a waveform generator 69.

In the case of a PWM generated time reversal signal, the functionality is provided to reverse the time of the received response signal, or a portion of the received response signal, and send the reversal signal as an input to the pulse width modulator.

The procedures of the following FIG. 11-13 can be used to generate specially adapted digital output signals for use in the measurement according to the present invention. Among other things, the phase and amplitude of the digital signal may change.

In FIG. 11 depicts an iterative procedure for generating an ultrasonic output matching predetermined criteria.

In the first step 80, the pulse generator generates a measurement signal. In the second step 81, the measurement signal is applied to the first measurement transducer 11. The response signal is measured in the second measurement transducer 13 in step 82.

In the next step 83, the response signal is evaluated according to predetermined criteria. For example, the response signal may be consistent with a predetermined waveform. If it is determined in step 84 that the response signal matches the predetermined criteria, the measurement signal parameters are determined and stored in step 86, preferably in the waveform database 70 for later use.

Otherwise, the frequency, amplitude and/or phase of the measurement signal, or also other signal parameters, are adjusted in step 85, and the procedure is looped back to the first step 80, in which a predetermined measurement signal is generated.

This iterative method can also be applied to a multi-transmitter arrangement. In the case of a plurality of sending measuring transducers, the individual frequencies, amplitudes or phases of the respective measuring signals are adjusted. In the case of multiple transducers, the criteria apply to the response signals received at the receiving transducers.

In FIG. 12 shows an additional method for obtaining an arbitrary waveform, which includes the following steps.

In step 90, the transducer bandwidth is measured. In one example, the transducer bandwidth is approximately 300 kHz around the transducer center frequency of approximately 1 MHz. One example of a signal for which this transducer bandwidth can be effectively used is a signal having a rectangular bandwidth of 300 kHz in the frequency domain.

A band-limited function such as a square wave is generated in the frequency domain. The corresponding time domain signal or function is obtained in step 92 by applying an inverse Fourier transform giving a function similar to a cardinal sine.

This cardinal sine-like function is then truncated in step 93 to a suitable signal duration that does not contain excessively high signal strength but contains sufficient information. The signal is then used in step 94 as input to a pulse generator or pulse width modulation to generate a pulse train.

The resulting signal is used as the measurement signal in the upstream and downstream directions in steps 95 and 96.

In FIG. 13 shows an additional method for obtaining a measurement signal. The method according to FIG. 13 is a variant of the method of FIG. 12. For brevity, such steps are not re-explained. According to the method of FIG. 13, the function or waveform is adjusted such that there is no offset in the zero fluid flow mode.

In step 105, a test is made as to whether the signal's time offset is below a predetermined threshold. If the offset is below a predetermined threshold value, at step 107 the signal parameters are stored. Otherwise, the procedure returns to the beginning of the loop at step 101.

An additional method (not shown in the figures) is explained below.

1) From the difference between the upstream and downstream measurements, the time-of-flight difference ΔT is obtained.

2) Adjust the frequency of the signal. The procedure is then returned to the beginning of the loop in step 1), and the amplitude and phase of the signal in the time domain are changed until zero offset is reached, which implies that for zero fluid flow, the time difference is measured, which is zero.

3) Alternatively, the signal of step 1) is generated in step 2) of the correction. According to predefined criteria, pre-emphasis is applied to the signal. The pre-emphasis may be chosen such that the receiver may be configured to be suitable for the particular received response signal. For example, the signals in steps 1) and 2) may be pre-emphasized so that the zero-crossings at the receiver occur at equally spaced time intervals. Thus, a narrowband receiver can be used. This applies to time of flight and TRA measurements.

Steps 1-3 use a digital arbitrary waveform instead of a simple square wave. The above procedure and the procedures of FIG. 11-13 can be used for all digital systems that are based on time-of-flight measurements and use time-reversed acoustics.

Flow measurements using the DTRAF method, using the DTRAC method, and using a time-reversed signal applied in pulse-width modulation are explained below in more detail.

Digital filtering using TRA (DTRAF) is a digital version of the known method using TRA. The basic idea of DTRAF is not to send a time-reversed signal and instead replace it with a digital filtering operation, however, without changing the underlying principles, it is also possible to send a time-reversed signal in combination with DTRAF (or DTRAC). Accordingly, in the DTRAF learning process, a digital FIR filter is obtained in which the TRA method is digitally executed. This simplifies the electronics in terms of TDC removal and lower specification DACs, for example in terms of accuracy and power consumption. On the contrary, an important component is the ADC. Ultimately, interpolation of the processed signals improves accuracy from the sampling period to the picosecond domain.

The proposed DTRAF method is shown in block diagram form in FIG. nineteen.

In the arrangement of FIG. 19 the layout digital buffer 67 of FIG. 10 has been replaced by an adjustable pulse generator unit (APGU) 76. This method is explained below in more detail.

в режиме обучения:First of all, either with the help of a high-precision or with the help of an on-board ADC, a learning process is carried out using the TRA for the flow measurement device 60 . The training output is a digital FIR filter. The system is excited by the input signal

in learning mode:

(1)

(one)

(2)

(2)

- период выборки. Этот период выборки должен согласовываться с периодом выборки последующих измерений. В случае использования более быстрого ADC, цифровой фильтр необходимо подвергнуть перевыборке до равенства периоду выборки режима измерения.where

- sampling period. This sampling period shall be consistent with the sampling period of subsequent measurements. If a faster ADC is used, the digital filter must be resampled to equal the sampling period of the measurement mode.

:In measuring mode, the system is again excited by the input signal

:

(3)

(3)

(4)

(4)

и

- оцифрованные измерительные сигналы. Здесь и далее переменная в квадратных скобках обозначает дискретный временной индекс, соответствующий n-кратному времени выборки Ts, в случае равномерной выборки, которая начинается в момент времени t = 0.

and

- digitized measuring signals. Here and below, the variable in square brackets denotes the discrete time index corresponding to n times the sampling time Ts, in the case of a uniform sampling that starts at time t = 0.

In the next step, these digitized measurement signals are convolved with time-reversed trained digital filters:

(5)

(5)

(6)

(6)

This additional filtering operation adds to the computational time due to the many multiplications and additions. It should be noted here that for both convolutions it may be advantageous to use the same digital filter banks instead of two different trained digital filter banks 120, 121 . Time reversal can be carried out in the memory by storing in reverse order or also by reading in reverse order.

и

подвергают корреляции (или согласованию с помощью какого-либо сопоставителя) друг с другом в блоке 122 взаимной корреляции:These two filtered measuring signals are then

and

subjected to correlation (or matching using any matcher) with each other in block 122 cross-correlation:

(7)

(7)

It is important to note that this cross-correlation by the cross-correlation block 122 does not need to be fully calculated. In particular, the correlation function need not be sampled at such a high frequency that the maximum value is always close to the sampling time. Cross-correlation is an estimator for time delays, and its maximum argument value corresponds to the time delay between two measurement signals:

(8)

(eight)

where n is the time index and r is the correlation function between two measurement signals y ₁₂ [n], y ₂₁ [n].

The time delay between the upstream and downstream signal is twice the time delay between the signal in zero flow mode and the signal at the current flow rate.

и

. Вычитание этих значений времени переноса по определению дает:However, in order to calculate not the entire cross-correlation, it is sufficient to have an estimate in which the maximum value of the argument is located. This can be done using estimates of the absolute values of the transfer time

and

. Subtracting these transfer times by definition gives:

(9)

(nine)

:At low sampling rates, this is exactly enough to estimate

:

(10)

(ten)

обозначает оценку

. В геометрическом смысле

представляет собой запаздывание в единицах целочисленных отсчетов между двумя подвергнутыми фильтрации измерительными сигналами

и

.where

denotes an estimate

. In a geometric sense

is the delay, in units of integer samples, between two filtered measurement signals

and

.

. Количество точек зависит от способа интерполяции. Способ интерполяции повышает точность от периода выборки вплоть до пикосекундного диапазона. В качестве примера, для таких сигналов ультразвукового измерительного преобразователя со столь небольшой шириной полосы пропускания превосходно действует косинусная интерполяция (требует трех точек). Косинусная интерполяция пиков дает расстояние

от запаздывания

до скрытого максимального значения непрерывной косинусной функции. Эта интерполяция изображена на фиг. 20.Finally, it is necessary to calculate only a few cross-correlation points (or any matcher) in the vicinity of the delay

. The number of points depends on the interpolation method. The interpolation method improves the accuracy from the sampling period up to the picosecond range. As an example, for such ultrasonic transducer signals with such a small bandwidth, cosine interpolation (requires three points) performs excellently. Cosine interpolation of the peaks gives the distance

from delay

up to the hidden maximum value of the continuous cosine function. This interpolation is shown in Fig. 20.

From: F. Viola, W. Walker, "A Spline-Based Algorithm for Continuous Time-Delay Estimation Using Sampled Data", IEEE transactions on ultrasonics, ferroelectrics and frequency control, Vol. 52, no. 1, January 2005; it is known that the cosine interpolation of the peaks of the function y[k] can be calculated as follows:

(11)

(eleven)

(12)

(12)

(13)

(thirteen)

затем вносят поправку в оценку запаздывания

для получения

с высокой точностью:Via

then correct the delay estimate

to receive

with high precision:

(14)

(fourteen)

The multiplication by ½ can be omitted when considering the flow rate in the final calculation.

There is a method slightly different from DTRAF (Digital Filtering Using TRA) called DTRAC (Digital Correlation Using TRA). The training and measurement processes for obtaining trained FIR filters and measurement signals are identical to DTRF. However, the signal processing chain is changed. Instead of filtering the entire signal with (time-reversed) trained FIR filters, it is only necessary to compute a few points by cross-correlation with non-time-reversed FIR filters. This is done for the measuring signal upstream and downstream.

путем вычитания. Пример способа DTRAC изображен на фиг. 21 в виде структурной схемы.By interpolating these correlation points, exact absolute transfer times are obtained, which can then be used to calculate

by subtraction. An example of the DTAC method is shown in FIG. 21 in block diagram form.

Similar to the DTRAF method, variable pulse generator block 76 can be used to perform DTRAC. Below, the proposed DTRAC method is explained in more detail.

, тогда как

- непрерывная во времени форма

:First of all, training using TRA is carried out in the same way as for DTRAF, using the input signal

, whereas

- time-continuous form

:

(15)

(fifteen)

(16)

(sixteen)

и

, где

- период выборки.This gives the FIR filter coefficients

and

, where

- sampling period.

:This measurement works in the same way as when training using TRA. Use the same input signal

:

(17)

(17)

(18)

(eighteen)

. Затем эти измерительные сигналы подвергаются взаимной корреляции с помощью цифровых фильтров:The measurement signals are quantized and sampled using

. These measurement signals are then cross-correlated using digital filters:

(19)

(nineteen)

(20)

(20)

Cross-correlation is an estimate function for time delays. However, other time delay estimators (or matchers) may be used, such as, for example, sum of squared difference (SSD). The two cross-correlations do not have to be computed completely, instead only a few points in the vicinity of the real maximum peak need to be computed. This is enough to calculate the interpolation method:

(21)

(21)

(22)

(22)

для низких частот выборки:Or, more generally, estimates

for low sampling rates:

(23)

(23)

(24)

(24)

соответствует оценке

. Большой период выборки снижает риск неверной оценки

или

. В способе интерполяции используется значение непрерывной во времени функции в основе дискретной во времени взаимной корреляции (или какого-либо сопоставителя). В случае узкополосных ультразвуковых измерительных преобразователей вывод взаимной корреляции, как правило, весьма подобен косинусному колебанию в окрестности реального максимального пикового значения. Данный эффект может быть даже больше в случае входного сигнала, который возбуждает только резонансную частоту пары ультразвуковых измерительных преобразователей, подобно длинному синусному, косинусному, треугольному или прямоугольному колебанию. В способе интерполяции оценивают индекс реального максимального значения дискретной во времени взаимной корреляции в подвыборках (и значений времени, кратных периоду выборки):

corresponds to the assessment

. Large sampling period reduces the risk of misestimation

or

. The interpolation method uses the value of a time-continuous function based on a time-discrete cross-correlation (or some matcher). In the case of narrowband ultrasonic transducers, the cross-correlation output is typically quite similar to a cosine waveform around the actual maximum peak value. This effect can be even greater for an input signal that only drives the resonant frequency of a pair of ultrasonic transducers, like a long sine, cosine, triangular, or square wave. In the interpolation method, the index of the real maximum value of the time-discrete cross-correlation in subsamples (and time values that are multiples of the sample period) is estimated:

(25)

(25)

(26)

(26)

и

соответствует коррекции подвыборки способа интерполяции. Так как абсолютные значения времени переноса

и

являются чрезвычайно точными,

можно получить путем вычитания следующего:

and

corresponds to the subsampling correction of the interpolation method. Since the absolute values of the transfer time

and

are extremely accurate

can be obtained by subtracting the following:

(27)

(27)

The multiplication by ½ can be omitted when considering the flow rate in the calculation.

The proposed methods DTRAF and DTRAC can simulate on a computer channel 77 with conduit 12 and fluid. This can be explained as follows: if an LTI system contains blocks or transfer functions A, B, C, D, then the resulting transfer function in the frequency domain is the product A*B*C*D of the individual transfer functions A, B, C, D This product does not depend on the order of the transfer functions A, B, C, D. Therefore, the calculation of the channel response can be "shifted" into the digital domain.

In FIG. 14 shows a PWM signal 129 of a pulse width modulator having a suitably modulated on and off cycle such that the PWM signal 129 approximates a time reversed TRA signal 130.

In FIG. 15 shows a comparison of the response signal 131 with the PWM signal 129 and the response signal 132 with the time reversed TRA signal 130.

According to the PWM-based TRA method, the time-reversed signal is approximated as:

U _A = A sign(U _E -U _D ), where sign(x) = 1 for x >= 0 and 0 otherwise.

As an example, Pulse Width Modulation can be based on a transducer example with a triangular signal frequency of 2 MHz and a PWM time resolution of approximately 83 ns.

An approximation of the time-reversed signal shown in FIG. 14 is quite satisfactory even if PWM generation is kept simple and additional filters are not used.

The PWM-based correlated TRA simulation shown in FIG. 16 shows how the behavior of ΔT changes. Since signal strength is computed for simulation and noise is scaled, the SNR for PWM-based correlated TRA should be increased in this simulation.

The results in FIG. 16 and 17 are obtained by simulation in which the system including signal generation, transducers, channel and signal reception is modeled as a linear stationary system. The simulation simulates measurements in zero flow mode.

In FIG. 16 shows measured time delays for PWM-based correlated TRA. The measured time delays are indicated by a histogram 133 and the resulting probability density is shown by a Gaussian curve 134 .

In FIG. 17 shows measurement results for signals generated by the above digital correlation method using TRA. The measured time delays are indicated by a histogram 135 and the resulting probability density is shown by a Gaussian curve 136 .

In FIG. 18 shows measurement results for signals generated by the above digital filtering method using TRA. The measured time delays are indicated by a histogram 135 and the resulting probability density is shown by a Gaussian curve 136 .

Comparison of the methods with each other gives relatively similar results (for example, in relation to the mean and standard deviation). It should be noted that the simulation parameters used in generating the measurement results shown in FIG. 16, 17 and 18 are different.

For the measurement in FIG. 18, pulse trains containing 25 pulses are used as input. The first 50 TRA training measurements are averaged and stored in two FIR filters. ΔT_DTRAF is the result of a digital filtering method using TRA (DTRAF). For measurements in zero flow mode, a test tube with closed ends was used.

In FIG. 19 shows a visualization of the DTRAF method in the form of a block diagram. Here, the adjustable pulse generator unit 76 is connected to the input signal source on the input side and to the first ultrasonic transducer 11 on the output side. The first ultrasonic transducer 11 is connected to a conduit 77, such as a fluid conduit 12. The second ultrasonic transducer 13 is connected to channel 77 and the output of the second ultrasonic transducer 13 is connected to ADC 64. A similar arrangement is also used in the measurement phase shown in FIG. 19 under the learning phase.

The bottom line shows the evaluation of the signal, which is the digitized received response signal, in the measurement phase. The first digital filter block 120 and the second digital filter block 121 are respectively connected to the output of the ADC 64. The respective outputs of the digital filter blocks 120, 121 are connected to the input of the cross-correlation block 122, and the output of the cross-correlation block 122 is connected to the input of the interpolation block 123.

In FIG. 20 shows an image of the overall interpolation of the peaks to determine the maximum value of the correlation function. The overall peak interpolation according to FIG. 20 is further explained above and is not repeated here in detail. Briefly, the correlation function is locally approximated by a cosine function or a higher order polynomial. This cosine is found by interpolation with three interpolation nodes y[k-1], y[k] and y[k+1]. Then the maximum value of the cosine function is determined. If more interpolation nodes are available, you can use the least squares method to find the cosine. The distance from the interpolation nodes to the cosine can also be minimized according to another type of normalization.

In FIG. 21 shows a visualization of the proposed DTRAC method in the form of a block diagram. The arrangement according to Fig. 21 is similar to the arrangement of FIG. 19. For brevity, only the components in the bottom row, which differ from the components of FIG. nineteen.

The input of cross correlation block 122 is connected to the output of ADC 64, and the output of cross correlation block 122 is connected to the input of interpolation block 123.

Fig. 19 and 21 can also be considered as a possible implementation of the DTRAC and DTRAF methods by hardware and/or software components. The various blocks shown in Fig. 19 and 21 may be implemented in separate components or in a single component, and they may be implemented in hardware and/or software. For example, block 122 cross-correlation and block 123 interpolation can be implemented on the same integrated circuit. The training and measurement procedures according to FIG. 19-21 have been further explained above and will not be repeated here.

In FIG. 22 is an example of a further embodiment of the proposed flow measurement device 60' for measuring the flow rate of fluid in fluid conduit 12 with an ultrasonic transit time based ultrasonic flow meter 60' using a digital filtering technique using time reversed acoustics.

The buffer 67 is connected to the block 76 of the adjustable pulse generator. ADC 64 is additionally connected to digital filter blocks 120, 121. Digital filter blocks 120, 121 are connected to a correlation block 122, which is connected to an interpolation block 123. The interpolation block 123 is connected to the speed calculation block 71, which is connected to both the result memory 72 and the buffer 67.

LIST OF LINKS

10 flow meter layout

11 piezoelectric element upstream

12 pipeline

13 downstream piezoelectric element

14 middle flow direction

15 first calculation block

16 second calculation block

17 signal path

20 signal path

22 piezoelectric element

23 piezoelectric element

31-52 piezoelectric elements

53 electronic processing unit

54 connector pin

55 connector pin

56 connector pin

60, 60' flow measuring device

61 first connector

62 second connector

63 multiplexer

64 AD

66 demultiplexer

67 digital buffer

68 matching module

69 waveform generator

70 waveform database

71 speed calculation blocks

72 result memory

73 measuring signal generator

74 first amplifier

75 second amplifier

76 adjustable pulse generator block

77 channel

90-96 method steps

100-106 method steps

110-117 method steps

120 1st ^digital filter

¹²¹ 2nd digital filter

122 block correlation

123 block interpolation

129 PWM signal

130 TRA signal

131 PWM signal response signal

132 TRA signal response signal

133 measured time delays

134 probability density

135 measured time delays

136 probability density

137 measured time delays

138 probability density

Claims (135)

1. A method for determining the fluid flow rate in a channel (77) with a pipeline (12) for a fluid medium using an ultrasonic flow meter (60), based on the measurement of transit time, using a digital filtering method using a time-reversed acoustic signal, and the method includes the following steps: - execution of the learning process using the block (76) of the adjustable pulse generator, and the learning process includes the steps: - supplying an input signal (x[n]) to the first ultrasonic transducer (11), wherein the first ultrasonic transducer (11) is installed on the pipeline (12) for the fluid at the first location; - receiving the first signal response to the input signal (x[n]) in the second ultrasonic transducer (13), wherein the second ultrasonic transducer (13) is installed on the conduit (12) for the fluid at a second location, the second position being upstream upstream or downstream from the first location relative to the direction (14) of fluid flow; - converting the first response signal into a first digitized response signal using an analog-to-digital converter (64); and - obtaining a first digital filter (120) with a response based on the first digitized response signal; - supplying the input signal (x[n]) to the second ultrasonic measuring transducer (13); - receiving a second response signal to the input signal (x[n]) in the first ultrasonic measuring transducer (11); - converting the second response signal into a second digitized response signal using an analog-to-digital converter (64); and - obtaining a second digital filter (121) with a response based on the second digitized response signal; - execution of the measurement process using the block (76) of the adjustable pulse generator, and the measurement process includes the following steps: - supplying the input signal (x[n]) to the first ultrasonic measuring transducer (11); - receiving a third response signal to the input signal (x[n]) in the second ultrasonic measuring transducer (13); - converting the response signal into a third digitized response signal (y ₁₂ ´[n]) using an analog-to-digital converter (64); - execution of the measurement process in the reverse direction using the block (76) of the adjustable pulse generator, and the measurement process in the reverse direction includes the steps: - supplying the input signal (x[n]) to the second ultrasonic measuring transducer (13); - receiving a fourth response signal to the input signal (x[n]) in the first ultrasonic measuring transducer (11); - converting the fourth response signal into a fourth digitized reverse direction response signal (y ₂₁ ´[n]) using an analog-to-digital converter (64); - obtaining the first input correlation signal (y ₁₂ [n]) based on the third digitized response signal (y ₁₂ ´[n]) and the first digital filter (120) with the response, and the obtaining includes reversal in time of either the third digitized response signal (y ₁₂ ´[n]), or the first digital filter (120) with a response; - obtaining the second input correlation signal (y ₂₁ [n]) based on the fourth digitized response signal (y ₂₁ ´[n]) and the second digital filter (121) with the response, and the obtaining includes reversal in time or the fourth digitized response signal (y ₂₁ ´[n]), or the second digital filter (121) with a response; and - obtaining the time-of-flight difference (ΔT) based on the first correlation input (y ₁₂ [n]) and the second correlation input (y ₂₁ [n]) by calculating the discrete correlation of the first correlation input (y ₁₂ [n]) with the second input signal of the correlation (y ₂₁ [n]), while determining the time index of the maximum value of the discrete correlation using the method of interpolation. 2. The method according to claim. 1, characterized in that the second digital filter (121) with a response is equal to the first digital filter (120) with a response. 3. The method according to claim 1, wherein the second response digital filter (121) is a reverse response digital filter, wherein the determination of the reverse response filter includes the following steps: - execution of the learning process in the opposite direction using the block (76) of the adjustable pulse generator; - feeding the input signal of the opposite direction (x[n]) to the second ultrasonic measuring transducer (13); - receiving a reverse direction response signal to the reverse direction training signal in the first ultrasonic measuring transducer (11); - converting the reverse direction response signal into a digitized reverse direction response signal using an analog-to-digital converter (64); - obtaining the digital filter coefficients with a reverse direction response (h _MF,21 [n]) based on the digitized reverse direction response signal; and - obtaining a second digital filter (121) with a response based on the digital filter coefficients with a reverse direction response (h _MF,21 [n]). 4. A method for determining the fluid flow rate in a channel (77) with a pipeline (12) for a fluid medium using an ultrasonic flow meter (60), based on the measurement of transit time, using a digital filtering method using a time-reversed acoustic signal, and the method includes steps: - execution of the learning process using the block (76) of the adjustable pulse generator, and the learning process includes the steps: - supplying an input signal (x[n]) to the first ultrasonic transducer (11), wherein the first ultrasonic transducer (11) is installed on the pipeline (12) for the fluid at the first location; - receiving the first signal response to the input signal (x[n]) in the second ultrasonic transducer (13), wherein the second ultrasonic transducer (13) is installed on the conduit (12) for the fluid at a second location, the second location being upstream upstream or downstream of the first location relative to the direction of fluid flow; - converting the first response signal into a first digitized response signal using an analog-to-digital converter (64); - determining a first digital response filter having filter coefficients (h _MF,12 [n]) based on the first digitized response signal; - supplying the input signal (x[n]) to the second ultrasonic measuring transducer (13); - receiving the second response signal in the first ultrasonic transducer (11); - converting the second response signal into a second digitized response signal using an analog-to-digital converter (64); and - determining a second digital response filter having filter coefficients (h _MF,21 [n]) based on the second digitized response signal; - execution of the measurement process using the block (76) of the adjustable pulse generator, and the measurement process includes the steps: - supplying the input signal (x[n]) to the first ultrasonic measuring transducer (11); - receiving a third response signal to the input signal (x[n]) in the second ultrasonic measuring transducer (13); - converting the third response signal into a third digitized response signal (y ₁₂ ´[n]) using an analog-to-digital converter (64); - execution of the measurement process in the reverse direction using the block (76) of the adjustable pulse generator, and the measurement process in the reverse direction includes the steps: - supplying the input signal (x[n]) to the second ultrasonic measuring transducer (13); - receiving a fourth response signal to the input signal (x[n]) in the first ultrasonic measuring transducer (11); - converting the fourth response signal into a fourth digitized response signal (y ₂₁ ´[n]) using an analog-to-digital converter (64); - obtaining the first value of the time-of-flight (T _abs,12 ) based on the third digitized response signal (y ₁₂ ´[n]) and the first digitized response signal by calculating the discrete correlation of the third digitized response signal (y ₁₂ ´[n]) with the first digitized the response signal, while determining the time index of the maximum value of the discrete correlation using the method of interpolation; - obtaining the second value of the time-of-flight (T _abs,21 ) based on the fourth digitized response signal (y ₂₁ ´[n]) and the second digitized response signal by calculating the discrete correlation of the fourth digitized response signal (y ₂₁ ´[n]) with the second digitized the response signal, while determining the time index of the maximum value of the discrete correlation using the method of interpolation; and - obtaining the difference in the values of the time of flight (ΔT) by subtracting the previously obtained values of the time of flight (T _abs,12 , T _abs,21 ). 5. The method of claim 4, wherein the second response digital filter is equal to the first response digital filter. 6. The method of claim 4, wherein the first digital response filter is a forward response filter; wherein the second digital response filter is a reverse response filter; wherein the coefficients of the filter (h _MF,12 ) representing the filter with a forward direction response are obtained based on the first digitized response signal; and wherein the filter coefficients (h _MF,21 ) representing the filter with a reverse direction response are obtained based on the second digitized response signal. 7. A method for determining the fluid flow rate in a fluid conduit (12) using an ultrasonic flow meter based on transit time measurement, the method comprising the steps of: - generating a training signal using a block (76) adjustable pulse generator; - supplying a training signal to the first ultrasonic measuring transducer (11), wherein the first ultrasonic measuring transducer (11) is installed on the pipeline (12) for the fluid medium in the first position; - receiving a response signal to the training signal in the second ultrasonic measuring transducer (13), wherein the second ultrasonic measuring transducer (13) is installed on the pipeline (12) for the fluid in the second position, the second position being upstream or downstream of the first positions relative to the direction of fluid flow; - converting the response signal into a digitized response signal using an analog-to-digital converter (64); - time reversal of the digitized response signal in order to obtain an inverted digitized response signal; - generating a measuring signal with pulse-width modulation based on the reversed digitized response signal; - supply of a measuring signal with pulse-width modulation to the first ultrasonic measuring transducer (11); - receiving a response signal to the measuring signal with pulse-width modulation in the second ultrasonic measuring transducer (13); and - obtaining the time of flight (ΔT) based on the response signal. 8. An electronic processing unit (53) for determining the fluid flow rate in the channel (77) with the pipeline (12) for fluid using an ultrasonic flow meter (60) based on the measurement of transit time, using a digital filtering method using reversed acoustic signal time, containing: - a block (76) of an adjustable pulse generator, and the block (76) of the pulse generator is configured to execute the learning process using the input signal (x[n]); - transmission means, the transmission means being configured to supply an input signal (x[n]) to the first ultrasonic transducer (11), the first ultrasonic transducer (11) being mounted on the fluid conduit (12) at a first location; - receiving means, wherein the receiving means are configured to receive the first signal response to the input signal (x[n]) in the second ultrasonic measuring transducer (13), the second ultrasonic measuring transducer (13) being installed on the pipeline (12) for the fluid in a second location, the second location being upstream or downstream of the first location relative to the direction of fluid flow; - an analog-to-digital converter (64), wherein the analog-to-digital converter (64) is configured to convert the first response signal into a first digitized response signal; processing means, the processing means being configured to determine a first digital response filter (120) having FIR filter coefficients ( _{h_MF12} [n]) based on the digitized response signal; - wherein the transmission means is further configured to supply an input signal (x[n]) to the second ultrasonic transducer (13), the second ultrasonic transducer (13) being mounted on the fluid conduit (12) at a second location; - moreover, the receiving means is additionally configured to receive a second response signal to the input signal (x[n]) in the first ultrasonic measuring transducer (11); - moreover, the analog-to-digital converter (64) is further configured to convert the second response signal into a second digitized response signal; wherein the processing means is further configured to determine a second digital response filter (121) having FIR filter coefficients (h _MF,21 [n]) based on the second digitized response signal; - moreover, the block (76) of the adjustable pulse generator is additionally configured to execute the measurement process using the input signal (x[n]); - moreover, the transmission means is additionally configured to supply the input signal (x[n]) of the measurement process to the first ultrasonic measuring transducer (11); - moreover, the receiving means is additionally configured to receive a third response signal to the input signal (x[n]) of the measurement process in the second ultrasonic measuring transducer (13); - moreover, the processing means are additionally configured to convert the third response signal into a third digitized response signal (y ₁₂ ´) using an analog-to-digital converter (64); - moreover, the block (76) of the adjustable pulse generator is additionally configured to execute the measurement process in the opposite direction using the input signal (x[n]); - moreover, the transmission means are additionally configured to feed the input signal (x[n]) of the measurement process in the opposite direction to the second ultrasonic measuring transducer (13); - moreover, the receiving means is additionally configured to receive the fourth response signal of the measurement process in the opposite direction in the first ultrasonic measuring transducer (11); - moreover, the analog-to-digital converter (64) is additionally configured to convert the fourth response signal of the measurement process in the opposite direction into the fourth digitized response signal (y ₂₁ ´) using the analog-to-digital converter (64); - moreover, the processing means are additionally configured to: - obtaining the first input correlation signal (y ₁₂ ) based on the third digitized response signal (y ₁₂ ´) and the first digital filter (120) with the response, and the obtaining includes reversal in time of either the third digitized response signal (y ₁₂ ´) or the first digital filter (120) with response; - obtaining a second input correlation signal (y ₂₁ ) based on the fourth digitized response signal (y ₂₁ ´) and a second digital filter (121) with a response, and the obtaining includes reversal in time of either the fourth digitized response signal or the second digital filter (121) with response; - obtaining the time-of-flight difference (ΔT) based on the first correlation input (y ₁₂ [n]) and the second correlation input (y ₂₁ [n]) by computing the discrete correlation of the first correlation input (y ₁₂ [n]), and a second correlation input signal (y ₂₁ [n]); and determining a time index of the maximum value of the discrete correlation using the interpolation method. 9. Electronic processing unit (53) according to claim 8, characterized in that the second digital filter (121) with a response is equal to the first digital filter (120) with a response. 10. The electronic processing unit (53) according to claim 8, characterized in that the second digital response filter (121) is a digital filter with a reverse direction response, and the electronic processing unit (53) is further configured to: - generating a training signal in the opposite direction using the block (76) adjustable pulse generator; - supplying a training signal in the opposite direction to the second ultrasonic measuring transducer (13); - receiving a reverse direction response signal to the reverse direction training signal in the first ultrasonic measuring transducer (11); - converting the reverse direction response signal into a digitized reverse direction response signal using an analog-to-digital converter (64); and - obtaining a second digital filter (121) with a response having FIR filter coefficients (h _MF,21 [n]) based on the digitized reverse direction response signal. 11. An electronic processing unit (53) for determining the fluid flow rate in the channel (77) with the pipeline (12) for the fluid using an ultrasonic flow meter (60) based on the measurement of transit time, using a digital filtering method using reversed the time of the acoustic signal, while the electronic processing unit (53) is configured to: - execution of the learning process using the block (76) of the adjustable pulse generator; - supplying an input signal (x[n]) to the first ultrasonic transducer (11), wherein the first ultrasonic transducer (11) is mounted on the fluid conduit (12) at a first location; - receiving a first signal response to an input signal (x[n]) in a second ultrasonic transducer (13), wherein the second ultrasonic transducer (13) is mounted on the fluid conduit (12) at a second location, the second location being upstream upstream or downstream of the first position relative to the direction of fluid flow; - converting the first response signal into a first digitized response signal using an analog-to-digital converter (64); - obtaining a digital filter with a response having the coefficients of the FIR filter (h _MF,12 [n]), based on the first digitized response signal; - execution of the learning process in the opposite direction using the block (76) of the adjustable pulse generator; - supplying the input signal (x[n]) to the second ultrasonic measuring transducer (13); - receiving a second response signal in the first ultrasonic transducer (11); - converting the second response signal into a second digitized response signal using an analog-to-digital converter (64); - determining a digital response filter having filter coefficients (h _MF,21 [n]) based on the second digitized reverse direction response signal; - execution of the measurement process using the block (76) of the adjustable pulse generator; - supplying the input signal (x[n]) to the first ultrasonic measuring transducer (11); - receiving a third response signal to the input signal (x[n]) in the second ultrasonic measuring transducer (13); - converting the third response signal into a third digitized response signal (y ₁₂ ´[n]) using an analog-to-digital converter (64); - execution of the measurement process in the opposite direction using the block (76) of the adjustable pulse generator; - supplying the input signal (x[n]) to the second ultrasonic measuring transducer (13); - receiving a fourth response signal to the input signal (x[n]) in the first ultrasonic measuring transducer (11); - converting the fourth response signal into a fourth digitized response signal (y ₂₁ ´[n]) using an analog-to-digital converter (64); - deriving a first correlation signal (r ₁₂ ) based on a third digitized response signal (y ₁₂ ´) and a digital response filter having filter coefficients (h _MF,12 [n]) by calculating the convolution, the derivation comprising time reversal, either a third digitized response signal (y ₁₂ ´) or a digital filter having filter coefficients (h _MF,12 [n]); - deriving a second correlation signal (r ₂₁ ) based on a fourth digitized response signal (y ₂₁ ´) and a second digital response filter having filter coefficients (h _MF,21 [n]) by calculating the convolution, the derivation including time reversal either a fourth digitized response signal or a digital filter having filter coefficients (h _MF,21 [n]); - obtaining the values of the time of flight (T _abs,12 , T _abs,21 ) based on each correlation signal (r ₁₂ , r ₂₁ ) using the interpolation method; and - obtaining the difference in the values of the time of flight (ΔT) by subtracting the previously obtained values of the time of flight (T _abs,12 - T _abs,21 ). 12. The electronic processing unit (53) according to claim 11, characterized in that the digital response filter having filter coefficients (h _MF,21 [n]) is equal to the digital response filter having filter coefficients (h _MF,12 [ n]). 13. Electronic processing unit (53) according to claim 12, characterized in that the digital response filter having the filter coefficients (h _MF,21 [n]) is a digital filter with a reverse response, and wherein obtaining a filter with a reverse response includes the steps: - execution of the learning process in the opposite direction using the block (76) of the adjustable pulse generator; - supplying a training signal in the opposite direction to the second ultrasonic measuring transducer (13); - receiving a reverse direction response signal to the reverse direction training signal in the first ultrasonic measuring transducer (11); - converting the reverse direction response signal into a digitized reverse direction response signal using an analog-to-digital converter (64); - obtaining a digital filter with a reverse direction response having FIR filter coefficients (h _MF,21 ), based on the digitized reverse direction response signal. 14. An electronic processing unit (53) for determining the fluid flow rate in the pipeline (12) for a fluid medium using an ultrasonic flow meter based on transit time measurement, wherein the electronic processing unit (53) is configured to: - generating a training signal using a block (76) of an adjustable pulse generator; - supplying an input signal (x[n]) to the first ultrasonic transducer (11), wherein the first ultrasonic transducer (11) is mounted on the fluid conduit (12) at a first location; - receiving a response signal to the training signal in the second ultrasonic transducer (13), wherein the second ultrasonic transducer (13) is installed on the fluid conduit (12) at a second location, the second location being upstream or downstream of the first locations relative to the direction of fluid flow; - converting the response signal into a digitized response signal using an analog-to-digital converter (64); - time reversal of the digitized response signal to obtain an inverted digitized response signal; - execution of the measurement process with pulse-width modulation using an inverted digitized response signal; - supplying a measuring signal with pulse-width modulation to the first ultrasonic measuring transducer (11); - receiving a response signal to the measuring signal with pulse-width modulation in the second ultrasonic measuring transducer (13); and - obtaining the time of flight (ΔT) based on the response signal.

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