CN115218971A - All digital time-of-flight flow meter using time-reversed acoustics - Google Patents

Abstract

A microcontroller and method for determining flow rate using an electronic processing unit of an ultrasonic time-of-flight flow meter using arbitrary waveform signals. The electronic processing unit comprises receiver and transmitter terminals, a signal processing unit and a signal generating unit configured to generate an oscillating electrical output signal having a time-dependent amplitude, wherein the time-dependent amplitude varies in accordance with the stored signal parameter.

Description

All-digital time-of-flight flow meter using time-reversed acoustics

Technical Field

The present application relates to flow meters, and more particularly, to ultrasonic time of flight flow meters.

Background

Currently, various types of flow meters are used to measure the volumetric flow rate of a fluid, such as a liquid or a gas, through a pipe. Ultrasonic flow meters are doppler flow meters that utilize acoustic doppler effects, or time-of-flight flow meters that utilize the difference in travel time caused by the relative motion of the source and the medium (sometimes also referred to as transmission flow meters). The travel time is also referred to as the time of flight or the time of flight.

An ultrasonic time of flight flow meter evaluates the difference in transit time of an ultrasonic pulse propagating in a flow direction and a counter-flow direction. Ultrasonic flow meters are provided as either on-line flow meters, also known as invasive or wet flow meters, or as clamp-on flow meters, also known as non-invasive flow meters. Other forms of flow meters include venturi channels, weirs, radar flow meters, coriolis flow meters, differential pressure flow meters, magnetic induction flow meters, and other types of flow meters.

When there is an irregular flow profile or open channels, more than one propagation path may be required to determine the average flow velocity. Therein, multipath processes are described in hydrological standards such as IEC41 or EN ISO 6416. As a further application, ultrasonic flow meters are also used to measure flow profiles, for example using acoustic doppler flow profilers (ADCP). ADCP is also suitable for measuring water flow rates and discharge in rivers and open waters.

The Theory of operation of Matched Filter based ultrasound Sensing technology and a single chip Implementation platform using (TI) MSP430FR6047 microcontroller was disclosed by "Mathed-Filter Ultrasonic Sensing: theory and Implementation" (White Paper SLAA814-2017, 12.M.).

Disclosure of Invention

It is an object of the present specification to provide an improved time of flight flow meter and corresponding computer implemented method for measuring the average flow velocity or flow profile of a fluid in general, and for liquids and/or gases such as water in particular.

In the flow measuring device according to the present description, an acoustic transducer, also referred to as a piezoelectric transducer, for example in the form of a piezoelectric element, is used to generate and receive a measurement signal.

Alternative acoustic emitters include lasers that excite vibration of a metal film or other light absorbing surface, or coil-driven speakers. According to other embodiments, the flow meter generates pressure waves in other ways, such as by MEMS devices, by using piezoelectric membranes, and the like. The receiver side can also be represented by other means than a piezoelectric transducer but detecting ultrasonic waves.

Although the term "piezoelectric transducer" is often used in this specification, it also stands for other acoustic transducers that generate or detect ultrasonic waves.

The measurement signal according to the present description may be modeled by a matched filter. When the word "signal" is used with reference to a signal manipulation step, it may particularly refer to a representation of a signal in a computer memory.

In particular, the signal representation may be defined by pairs of digitized amplitudes and associated discrete-time values.

An ultrasonic flow meter according to the present description may provide the desired characteristics of the focusing characteristics by using signals of arbitrary shape so that the ultrasonic flow meter obtains signals with the desired characteristics at the receiving transducer or by calculation after the receiving ADC.

As an example, the frequency of the acoustic waves used in the flow meter according to the invention may be between 20kHz and 2MHz, which corresponds to an oscillation period of 0.5 microseconds (μ s), but it may even be as high as 800MHz.

In one aspect, the present specification discloses a microcontroller for determining flow rate with an ultrasonic time-of-flight flow meter using time-reversed digitized signals. In particular, the flow meter may be "all-digital" in the sense that signal evaluation is done by digital signal processing and the electrical signal applied to the ultrasonic transducer is a step signal (which may also be considered a digital signal). That is, the electrical signal has a sequence of discrete voltage levels that is constant over a predetermined sampling time. Therefore, a DAC with an analog part is not required, and the DAC can be replaced by a pulse generator.

The microcontroller can generate any pulse signal. In particular, the microcontroller may be configured to output a pulse train having an arbitrary frequency and length.

A microcontroller according to the present description may comprise a plurality of buffer output terminals, for example four or eight output terminals, instead of only one output terminal. To this end, the microcontroller may comprise a plurality of buffers in a parallel configuration, for example four or eight buffers, which allow time-multiplexed output of all signals to all respective channels or buffers.

That is, the transmitter path or signal generation path in a chip or FGPA, or a path connected to such a chip or FPGA, is operable to generate an arbitrary burst of pulses having an arbitrarily selected pulse length. In this context, "arbitrary" means any within a predetermined range and accuracy, such as within predetermined minimum and maximum frequency ranges and within a predetermined accuracy.

The microcontroller comprises a first receiver terminal for connecting the first ultrasonic transducer, a first transmitter terminal for connecting the first ultrasonic transducer, a second receiver terminal for connecting the second ultrasonic transducer, a second transmitter terminal for connecting the second ultrasonic transducer. The receiver terminal and the transmitter terminal are also referred to as "connectors".

In one embodiment, the first receiver terminal and the first transmitter terminal and the second receiver terminal and the second transmitter terminal coincide. This may be particularly useful when there is no overlapping transmission and reception.

The first and second receiver terminals are connected with a signal processing unit or signal processing means provided for evaluating signals received from the transducer. The signal processing unit includes an analog-to-digital converter (ADC) and an evaluation circuit. Unlike a simple timer circuit, the ADC may provide various options to calculate the shape of the signal.

The predicted signal shape may be used to reduce the sampling rate and/or amplitude resolution of the ADC. In particular, the power saving features may utilize signal shapes according to the present description. The sampling rate and/or the exact amplitude of the amplitude digitization of the ADC may also be non-uniform and time dependent to further reduce power consumption, and the time dependence may depend on the predicted signal shape. Furthermore, the ADC may be turned on only after a specified time after the measurement signal is transmitted, and turned off again after the response signal has been received.

The first and second transmitter terminals are connected to a signal generating unit. The signal generation unit comprises a memory for storing signal parameters and a buffer.

The signal generating unit and the signal processing unit are configured to send the oscillating electrical output signal to the first transmitter terminal and to receive the response signal at the second receiver terminal, and to derive the flow rate of the fluid from at least the received response signal.

In one embodiment, the buffer of the signal generation unit is connected to a Pulse Width Modulation (PWM) unit, which is connected to a low pass filter. Pulse width modulation may provide a simple and robust way to generate an analog electrical signal from a digital signal and then apply the analog electrical signal to an ultrasound transducer, such as a piezoelectric element, a piezoelectric or speaker membrane, a microelectromechanical element, or another type of transducer. By using power electronics, the pulse width modulated output signal can be made sufficiently large that in many cases no additional signal amplification is required. PWM can provide a resulting analog signal of sufficient quality.

The PWM of the microcontroller may also be used with or without weak low-pass filtering after the output of the buffer or PWM to generate an output signal having a staircase shape or a staircase-like shape. To this end, the low-pass filter may be provided as an adjustable or switchable low-pass filter.

The PWM signal stores amplitude information in time information of the signal. The granularity of the time resolution has an effect on the resolution of the PWM signal. To this end, the PWM signal generator may be designed to provide a higher frequency than a comparable ADC, e.g. 10 times as high. The amplitude of the PWM signal generator may be the same or similar to the pulse generator.

In another embodiment, the signal generating unit of the microcontroller comprises a Field Programmable Gate Array (FPGA) connected to a digital input/output connector. The signal generation unit is operative to derive an oscillation signal from an output signal of the FPGA, wherein the output signal of the FPGA is modified by providing programming instructions through the digital input/output connector.

In particular, an FPGA may contain logic or circuit arrangements to derive desired output signals from provided signal parameters. Wherein the signal parameters may be read in from a memory, pre-calculated or calculated in real time. In one embodiment, the calculation of the signal parameters from which the output signal is derived depends on the received signal.

In another embodiment, the signal generation unit is configured to generate a signal string comprising oscillating signal portions separated by predetermined quiet periods in which the output voltage is constant, in particular the output voltage may be zero during the quiet periods. In one embodiment, the oscillating signal portion is repeated at regular intervals. In another embodiment, the duration of the oscillation signal portions and the duration of the time intervals therebetween are determined by calculation. In particular, the calculation of the time interval may comprise a random component or jitter within a predetermined amplitude range, whereby the oscillator signal portions have a randomly varying distance in time from each other.

According to another embodiment, the signal generation of the oscillating signal comprises applying a pulse width modulation and applying a low pass filter to the pulse width modulated output signal.

In yet another embodiment, the signal generation of the oscillating signal comprises generation of a signal train comprising parts of the oscillating signal separated by predetermined quiet periods in which the output voltage is constant, wherein the constant voltage level may be zero.

During measurement of the flow rate, the microcontroller applies a second measurement signal to one of the transducers, receives a response signal to the second measurement signal at the other of the transducers, and processes the response signal to derive the flow rate.

According to one aspect, the present specification discloses a computer-implemented method for determining the flow rate of a fluid in a fluid conduit or channel, in particular a flow rate of a fluid in a pipe or tube, using a transit time ultrasonic flow meter. In a preferred embodiment, "computer-implemented" refers to execution on small-scale electronic components, such as microprocessors, ASICs, FPGAs, etc., which may be used in portable or compact stationary digital signal processing devices, which typically have smaller dimensions than workstations or mainframe computers, and which may be placed at desired locations along a fluid conduit.

Hereinafter, the terms "channel", "conduit", "passageway", and the like are used synonymously. The subject matter of the present application can be applied to all types of conduits for fluids, irrespective of their respective shape, and irrespective of whether they are open or closed. The subject matter of the present application can also be applied to all types of fluids or gases, whether they are gases or liquids, or mixtures of both.

Throughout this application, the term "computer" is often used. Although the computer includes devices such as a laptop computer or a desktop computer, signal transmission and reception may also be accomplished by a microcontroller, an Application Specific Integrated Circuit (ASIC), an FPGA, or the like.

Furthermore, the connecting lines between the transducers may be offset with respect to the centre of the fluid conduit in order to obtain a flow velocity in a predetermined layer, and more than one pair of transducers may be present. Furthermore, the measurement signal may be provided by more than one transducer and/or the response signal to the measurement signal may be measured by more than one transducer.

Further, the present specification discloses a flow measurement device having a first piezoelectric transducer connected to a first connector and having a second ultrasonic transducer, such as a piezoelectric transducer, connected to a second connector. In particular, ultrasonic transducers, such as piezoelectric transducers, may be provided with attachment regions, such as clamping mechanisms for attaching them to a pipe.

Further, the present specification discloses a flow measuring device having a pipe portion. A first ultrasonic transducer, such as a piezoelectric transducer, is mounted to the pipe portion at a first location, and a second ultrasonic transducer, such as a piezoelectric transducer, is mounted to the pipe portion at a second location. In particular, the transducer may be clamped to the pipe portion. Providing the device with a conduit portion may provide benefits when the device is pre-calibrated relative to the conduit portion.

The device can be made compact and portable. The portable device according to the present description, which is equipped with a surface-mountable transducer, such as a clamp transducer, can be used for inspecting pipes in any accessible position. Generally, the device may be stationary or portable. Preferably, the device is compact enough to be placed in a desired location and is sufficiently protected from environmental conditions, such as humidity, heat and corrosive substances.

In particular, the dedicated electronic components may be provided by electronic components comprising the computer readable memory described above. According to other embodiments, the application specific electronic components are provided by components having hardwired or having configurable circuitry, such as an Application Specific Integrated Circuit (ASIC) or a Field Programmable Gate Array (FPGA).

The learning process for learning the signal may be performed as follows. In this context, the signal generated by the learning process may be a digital filter or other discrete time series. The signal may be used for generating a sound signal or may be used for computational purposes for processing a digitized version of a received sound signal.

Learning is accomplished by capturing the step response of the channel under controlled conditions, such as at zero flow. This may be accomplished by using one or more pulses as the transmit signal transmitted through the channel. Here, "channel" means a communication channel including a fluid.

The step response is then processed as follows:

in the case of a PWM generated signal, the step response is time-reversed and digitized such that the PWM emitted signal corresponds to the time-reversed step response.

In the case of a burst input, the step response of the channel is stored in the digital receive filter after the analog-to-digital conversion.

The measurement phase can then be performed as follows:

in the case of using a PWM signal, the time-reversed channel impulse response obtained by the learning process is used as the transmission signal by supplying a time-reversed digital signal as an input to the PWM device and connecting the output of the PWM device to the ultrasonic transducer.

In case a pulse train is used, the same signal as used during the learning phase may be used as the transmission signal or the measurement signal in the measurement phase. For example, a step signal or a single pulse or a pulse train may be used as the transmission signal or the measurement signal. In one example, the pulse train has 10 to 30 pulses. The measurement signal may alternatively be longer or shorter than the learning phase.

In the case where a PWM signal is used as the transmit signal, the transmit signal is folded with the channel impulse response and a distinct peak is received at the input of the ADC. That is, in an approximation of the LTI system (linear time invariant system), the received response signal is a convolution of the transmitted signal with the previously determined channel impulse response.

The received signal is digitized and the peaks of the signal are used, for example, to derive time of flight information. This may be done by a high resolution sampling process or alternatively by interpolation of the received signal. The signal peak detection is PWM, and if the correlation process occurs in the channel, the peak only appears immediately after the ADC.

The peak value can also be calculated after the ADC if the stored inverse filter is associated with the receive filter.

In the case of using a burst as a transmission signal, we explain that signal processing can be performed by the DTRAF or DTRAC method according to the present invention. The DTRAC method may provide improved power efficiency compared to the DTRAF method. One reason for the improved efficiency is that the DTRAF method filters the entire received signal using an FIR filter. On the other hand, the DTRAC method does not perform such power consuming operations, but directly correlates the digitized measurement signals.

In the case of DTRAF, after the learning phase, the received signal is filtered with the inverse of the previously stored channel response, typically in a finite impulse response filter. As a result of this filtering operation, a distinct peak is obtained. This operation may be performed for upstream and downstream signals. The resulting signals are then cross-correlated and interpolated.

Second, in the case of DTRAC, the ADC signal of the measurement process is cross-correlated with the training signal and then interpolated. This can likewise be done for the upstream and downstream signals, so that the travel time difference Δ T between the upstream and downstream signals can be calculated.

Drawings

The subject matter of this specification is now explained in more detail with reference to several drawings, in which:

figure 1 shows a first flow meter arrangement with two piezoelectric elements,

fig. 2 shows the flow meter arrangement of fig. 1, one direct signal and two scattered signals,

figure 3 shows the flow meter arrangement of figure 1 when viewed in the flow direction,

figure 4 shows a second flow meter arrangement with four piezoelectric elements and four direct signals,

figure 5 shows the flow meter arrangement of figure 4 when viewed in the flow direction,

figure 6 shows a many-to-one sensor arrangement for flow measurement according to the invention,

figure 7 shows a one-to-many sensor arrangement for flow measurement according to the present invention,

figure 8 shows a one-to-one sensor arrangement for flow measurement in a layer according to the invention,

figure 9 shows a multisensor arrangement for flow measurement in multiple layers according to the present invention,

figure 10 shows a device for measuring a flow rate according to the invention,

figure 11 shows an iterative process for deriving waveforms for use in the flow meter of figure 10,

figure 12 shows another process for deriving waveforms for use in the flow meter of figure 10,

figure 13 shows another process for deriving waveforms for use in the flow meter of figure 10,

fig. 14 shows a comparison between the PWM signal of the pulse width modulator and the time-reversed signal, wherein the PWM signal approximates the time-reversed TRA signal,

figure 15 shows a comparison between the response signal to the PWM signal of figure 15 and the response signal to the time reversed signal of figure 15,

figure 16 shows simulation results for a PWM based correlation TRA,

figure 17 shows simulation results for signals generated by the digital TRA correlation method shown in figure 21,

figure 18 shows simulation results for signals generated using the digital TRA filtering method shown in figure 19,

figure 19 shows in block diagram form a visual display of the DTRAF method,

figure 20 shows a diagram of a general peak interpolation for determining the maximum of the correlation function,

figure 21 shows in block diagram form a visual display of the DTRAC method, an

Fig. 22 shows another embodiment of the proposed flow rate measuring device.

Detailed Description

In the following description, details are provided to describe embodiments of the present invention. It will be apparent, however, to one skilled in the art that the embodiments may be practiced without these specific details.

Some parts of the embodiments shown in the figures have similar parts. Like parts have the same name or like part number with an apostrophe or letter designation. Where appropriate, the description of such similar parts is also applicable by reference to other similar parts, thereby reducing repetition of the text without limiting the disclosure.

Fig. 1 shows a first flow meter arrangement 10. In this flow meter arrangement, the first piezoelectric element 11 is placed at the outer wall of a pipe 12, also referred to as tube 12. The second piezoelectric element 13 is placed on the opposite side of the pipe 12 so that the straight line between the piezoelectric element 11 and the downstream piezoelectric element 13 is oriented at an angle β to the mean flow direction 14, which is at the same time the direction of the symmetry axis of the pipe 12. In the example of fig. 1, the angle β is chosen to be about 45 degrees, but it may also be steeper, such as 60 degrees, or shallower, such as 30 degrees.

Piezoelectric elements, such as piezoelectric elements 11, 13 of fig. 1, may generally operate as acoustic emitters and acoustic sensors. The acoustic transmitter and acoustic sensor may be provided by the same piezoelectric element or by different regions of the same piezoelectric element. In this case, the piezoelectric element or transducer is also referred to as a piezoelectric transmitter when it operates as a transmitter or sound source, and as an acoustic sensor or receiver when it operates as an acoustic sensor.

When the flow direction is as shown in fig. 1, the first piezoelectric element 11 is also referred to as the "upstream" piezoelectric element, and the second piezoelectric element 13 is also referred to as the "downstream" piezoelectric element. The flow meter according to the present description functions in substantially the same way for both flow directions, and the flow direction of fig. 1 is provided as an example only.

Fig. 1 shows the flow of the electrical signal of fig. 1 in a configuration in which the upstream piezoelectric element 11 operates as a piezoelectric transducer and the downstream piezoelectric element 13 operates as an acoustic sensor. For the sake of clarity, the application works both upstream and downstream, i.e. the positions of the piezoelectric elements may be interchanged.

The first computing unit 15 is connected to the upstream piezoelectric element 11, and the second computing unit 16 is connected to the downstream piezoelectric element 13. The first calculation unit 15 includes a first digital signal processor, a first digital buffer, and a first analog-to-digital converter (ADC). Likewise, the second calculation unit 16 includes a second digital signal processor, a second digital buffer, and a second analog-to-digital converter (ADC). The first calculation unit 15 is connected to the second calculation unit 16.

The arrangement shown in fig. 1 with two calculation units 15, 16 is provided as an example only. Other embodiments may have different numbers and arrangements of computing units. For example, there may be only one central computing unit, or there may be two ADC or buffer units and one central computing unit, or there may be two small scale computing units and one larger central computing unit at the transducer.

For example, the one or more computational units may be provided by a microcontroller or an Application Specific Integrated Circuit (ASIC) or a Field Programmable Gate Array (FPGA).

A method for performing a measurement procedure according to the invention comprises the steps of:

a predetermined digital measuring signal is generated by synthesizing the electrical signals with the digital signal processor of the first calculation unit 15. The electrical signal is sent from the first calculation unit 15 to the piezoelectric transducer 11 along a signal path 17. The piezoelectric transducer 11 generates a corresponding ultrasonic test signal. The units 15 and 16 may also be arranged in one single unit.

The measurement signal is provided as an arbitrary waveform. For example, the arbitrary waveform may be provided by a pulse width modulated oscillation (such as a 1MHz oscillation) having a fundamental frequency in the MHz range. An arbitrary waveform can also be represented as a unit pulse.

The ultrasonic test signal propagates through the fluid (e.g., liquid) in the pipe 12 to the piezoelectric transducer 13. In fig. 1, the direct signal path of the ultrasound signal is indicated by arrow 18. Likewise, the direct signal path of the ultrasound signal in the reverse direction is indicated by arrow 19. The response signal is picked up by the piezoelectric sensor 13, sent along a signal path 20 to the second calculation unit 16, and digitized by the second calculation unit 16.

In a further step, a digital measurement signal is derived from the digitized response signal by signal processing. According to a further embodiment, the derivation of the measurement signal comprises further processing steps.

In a flow meter according to one embodiment of the present description, the same measurement signal is used for both directions 18, 19, i.e. the downstream and upstream directions, thereby providing a simple and efficient arrangement. According to other embodiments, different measurement signals are used for both directions. In particular, the measurement signal may be applied to the original receiver of the test signal. Such an arrangement may provide benefits for asymmetric conditions and pipe shapes.

The operation of the flow meter will now be explained in more detail with reference to fig. 1.

The ultrasonic measurement signal propagates through the liquid in the pipe 12 to the piezoelectric sensor 13. The response signal is picked up by the piezoelectric sensor 13, sent along a signal path 20 to the second calculation unit 16, and digitized by the second calculation unit 16.

A similar procedure is performed for signals propagating in the reverse direction 19, i.e. applying the above-mentioned measurement signal to the downstream piezoelectric element 13 and measuring the response signal by the upstream piezoelectric element 11 to obtain the upstream time of flight TOF in the reverse direction 19 _up . The first calculation unit 15 determines the velocity of the flow, for example according to the following formula

Where L is the length of the direct path between the piezoelectric elements 11, 13, β is the angle of inclination of the direct path between the piezoelectric elements 11, 13 to the mean flow direction, and c is the speed of sound in the liquid at a given pressure and temperature.

Upstream time of flight is given by

And the downstream time of flight is given by

Which generates the formula

By using this formula, there is no need to determine the temperature or pressure used to in turn determine the fluid density and sound velocity, or to directly measure the sound velocity or fluid density. Instead, first order errors do not cancel out for only one measurement direction.

Instead of using a factor of L times cos (β), the flow-related value can be derived from a calibration measurement with a known flow rate. These calibration values take into account further influences such as flow profiles and contributions from acoustic waves that are scattered without propagating along straight lines.

According to a further embodiment, the measurement signal to be supplied to the transmitting piezoelectric element is synthesized using an arbitrary signal form.

According to another embodiment of the present description, a cross-correlation technique is used to estimate the time of flight of a signal. In particular, the corresponding time shift may be evaluated by cross-correlating the received downstream or upstream signal with the received signal at zero flow rate according to the following formula:

where t and τ are time variables, sig _Flow Representing an upstream or downstream signal under measured conditions when fluid is flowing through the conduit, and wherein Sig _Flow Representing the signal at calibration conditions at zero flow.The infinite sum limit represents a sufficiently large time window [ -T1, + T2) from time T1 to time T2]. More generally, it is not necessary that-T1 and + T2 are the same, and for practical reasons this is advantageous for the flow meter.

The time-shifted TOF is then obtained by comparing the time of the maximum of the upstream correlation function with the time of the maximum of the downstream correlation function _up –TOF _down . The envelope of the correlation function may be used to more accurately determine the location of the maximum.

In another embodiment, a separate evaluation unit is provided between the first calculation unit 15 and the second calculation unit 16, which performs the calculation of the signal arrival time and the flow rate.

In general, the measurement signal of the acoustic sensor results from the superposition of a scattered signal and a direct signal. The scattered signal is reflected one or more times from the inner and outer walls of the pipe, including additional scattering processes within the pipe wall. This is shown by way of example in fig. 2.

The time of arrival may also be determined by using a matched filter technique. The received response signals can be modeled as a simple model based on assumptions of linear signal propagation and mirror-like reflections on the pipe wall

x(t)＝A*s(t-TOA)+n(t)

Where t is a time variable, a is an attenuation factor, s is the transmitted signal time-shifted by the unknown time of arrival TOA, and n (t) is a noise term. The time of arrival TOA is then derived by correlating the received signal x (t) with the time shift measurement signal s (t) according to,

y(t)＝Integral(-inf,inf,x(τ)s(t-τ))

where-inf, inf are negative infinity and positive infinity integral boundaries, and the convolution x (τ) s (t- τ) is a function of the integral with respect to the time variable τ.

The time of arrival being a time value or function argument at which the correlation becomes maximum

TOA＝argmax[y(t)]

The correlation can also be expressed by convolution with a "matched filter" h (t) which takes the form

h(t)＝a·s(t ₀ -t)，

Where a is the normalization factor and s (t) is the measurement signal. This procedure can be generalized to the case of multiple receivers as follows. A set of phase shifts is determined that maximizes the sum of the response signal amplitudes for each receiving transducer. The time of arrival is determined by applying a matched filter to the sum function of the individual received response signals shifted by the previously determined set of phase shifts.

The transducer configuration of fig. 1 is a straight line. Other arrangements are possible, such as "V" and "W" shaped configurations, which utilize reflections on opposite sides of the duct. The V and W shaped configurations act based on reflections on the pipe wall, which causes more scattering than the straight configuration. The subject matter of the present application will benefit from these configurations as long as the path is properly understood.

In the V-configuration, the two transducers are mounted on the same side of the pipe. To record the 45 degree reflections, they were placed about the diameter of the pipe apart in the flow direction. The W-shaped configuration utilizes triple reflection. Similar to the V-shaped configuration, the two transducers are mounted on the same side of the pipe. To record the signals after two 45 degree reflections, they were placed two pipe diameters apart in the flow direction.

Fig. 2 shows, by way of example, a first acoustic signal "1" propagating directly from the piezoelectric element 11 to the piezoelectric element 13, and a second acoustic signal "2" scattered twice at the periphery of the pipe 12.

For simplicity, the scattering events are shown as reflections in fig. 2 to 5, but the actual scattering process may be more complex. In particular, the most relevant scattering usually occurs in the pipe wall or at the material mounted in front of the piezoelectric transducer. Fig. 3 shows a view of the flow direction in fig. 2 in the viewing direction 3-3.

Figures 4 and 5 show a second sensor arrangement in which the further piezoelectric element 22 is at a 45 degree angle to the piezoelectric element 11 and the further piezoelectric element 23 is at a 45 degree angle to the piezoelectric element 13.

Furthermore, fig. 4 and 5 show direct or straight acoustic signal paths in the case of piezoelectric elements 11, 22 operating as piezoelectric transducers and piezoelectric elements 13, 23 operating as acoustic sensors. The piezoelectric element 23, which is located on the rear side of the pipe 12 in the view of fig. 4, is shown by a dashed line in fig. 4.

Fig. 6 to 9 show by way of example different arrangements of clamping-type piezoelectric transducers for which flow measurement according to the present description can be used. By providing a plurality of transmitting transducers with signals based on arbitrary waveforms, improved signals, e.g. signals with improved beamforming properties, may be obtained at the receiving transducers. By providing a plurality of receiving sensors, the received measuring signals can be evaluated more efficiently and/or a greater freedom of design for arbitrarily shaped measuring signals can be achieved.

Figures 6 to 9 are aligned so that the gravitational force on the liquid in the pipe 12 is directed downwards. However, arrangements that are rotated relative to the arrangements of fig. 6 to 9 may also be used. The viewing direction of fig. 6 to 9 is along the longitudinal axis of the pipe 12. The upstream or downstream position of the transducer is not shown in fig. 6 to 9.

In the arrangement of fig. 6, an array of five piezoelectric elements 31-35 is provided at a first location, and another piezoelectric element 36 is placed upstream or downstream of the first location. When the array of five elements 31-35 is used as a transmitter and the other element 36 is used as a receiver, the array of piezoelectric elements 31-35 can be used to obtain a predetermined wave front and achieve an improved focusing of the acoustic wave in a predetermined direction.

In the arrangement of fig. 7, a single piezoelectric element 37 is provided at a first location and an array of five piezoelectric elements 38-42 is placed upstream or downstream of the first location. An array of piezoelectric elements 38-42 may be used to obtain improved recording of the wavefront of the response signal. The improved recording may then be used to obtain an improved flow measurement signal, which is then applied to the single piezoelectric element 37.

Fig. 8 shows an arrangement of two piezoelectric elements 43, 44, one of which is placed downstream with respect to the other. The distance d of the connecting line between the piezoelectric elements 43, 44 to the axis of symmetry of the pipe 12 is approximately half the radius of the pipe 12, so that the flow layer at the distance d from the central axis of the pipe 12 can be measured.

Especially for a clamped transducer such as the piezoelectric elements 43, 44 shown in fig. 8, flow measurement according to the present description provides improved signals at the receiving piezoelectric elements 44, 43 by beamforming.

Figure 9 shows an arrangement of eight piezoelectric elements 45-52, which are spaced at 45 degrees. With respect to upstream-downstream placement, several arrangements are possible.

In one arrangement, the sensor locations alternate between upstream and downstream along the perimeter, such as upstream 45, 47, 49, 51 and downstream 46, 48, 50, 52.

In another arrangement, the first four consecutive elements, e.g., 45-48, are placed circumferentially upstream or downstream relative to the other four elements, e.g., 49-52. In another arrangement with 16 piezoelectric elements, all piezoelectric elements 45-52 of fig. 9 are placed in one plane and the arrangement of fig. 9 is repeated in the upstream or downstream direction.

In the arrangements of fig. 6 to 9, the receiving transducer is offset relative to the transmitting transducer with respect to the longitudinal direction or flow direction of the conduit. In particular, transducer 36 of fig. 6 is offset with respect to transducers 31-35, transducer 37 of fig. 7 is offset with respect to transducers 38-42, transducer 44 of fig. 8 is offset with respect to transducer 43, and in fig. 9, the opposing transducers are offset with respect to each other in the longitudinal direction of the catheter. For example, transducer 51 is offset relative to transducer 47, while transducer 52 is offset relative to transducer 46.

Fig. 10 shows by way of example a flow measuring device 60 for measuring the flow in the arrangement in fig. 1 or in other arrangements according to the invention. In the arrangement of fig. 1, the flow measuring device 60 is provided by a first computing unit 15 and a second computing unit 16 (not shown in fig. 10).

The first connector 61 of the flow measuring device 60 is connected to the first piezoelectric transducer 11 at the fluid conduit 12 and the second connector 62 of the flow measuring device is connected to the second piezoelectric transducer 13 at the fluid conduit 12.

Inside the flow measuring device 60, the first connector 61 is connected to the analog-to-digital converter 64 through the multiplexer 63 and the first amplifier 74. The second connector 62 is connected to a digital buffer 67 via a second amplifier 75 and a demultiplexer 66.

The buffer 67 is connected to a waveform generator 69, the waveform generator 69 being connected to a waveform database 70. The waveform database 70 is connected to the ADC64 through a matching module 68, wherein the matching module 68 is foreseen to match parameters to the specific environment of the catheter under test 12. The ADC64 is also connected to a speed calculation unit 71, the speed calculation unit 71 being connected to a result memory 72.

During the signal generation phase, the waveform generator 69 extracts waveform parameters from the waveform database 70, derives electrical signals from the extracted waveform parameters, and sends the signals to the digital buffer 67.

In particular, the flow measuring device 60 may be provided with an electronic processing unit 53 in the form of a microcontroller or FPGA, which comprises several ADCs 64 with high resolution, a digital correlator (not shown in fig. 10) and modules supporting signal envelope detection, etc. In the transmission direction, the flow measuring device 60 generates a digital signal with a variable frequency. In the case of a square wave signal, the duty cycle of the signal can be varied by modulating the pulse width of the square wave signal, which is also referred to as PWM.

The ADC64, the matching module 68, the waveform database 70, the waveform generator 69, the buffer 67 are for example provided on an electronic processing unit 53, such as a microcontroller or a microprocessor. The electronic processing unit 53 has a connector pin 54 for connecting the first amplifier 74, a connector pin 55 for connecting the second amplifier 75, a connector pin (not shown) for connecting a power supply battery (not shown), a connector pin (not shown) for connecting a ground potential (not shown), and the like.

Other components not shown in fig. 10, such as a power converter, may be provided between the power supply battery and the connector pins.

In an alternative embodiment, the "current fingerprint" of the electronic processing unit 53 may be measured when the proposed method is performed. It has been found that the calculated part of the method is not in the peak (which would be the apparent current fingerprint), but in a flat area after the peak.

Flow measuring device 60 may transmit a signal at a predetermined frequency and amplitude. These signals are output by a digital buffer 67. Alternatively, the output may be provided by a component that converts a predetermined output signal into a PWM signal.

The device of fig. 10 is shown by way of illustration. An apparatus for performing flow measurements in accordance with the present description may include more or fewer components than shown in FIG. 10. In particular, digital signals according to the DTRAF and DTRAC methods mentioned below may be generated by the adjustable pulse generator unit 76 and do not require the matching module 68 and/or the waveform database 70 and/or the waveform generator 69.

In the case of a PWM-generated time-reversed signal, a function of time-reversing the received response signal or a part of the received response signal and transmitting the reversed signal as an input signal to the pulse width modulator is provided.

The processes of fig. 11 to 13 below can be used to generate a particularly suitable digital output signal for the measurement according to the invention. Wherein the phase and amplitude of the digital signal may vary.

Fig. 11 shows an iterative process for generating ultrasound output signals that meet predefined criteria.

In a first step 80, a measurement signal is generated with a pulse generator. In a second step 81, the measurement signal is applied to the first transducer 11. In step 82, the response signal is measured at the second transducer 13.

In a further step 83, the response signal is evaluated according to predetermined criteria. For example, the response signal may be matched to a predetermined waveform. If it is determined in step 84 that the response signal meets the predetermined criteria, then in step 86 the parameters of the measurement signal are determined and stored, preferably in the waveform database 70, for later use.

Otherwise, the frequency, amplitude and/or phase of the measurement signal or also other signal parameters are adjusted in step 85 and the process loops back to the first step 80 of generating the predetermined measurement signal.

This iterative approach may also be applied to multiple transducer arrangements. In the case of a plurality of transmitting transducers, the respective frequency, amplitude or phase of the respective measuring signal is adjusted. In the case of multiple receiving transducers, the criteria is applied to the response signals received at the receiving transducers.

Fig. 12 shows another method of deriving an arbitrary waveform signal, comprising the steps of:

in step 90, the transducer bandwidth is measured. In one example, the transducer bandwidth is about 300kHz near the transducer center frequency of about 1 MHz. One example of a signal that takes full advantage of this transducer bandwidth is a signal having a rectangular bandwidth of 300kHz in the frequency domain.

A bandwidth limited function such as a rectangular signal is generated in the frequency domain. In step 92, the corresponding signal or function in the time domain is obtained by applying an inverse fourier transform, which results in a sinc-like function.

Then in step 93 the sinc-like function is truncated to an appropriate signal length that does not have too much signal power but sufficient information. This signal is then used as an input signal for a pulse generator or pulse width modulation to generate a pulse train in step 94.

In steps 95 and 96, the resulting signal is used as a measurement signal in the upstream and downstream directions.

Fig. 13 shows another method of deriving a measurement signal. The method of fig. 13 is a variation of the method of fig. 12. For the sake of brevity, similar steps are not explained again. According to the method of fig. 13, the function or waveform is adjusted so that there is no offset at zero flow of the fluid.

In step 105, it is tested whether the signal time offset is below a predetermined threshold. If the offset is below a predetermined threshold, the signal parameter is stored in step 107. Otherwise, the process loops back to step 101.

Another method (not shown in the figure) is explained below:

1) The time-of-flight difference Δ T is obtained from the difference between the upstream and downstream measurements.

2) The signal frequency is adjusted. Thereafter, the process loops back to 1) and the amplitude and phase of the time domain signal are changed until there is a zero offset, which means that a zero time difference is measured for a zero fluid flow.

3) Alternatively, the signal according to 1) is generated with a correction step 2). Predistortion is applied to the signal according to a predetermined criterion. The predistortion may be selected such that the receiver may be designed in a manner suitable for the particular received response signal. For example, the signals according to 1) and 2) may be pre-distorted such that the zero crossings at the receiver occur at equidistant time intervals. Thus, a narrow bandwidth receiver may be used. This applies to both time of flight and TRA measurements.

In steps 1-3, an arbitrary digital waveform is used, rather than just oscillations with a rectangular envelope. The above process and the processes of fig. 11 to 13 can be used for all digital time-of-flight and time-reversed acoustic systems.

Hereinafter, flow measurement using the DTRAF method, the DTRAC method, and the time-reversal signal applied to pulse width modulation is explained in more detail.

Digital TRA filtering (DTRAF) is a digital version of the known TRA method. The basic concept of DTRAF is not to transmit a time-reversed signal but to replace it with a digital filter operation, but the time-reversed signal can also be transmitted in combination with DTRAF (or DTRAC) without changing the basic principle. Thus, the DTRAF training process produces a digital FIR filter in which the TRA method is performed digitally. This simplifies the electronics in terms of removing lower specifications on TDC and DAC, for example in terms of accuracy and electrical power consumption. Rather, the ADC is an important component. Finally, interpolation of the processed signal increases the accuracy in the range from the sampling period to picoseconds.

Fig. 19 shows the proposed DTRAF method in block diagram form.

In the arrangement of fig. 19, the digital buffer 67 of the arrangement of fig. 10 is replaced by an Adjustable Pulse Generator Unit (APGU) 76. The method is now explained in more detail:

first, the TRA training process for the flow measurement device 60 is completed with high accuracy or with an on-board ADC. The output of the training is a digital FIR filter. The system is excited in a training mode by an input signal x [ n ]:

h _MF，12 (t)＝h ₁₂ (t)*x(t)→h _MF，12 [n]＝Q(h _MF，12 (n·T _s ))， (1)

h _MF，21 (t)＝h ₂₁ (t)*x(t)→h _MF，21 [n]＝Q(h _MF，21 (n·T _s ))， (2)

wherein, T _s Is the sampling period. This sampling period must match the sampling period measured later. With a faster ADC, the digital filter needs to be resampled to the sampling period of the measurement mode.

In the measurement mode, the system is excited again with the input signal x [ n ]:

y′ ₁₂ (t)＝h ₁₂ (t)*x(t)→y′ ₁₂ [n]＝Q(y′ ₁₂ (n·T _s ))， (3)

y′ ₂₁ (t)＝h ₂₁ (t)*x(t)→y′ ₂₁ [n]＝Q(y′ ₂₁ (n·T _s )). (4)

y′ ₁₂ [n]and y' ₂₁ [n]Is a digitized measurement signal. Here and in the following, the variable in square brackets represents a discrete time index, which in the case of uniform sampling starting from time t =0 corresponds to n times the sampling time Ts.

In a next step, these digitized measurement signals are convolved with a time-reversed training digital filter:

y ₁₂ [n]＝h _MF，12 [-n]*y′ ₁₂ [n]， (5)

y ₂₁ [n]＝h _MF，21 [n]*y′ ₂₁ [n]. (6)

this additional filtering operation increases the computation time due to the many multiplication and addition operations. It is noted here that it may be beneficial when using the same digital filter unit in both convolutions instead of two different trained digital filter units 120, 121. The time reversal can be performed in the memory by storing in the reverse order or also by reading out in the reverse order.

The two filtered measurement signals y are then ₁₂ [n]And y ₂₁ [n]In a cross-correlation unit 122Correlated with each other (or matched by any pattern matcher):

r[n]＝y ₁₂ [n]★y ₂₁ [n]. (7)

it is important to note here that the cross-correlation with the cross-correlation unit 122 need not be fully calculated. In particular, the correlation function need not be sampled at a high frequency where the maximum value is always close to the sampling time. The cross-correlation is an estimator of the time delay and its argument maximum corresponds to the time delay between two measured signals:

n ^* ＝arg max(r[n])， (8)

where n is the time index and r is the two measurement signals y ₁₂ [n]、y ₂₁ [n]A correlation function between.

The time delay between the upstream and downstream signals is twice the time delay between the zero flow signal and the current fluid flow signal.

However, in order not to calculate the entire cross-correlation, it is sufficient to have an estimate of where the argument maximum is located. This can be done by estimating the absolute transit time T _abs，12 And T _abs，21 To complete. Subtracting these transit times, we obtain by the following definition:

2·ΔT＝T _abs，12 -T _abs，21 . (9)

at low sampling rates, its accuracy is sufficient to estimate n ^* ：

Wherein the content of the first and second substances,

represents n ^* Is estimated. In terms of the geometry, it is preferred that,

is two filtered measurement signals y ₁₂ [n]And y ₂₁ [n]Integer number of samples in between.

Finally, only after hysteresis

Several points around the cross-correlation (or any pattern matcher) are computed. The number of points depends on the interpolation method. The interpolation method increases the accuracy in the range from the sampling period to picoseconds. As an example, cosine interpolation (requiring three points) works well for such a small band ultrasound transducer signal with cross-correlation method. Cosine peak interpolation yields slave lag

Distance d to the hidden maximum of the continuous cosine function. This interpolation is shown in fig. 20.

From "A line-Based Algorithm for Continuous Time-Delay Estimation Using Sampled Data" (IEEE transactions on ultrasound, ferroelectronics and frequency control, vol.52, no.1, 1 month 2005) of F.Viola, W.Walker, it is known that the cosine peak interpolation of the function y [ k ] can be calculated in the following way:

this d-corrected lag estimate is then

To obtain a highly accurate Δ T:

the multiplication by 1/2 can be omitted when considered in the final calculation of the flow rate.

Compared to DTRAF (digital TRA-filtering), there is a slightly different approach, called DTRAC (digital TRA-correlation). The training and measurement procedure to get the trained FIR filter and measurement signal is the same as DTRAF. However, the signal processing chain changes. Instead of filtering the entire signal with a (time-reversed) trained FIR filter, only a few points are calculated by cross-correlation with a non-time-reversed FIR filter. This is done for the upstream and downstream measurement signals.

By interpolation of these correlation points, the exact absolute transit time is obtained, which can then be used to calculate Δ T by subtraction. The DTRAC method is exemplarily shown as a block diagram in fig. 21.

Similar to the DTRAF method, DTRAC may be performed using an adjustable pulse generator unit 76. In the following, the proposed DTRAC method is explained in more detail:

first, TRA training is completed using an input signal x [ n ] in the same manner as DTRAF, and x (t) is x [ n ] in a time-continuous form:

h _MF，12 (t)＝h ₁₂ (t)*x(t)→h _MF，12 [n]＝Q(h _MF，12 (n·T _s ))， (15)

h _MF，21 (t)＝h ₂₁ (t)*x(t)→h _MF，21 [n]＝Q(h _MF，21 (n·T _s )). (16)

this yields the FIR filter coefficients h _MF，12 [n]And h _MF，21 [n]Wherein T is _s Is the sampling period.

Measurements are made in the same manner as TRA training. Using the same input signal x [ n ]:

y′ ₁₂ (t)＝h ₁₂ (t)*x(t)→y′ ₁₂ [n]＝Q(y′ ₁₂ (n·T _s ))， (17)

y′ ₂₁ (t)＝h ₂₁ (t)*x(t)→y′ ₂₁ [n]＝Q(y′ ₂₁ (n·T _s )). (18)

quantizing the measurement signal and using T _s And (6) sampling. The measurement signal is then cross-correlated with a digital filter:

r ₁₂ [n]＝y′ ₁₂ [n]★h _MF，12 [n]， (19)

r ₂₁ [n]＝y′ ₂₁ [n]★h _MF，21 [n]. (20)

the cross-correlation is an estimator of the time delay. However, other time delay estimators (or pattern matchers) may be used, for example, as a Sum of Squared Differences (SSD). It is not necessary to compute the two cross-correlations completely, but only a few points around the actual maximum peak. This is sufficient for the calculation of the interpolation method:

or, in a more general case, n at a low sampling rate ^* Estimation of (2):

corresponds to n ^* Is estimated. Large sampling period reduces errors

Or

The estimated risk. The interpolation method uses time discrete interactionKnowledge of the time-continuous function after correlation (or any pattern matcher). In the case of a narrowband ultrasound transducer, the cross-correlation output tends to be very similar to a cosine oscillation around the actual maximum peak. This effect is even greater in the case of an input signal that only excites the resonant frequency of the ultrasound transducer pair, such as a long sine, cosine, triangle or rectangular oscillation. The interpolation method estimates the index of the actual maximum of the time-discrete cross-correlation in the subsamples (and the number of times multiplied by the sampling period):

d ₁₂ and d ₂₁ Sub-sample correction corresponding to the interpolation method. Due to absolute transit time T _abs,12 And T _abs,21 Very precisely, Δ T can be obtained by subtracting these:

when considered in the calculation of the flow rate, the multiplication by 1/2 may be omitted.

The proposed DTRAF and DTRAC methods can simulate in a computer the channel 77 with the catheter 12 and the fluid. This can be explained as follows: if the LTI system has blocks or transfer functions a, B, C, D, the resulting transfer function of the frequency domain is the product a x B C D of the respective transfer functions a, B, C, D. The product does not depend on the order of the transfer functions a, B, C, D. Thus, the computation of the channel response may be "shifted" into the digital domain.

Fig. 14 shows a PWM signal 129 of the pulse width modulator with a suitable modulation on/off period such that the PWM signal 129 approximates a time-reversed TRA signal 130.

Fig. 15 shows a comparison of the response signal 131 to the PWM signal 129 and the response signal 132 to the time-inverted TRA signal 130.

According to the PWM-based TRA method, according to U _A ＝A·sign(U _E -U _D ) To approximate a time-reversed signal, wherein for x>=0,sign (x) =1, otherwise 0. As an example, pulse width modulation may be based on an example with a transducer, a triangular wave signal frequency of 2MHz and a PWM time resolution of about 83 ns.

The approximation of the time reversed signal shown in fig. 14 is quite good, even though the PWM generation remains simple and no additional filter is used.

The simulation shown in fig. 16 with the associated TRA based on PWM illustrates how the performance of the delta T varies. For simulations in which the SNR of the PWM based correlated TRA must be increased, the signal power is calculated and the noise is scaled.

The results of fig. 16 and 17 were obtained by simulations in which the system including signal generation, transducers, channels and signal reception was modeled as a linear time-invariant system. The simulation simulates a measurement with zero flow.

Fig. 16 shows the measured time delay of the relevant TRA based on PWM. The measured time delays are indicated by a histogram 133 and the resulting probability densities are shown by gaussian curves 134.

Fig. 17 shows a measurement of a signal generated using the digital TRA correlation method described above. The measured time delays are indicated by a histogram 135 and the resulting probability densities are shown by gaussian curves 136.

Fig. 18 shows a measurement of a signal generated using the digital TRA filtering method described above. The measured time delays are indicated by a histogram 135 and the resulting probability densities are shown by gaussian curves 136.

Comparisons between methods yield relatively similar results (e.g., with respect to mean, standard deviation). Note that the simulation parameters used to generate the measurements shown in fig. 16, 17 and 18 are differential.

For the measurement of fig. 18, a pulse train with 25 pulses was used as input signal. The first 50 measurements of TRA training are averaged and stored in two FIR filters. Δ T _ DTRAF is the result of the digital TRA filtering method (DTRAF). The test tube has a closed end for making a zero flow measurement.

Fig. 19 shows in block diagram form a visual display of the DTRAF method. Here, the adjustable pulse generator unit 76 is connected on the input side to the input signal source and on the output side to the first ultrasonic transducer 11. A first ultrasonic transducer 11 is coupled to a channel 77, such as a conduit 12 with a fluid. The second ultrasonic transducer 13 is coupled to channel 77 and the output of the second ultrasonic transducer 13 is coupled to the ADC 64. The same arrangement is also used during the measurement phase, which is shown in fig. 19 below the training phase.

The bottom row shows the signal evaluation of the digitized received response signal during the measurement phase. The first digital filter unit 120 and the second digital filter unit 121 are respectively connected to the output terminal of the ADC 64. Respective outputs of the digital filter units 120, 121 are connected to an input of a cross-correlation unit 122, an output of the cross-correlation unit 122 being connected to an input of an interpolation unit 123.

Fig. 20 shows a diagram of a general peak interpolation for determining the maximum of the correlation function. The general peak interpolation according to fig. 20 has been explained further above and is not repeated here in detail. In short, the correlation function is locally approximated by a cosine function or a higher order polynomial. The cosine is obtained by interpolation using three interpolation points y [ k-1], y [ k ], and y [ k +1 ]. Then, the maximum value of the cosine function is determined. If more interpolation points are available, the cosine can be found using least squares. The distance of the interpolation point to the cosine can also be minimized according to different kinds of norms.

Figure 21 shows in block diagram form a visual display of the proposed DTRAC method. The arrangement of fig. 21 is similar to that of fig. 19. For the sake of brevity, only the components in the bottom row that differ from those in fig. 19 are described below.

An input of the cross-correlation unit 122 is connected to an output of the ADC64 and an output of the cross-correlation unit 122 is connected to an input of the interpolation unit 123.

Fig. 19 and 21 can also be seen as possible implementations of the DTRAC and DTRAF methods by hardware and/or software components. The various units shown in fig. 19 and 21 may be implemented by separate components or on one component, and they may be implemented in hardware and/or software. For example, the cross-correlation unit 122 and the interpolation unit 123 may be implemented on the same integrated circuit. The training and measuring process according to fig. 19 to 21 has been explained further above and is not repeated here.

Fig. 22 shows by way of example a further embodiment of the proposed flow measurement device 60 'for measuring a flow velocity of a fluid in a fluid conduit 12 using an ultrasonic time-of-flight flow meter 60' using a digital time-reversed acoustic filtering method.

The buffer 67 is connected to an adjustable pulse generator unit 76. The ADC64 is also connected to digital filter units 120, 121. The digital filter units 120, 121 are connected to a correlation unit 122, which is connected to an interpolation unit 123. The interpolation unit 123 is connected to a velocity calculation unit 71 which is connected to a result memory 72 and a buffer 67.

Reference numerals are as follows:

10. flow meter device

11. Upstream piezoelectric element

12. Catheter tube

13. Downstream piezoelectric element

14. Mean direction of flow

15. First computing unit

16. Second computing unit

17. Signal path

20. Signal path

22. Piezoelectric element

23. Piezoelectric element

31-52 piezoelectric element

53. Electronic processing unit

54. Connector pin

55. Connector pin

56. Connector pin

60. 60' flow measuring device

61. First connector

62. Second connector

63. Multiplexer

64 ADC

66. Demultiplexer and method of demultiplexing data

67. Digital buffer

68. Matching module

69. Waveform generator

70. Waveform database

71. Speed calculation unit

72. Result memory

73. Measuring signal generator

74. First amplifier

75. A second amplifier

76. Adjustable pulse generator unit

77. Channel

90-96 method steps

100-106 method steps

110-117 method steps

120. First digital filter

121. Second digital filter

122. Correlation unit

123. Interpolation unit

129 PWM signal

130 TRA signal

131. Response signal to PWM signal

132. Response signal to TRA signal

133. Measured time delay

134. Density of probability

135. Measured time delay

136. Density of probability

137. Measured time delay

138. Density of probability

Claims (14)

1. A method for determining a flow rate of a fluid in a channel (77) having a fluid conduit (12) with an ultrasonic time of flight flow meter (60) using a digital time-reversal acoustic filtering method, the method comprising the steps of:

-performing a training process with an adjustable pulse generator unit (76), the training process comprising the steps of:

-applying an input signal (x [ n ]) to a first ultrasound transducer (11), the first ultrasound transducer (11) being mounted to the fluid conduit (12) at a first location;

-receiving a first response signal to the input signal (x [ n ]) at a second ultrasonic transducer (13), the second ultrasonic transducer (13) being mounted to the fluid conduit (12) at a second location, the second location being upstream or downstream of the first location with respect to a flow direction (14) of the fluid;

-converting the first response signal into a first digitized response signal using an analog-to-digital converter (64); and

-deriving a first digital response filter (120) from the first digitized response signal;

-applying the input signal (x [ n ]) to the second ultrasonic transducer (13);

-receiving a second response signal to the input signal (x [ n ]) at the first ultrasonic transducer (11);

-converting the second response signal into a second digitized response signal using the analog-to-digital converter (64); and

-deriving a second digital response filter (121) from the second digitized response signal;

-performing a measurement procedure with the adjustable pulse generator unit (76), the measurement procedure comprising the steps of:

-applying the input signal (x [ n ]) to the first ultrasonic transducer (11);

-receiving a third response signal of the input signal (x [ n ]) at the second ultrasound transducer (13);

-converting the response signal into a third digitized response signal (y) using the analog-to-digital converter (64) ₁₂ '[n])；

-performing a reverse measurement procedure with the adjustable pulse generator unit (76), the reverse measurement procedure comprising the steps of:

-applying the input signal (x [ n ]) to the second ultrasonic transducer (13);

-receiving a fourth response signal to the input signal (x [ n ]) at the first ultrasonic transducer (11);

-converting the fourth response signal into a fourth digitized inverted response signal (y) using the analog-to-digital converter (64) ₂₁ '[n])；

-from said third digitized response signal (y) ₁₂ '[n]) And said first digital response filter (120) deriving a first correlation input signal (y) ₁₂ [n]) Said deriving comprising inverting said third digitized response signal (y) with respect to time ₁₂ '[n]) Or the first digital response filter (120);

-from said fourth digitized response signal (y) ₂₁ '[n]) And said second digital response filter (121) to derive a second correlation input signal (y) ₂₁ [n]) Said deriving comprising inverting said fourth digitized response signal (y) with respect to time ₂₁ '[n]) Or said second digital response filter (121); and

-by calculating said first correlation input signal (y) ₁₂ [n]) Input signal (y) with said second correlation ₂₁ [n]) According to said first correlation, inputting a signal (y) ₁₂ [n]) And said second correlation input signal (y) ₂₁ [n]) Deriving a time-of-flight difference (Δ T), wherein a time index of a maximum of the discrete correlations is determined using an interpolation method.

2. The method of claim 1, wherein the second digital response filter (121) is equal to the first digital response filter (120).

3. The method of claim 1, wherein the second digital response filter (121) is a digital inverse response filter, wherein the determining of the inverse response filter comprises the steps of:

-performing a reverse training procedure with the adjustable pulse generator unit (76);

-applying the inverted input signal (x [ n ]) to the second ultrasonic transducer (13);

-receiving an inverse response signal to the inverse training signal at the first ultrasound transducer (11);

-converting the inverse response signal into a digitized inverse response signal using the analog-to-digital converter (64);

-deriving digital inverse response filter taps (h) from said digitized inverse response signals _MF,21 [n]) (ii) a And

-filter taps (h) according to the digital inverse response _MF,21 [n]) -deriving the second digital response filter (121).

4. A method for determining a flow rate of a fluid in a channel (77) having a fluid conduit (12) with an ultrasonic time of flight flow meter (60) using a digital time-reversal acoustic filtering method, the method comprising the steps of:

-performing a training procedure with an adjustable pulse generator unit (76), the training procedure comprising the steps of:

-applying an input signal (x [ n ]) to a first ultrasound transducer (11), the first ultrasound transducer (11) being mounted to the fluid conduit (12) at a first location;

-receiving a first response signal to the input signal (x [ n ]) at a second ultrasonic transducer (13), the second ultrasonic transducer (13) being mounted to the fluid conduit (12) at a second location, the second location being upstream or downstream of the first location with respect to a flow direction of the fluid;

-converting the first response signal into a first digitized response signal using an analog-to-digital converter (64);

-determining from the first digitized response signal to have filter taps (h) _MF,21 [n]) The first digital response filter of (a);

-applying the input signal (x [ n ]) to the second ultrasonic transducer (13);

-receiving a second response signal at the first ultrasound transducer (11);

-converting the second response signal into a second digitized response signal using the analog-to-digital converter (64); and

-determining from the second digitized response signal to have filter taps (h) _MF,21 [n]) The second digital response filter of (2) is,

-performing a measurement procedure with the adjustable pulse generator unit (76), the measurement procedure comprising the steps of:

-applying the input signal (x [ n ]) to the first ultrasonic transducer (11);

-receiving a third response signal of the input signal (x [ n ]) at the second ultrasound transducer (13);

-converting the third response signal into a third digitized response signal (y) using the analog-to-digital converter (64) ₁₂ '[n])；

-performing a reverse measurement procedure with the adjustable pulse generator unit (76), the reverse measurement procedure comprising the steps of:

-applying the input signal (x [ n ]) to the second ultrasonic transducer (13);

-receiving a fourth response signal to the input signal (x [ n ]) at the first ultrasonic transducer (11);

-converting the fourth response signal into a fourth digitized response signal (y) using the analog-to-digital converter (64) ₂₁ '[n])；

-by calculating said third digitized response signal (y) ₁₂ '[n]) Discrete correlation with said first digitized response signal, in dependence on said third digitized response signal (y) ₁₂ '[n]) And said first digitized response signal to derive a first time of flight (T) _abs,12 ) Wherein a time index of a maximum value of the discrete correlation is determined using an interpolation method;

-by calculating said fourth digitized response signal (y) ₂₁ '[n]) Discrete correlation with said second digitized response signal, in dependence on said fourth digitized response signal (y) ₂₁ '[n]) And said second digitized response signal to derive a second time of flight (T) _abs,21 ) Wherein the determination is made using an interpolation methodA time index of a maximum value of the discrete correlations; and

-by means of a previously derived time of flight (T) _abs,12 、T _abs,21 ) The subtraction of (d) yields a time-of-flight difference (Δ T).

5. The method of claim 4, wherein the second digital response filter is equal to the first digital response filter.

6. The method of claim 4, wherein the first digital response filter is a forward response filter;

wherein the second digital response filter is an inverse response filter;

wherein a filter tap (h) of the forward response filter is derived from the first digitized response signal _MF,12 ) (ii) a And

wherein a filter tap (h) of the inverse response filter is derived from the second digitized response signal _MF,21 )。

7. A method for determining a flow rate of a fluid in a fluid conduit (12) using an ultrasonic time-of-flight flow meter, the method comprising the steps of:

-generating a training signal with an adjustable pulse generator unit (76);

-applying the training signal to a first ultrasound transducer (11), the first ultrasound transducer (11) being mounted to the fluid conduit (12) at a first location;

-receiving a response signal to the training signal at a second ultrasound transducer (13), the second ultrasound transducer (13) being mounted to the fluid conduit (12) at a second location, the second location being upstream or downstream of the first location with respect to the flow direction of the fluid;

-converting the response signal into a digitized response signal using an analog-to-digital converter (64);

-inverting the digitized response signal with respect to time so as to obtain an inverted digitized response signal;

-generating a pulse width modulated measurement signal from the inverted digitized response signal;

-applying the pulse width modulated measurement signal to the first ultrasonic transducer (11);

-receiving a response signal to the pulse width modulated measurement signal at the second ultrasound transducer (13); and

-deriving a time of flight (Δ T) from the response signal.

8. An electronic processing unit (53) for determining a flow rate of a fluid in a channel (77) having a fluid conduit (12) with an ultrasonic time-of-flight flow meter (60) using a digital time-reversal acoustic filtering method, comprising:

-an adjustable pulse generator unit (76), the pulse generator unit (76) being operable to perform a training process with an input signal (x [ n ]);

-a transmission module operable to apply the input signal (x [ n ]) to a first ultrasound transducer (11), the first ultrasound transducer (11) being mounted to the fluid conduit (12) at a first location;

-a receiving module operable to receive a first response signal to the input signal (x [ n ]) at a second ultrasonic transducer (13), the second ultrasonic transducer (13) being mounted to the fluid conduit (12) at a second location, the second location being upstream or downstream of the first location with respect to a flow direction of the fluid;

-an analog-to-digital converter (64), the analog-to-digital converter (64) being operable to convert the first response signal into a first digitized response signal;

-a processing module operable to determine from the digitized response signal that there are FIR filter taps (h _ \ } _MF12 [n]) The first digital response filter (120);

-the transmission module is further operable to apply the input signal (x [ n ]) to a second ultrasound transducer (13), the second ultrasound transducer (13) being mounted to the fluid conduit (12) at the second location;

-the receiving module is further operable to receive a second response signal to the input signal (x [ n ]) at the first ultrasonic transducer (11);

-the analog-to-digital converter (64) is further operable to convert the second response signal into a second digitized response signal;

-said processing module is further operable to determine from said second digitized response signal that there are FIR filter taps (h) _MF,21 [n]) A second digital response filter (121);

-the adjustable pulse generator unit (76) is further operable to perform a measurement procedure using the input signal (x [ n ]);

-the transmission module is further operable to apply the input signal (x [ n ]) of the measurement process to the first ultrasonic transducer (11);

-the receiving module is further operable to receive a third response signal to the input signal (x [ n ]) of the measurement process at the second ultrasound transducer (13);

-the processing module is further operable to convert the third response signal into a third digitized response signal (y) using the analog-to-digital converter (64) ₁₂ ')；

-the adjustable pulse generator unit (76) is further operable to perform an inverse measurement procedure using the input signal (x [ n ]);

-the transmission module is further operable to apply the input signal (x [ n ]) of the inverse measurement process to the second ultrasonic transducer (13);

-the receiving means is further operable to receive a fourth response signal of the inverse measurement process at the first ultrasound transducer (11);

-the analog-to-digital converter (64) is further operable to convert the fourth response signal of the inverse measurement process into a fourth digitized response signal (y) using the analog-to-digital converter (64) ₂₁ ')；

-the processing module is further operable to:

-from said third digitized response signal (y) ₁₂ ') and said first digital response filter (120) to derive a first correlationInput signal (y) ₁₂ ) Said deriving comprising inverting said third digitized response signal (y) with respect to time ₁₂ ') or the first digital response filter (120);

-from said fourth digitized response signal (y) ₂₁ ') and said second digital response filter (121) to derive a second correlation input signal (y) ₂₁ ) -said deriving comprises inverting said fourth digitized response signal or said second digital response filter (121) with respect to time;

-by calculating said first correlation input signal (y) ₁₂ [n]) And said second correlation input signal (y) ₂₁ [n]) And determining a time index of a maximum value of said discrete correlation using interpolation to input a signal (y) from said first correlation ₁₂ [n]) And said second correlation input signal (y) ₂₁ [n]) A time-of-flight difference (Δ T) is derived.

9. The electronic processing unit (53) of claim 8, wherein the second digital response filter (121) is equal to the first digital response filter (120).

10. The electronic processing unit (53) of claim 8, wherein the second digital response filter (121) is a digital inverse response filter, and wherein the electronic processing unit (53) is further operable to

-generating an inverse training signal with an adjustable pulse generator unit (76);

-applying the inverse training signal to the second ultrasound transducer (13);

-receiving an inverse response signal to the inverse training signal at the first ultrasound transducer (11);

-converting the inverse response signal into a digitized inverse response signal using an analog-to-digital converter (64); and

-deriving a signal having FIR filter taps (h) from the digitized inverse response signal _MF,21 [n]) The second digital response filter (121).

11. An electronic processing unit (53) for determining a flow rate of a fluid in a channel (77) having a fluid conduit (12) with an ultrasonic time of flight flow meter (60) using a digital time-reversal acoustic filtering method, wherein the electronic processing unit (53) is operable to:

-performing a training procedure with an adjustable pulse generator unit (76);

-applying an input signal (x [ n ]) to a first ultrasound transducer (11), the first ultrasound transducer (11) being mounted to the fluid conduit (12) at a first location;

-receiving a first response signal to the input signal (x [ n ]) at a second ultrasonic transducer (13), the second ultrasonic transducer (13) being mounted to the fluid conduit (12) at a second location, the second location being upstream or downstream of the first location with respect to a flow direction of the fluid;

-converting the first response signal into a first digitized response signal using an analog-to-digital converter (64);

-deriving a signal having FIR filter taps (h) from the first digitized response signal _MF,12 [n]) The digital response filter of (1);

-performing a reverse training procedure with the adjustable pulse generator unit (76);

-applying the input signal (x [ n ]) to the second ultrasonic transducer (13);

-receiving a second response signal at the first ultrasound transducer (11);

-converting the second response signal into a second digitized response signal using the analog-to-digital converter (64);

-determining from the second digitized inverse response signal having filter taps (h) _MF,21 [n]) The digital response filter of (2);

-performing a measurement procedure with the adjustable pulse generator unit (76);

-applying the input signal (x [ n ]) to the first ultrasonic transducer (11);

-receiving a third response signal to the input signal (x [ n ]) at the second ultrasonic transducer (13);

-converting the third response signal into a third digitized response signal (y) using the analog-to-digital converter (64) ₁₂ '[n])；

-performing a reverse measurement procedure with the adjustable pulse generator unit (76);

-applying the input signal (x [ n ]) to the second ultrasonic transducer (13);

-receiving a fourth response signal to the input signal (x [ n ]) at the first ultrasound transducer (11);

-converting the fourth response signal into a fourth digitized response signal (y) using the analog-to-digital converter (64) ₂₁ '[n])；

-from said third digitized response signal (y) by calculating a convolution ₁₂ ') and has filter taps (h) _MF,12 [n]) Derives a first correlation signal (r) ₁₂ ) Said deriving comprising inverting said third digitized response signal (y) with respect to time ₁₂ ') or with filter taps (h) _MF,12 [n]) The digital filter of (1);

-from said fourth digitized response signal (y) by calculating a convolution ₂₁ ') and has filter taps (h) _MF,21 [n]) Derives a second correlation signal (r) ₂₁ ) Said deriving comprising inverting said fourth digitized response signal with respect to time or having filter taps (h) _MF,21 [n]) The digital filter of (a);

-from each correlation signal (r) by means of an interpolation method ₁₂ ，r ₂₁ ) Deriving time of flight (T) _abs,12 ，T _abs,21 ) (ii) a And

-subtraction (T) by previously derived time of flight _abs,12 -T _abs,21 ) To derive a time-of-flight difference (Δ T).

12. Electronic processing unit (53) according to claim 11, wherein there are filter taps (h) _MF,21 [n]) Is equal to having a filter tap (h) _MF,12 [n]) Digital soundA filter is applied.

13. The electronic processing unit (53) according to claim 12,

therein, having filter taps (h) _MF,21 [n]) Is a digital inverse response filter, and wherein the derivation of the inverse response filter comprises the steps of:

-performing a reverse training procedure with the adjustable pulse generator unit (76);

-applying the inverse training signal to the second ultrasound transducer (13);

-receiving an inverse response signal to the inverse training signal at the first ultrasound transducer (11);

-converting the inverse response signal into a digitized inverse response signal using an analog-to-digital converter (64);

-deriving a signal having FIR filter taps (h) from the digitized inverse response signal _MF,21 ) The digital inverse response filter of (1).

14. An electronic processing unit (53) for determining a flow rate of a fluid in a fluid conduit (12) using an ultrasonic time-of-flight flow meter, the electronic processing unit (53) being operable to:

-generating a training signal with an adjustable pulse generator unit (76);

-applying an input signal (x [ n ]) to a first ultrasound transducer (11), the first ultrasound transducer (11) being mounted to the fluid conduit (12) at a first location;

-receiving a response signal to the training signal at a second ultrasound transducer (13), the second ultrasound transducer (13) being mounted to the fluid conduit (12) at a second location, the second location being upstream or downstream of the first location with respect to the flow direction of the fluid;

-converting the response signal into a digitized response signal using an analog-to-digital converter (64);

-inverting the digitized response signal with respect to time to obtain an inverted digitized response signal;

-performing a pulse width modulation measurement procedure using the inverted digitized response signal,

-applying the pulse width modulated measurement signal to the first ultrasonic transducer (11);

-receiving a response signal to the pulse width modulated measurement signal at the second ultrasound transducer (13); and

-deriving a time of flight (Δ T) from the response signal.

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