RU2649050C1 - Microwave analogue phase shifter and the system containing it - Google Patents

Abstract

FIELD: instrument engineering.

SUBSTANCE: invention refers to the field of microwave analog devices for phase shifting, in particular to the microwave analog phase shifter, based on customized capacities. Phase shifter contains the four-terminal network with the radio frequency input port and an RF output port, the 3-dB hybrid bridge, which is configured with the possibility to electrically connect the input port and the output port with the two load ports. And two identical configurable reflective loads are connected with the loading ports of the 3-dB hybrid bridge and contain two controllable similar varactors each. Two independent analog control channels are provided for independent adjustment of each varactor for each reflective load of the adjustable reflective loads with the voltage that is applied to the corresponding DC ports.

EFFECT: technical result consists in reducing the average losses in the analog phase shifter and in weakening the internal resonance of the reflective loads.

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Description

FIELD OF TECHNOLOGY

The present invention relates to the field of microwave analog phase shifting devices, in particular to a microwave analog phase shifter based on custom capacitances.

BACKGROUND OF THE INVENTION

The main areas of application for microwave analog phase shifters are antenna arrays, for example array control and focus control for radar and communication antennas, and their use as microwave wireless power transfer (WPT) chargers. In addition, microwave analog phase shifters can be used as phase modulators to modulate the phase of the input radio frequency (RF or RF) signal.

The general requirements for microwave analog phase shifters are the following: 360 ° phase shift, low insertion loss, low transfer coefficient ripple, compact size, PCB integration capability, and low cost.

Customizable elements that are traditionally used in microwave analog phase shifters include semiconductor and dielectric varactors and piezoelectric capacitors. The most common design of analog phase shifters with adjustable capacities includes a 3-dB hybrid bridge (dual-loop directional bridge, hybrid ring bridge, Lange bridge, etc.) with a custom reflective load. Specified custom reflective load typically contains two custom capacitors, which are configured simultaneously by applying a variable DC voltage. As a result, losses occur that are caused by internal resonance in the structure of the reflective load. Resonant loss leads to a high attenuation of the RF output signal and spurious amplitude modulation. Such phenomena impair the performance of the system, i.e. reduce the gain (when used in antenna arrays), distort the spectrum of the signal (when used in phase modulators), etc.

Known phase shifter disclosed in US 7,969,359 B2. US-359 discloses a phase shifter comprising a hybrid bridge with a grounded shield, including differential coplanar strip lines arranged one above the other using different metal layers, so that the bond occurs in a vertical plane; and terminal reflective loads connected to the hybrid bridge such that when the hybrid bridge is connected to the terminal reflective loads, a phase shifter is formed, the terminal reflective loads each include a parallel LC circuit. The disadvantages of this known solution are high resonant losses, high ripple and high sensitivity of the phase response to voltage changes.

The phase shifter disclosed in US 6,710,679 B2 is also known. Mentioned document US-679 discloses a 360-degree analog dielectric varactor phase shifter including a 180-degree analog phase shifter on a hybrid ring bridge and a 180-degree digital phase shifter on switched lines, said digital phase shifter including first and second microstrip lines connected to each other through capacitors. One of the above microstrip lines serves as a 180 degree phase shift line, and the other microstrip line serves as a reference line. The disadvantages of this known solution are a very bulky design and high resonant losses.

Thus, it is desirable to provide a solution having a simplified design, reduced dimensions with minimal loss, in order to maintain the same amplitude ripple and phase range of 360 °.

SUMMARY OF THE INVENTION

The aim of the present invention is to reduce the average loss in the analog phase shifter and weaken the internal resonance of the reflective loads in order to reduce ripple during transmission. To achieve this goal, the problem of intrinsic resonance of reflective loads must be solved.

In order to overcome, at least in part, the disadvantages of the prior art and to solve the above problem, in accordance with one aspect of the invention, there is provided a microwave analog phase shifter, said phase shifter comprising a four-terminal network (two-port circuit or two-input circuit) with a radio frequency input port and a radio frequency output port A 3-dB hybrid bridge configured to electrically connect the input port and output port with two load ports, two of which are identical to traivaemye reflective load connected Terminating ports 3-dB hybrid bridge, with two independent control channel provided for independent tuning of the tunable reflector load voltage applied to the respective DC-ports.

The hybrid bridge is implemented as a microstrip or coplanar two-loop directional bridge, and two similar reflective loads are connected to the load ports of the bridge, while the said load ports are connected to the input port of the bridge. Each reflective load consists of two similar varactor diodes (also referred to as varactors) with a microstrip or coplanar transmission line with a given length and impedance between them, as described in detail below. Each varactor forms a varactor segment, which includes the specified varactor and a segment of the tuning microstrip or coplanar line, short-circuited or open at its end and connected to the varactor. Each varactor has its own control DC port, connected to it through a DC filter (DC filter). In the case where the varactor training lines are open, the DC port is connected to these lines, while the transmission line between the two varactors is shorted through another DC filter. In the other case, when the training lines are shorted, each varactor is offset by a separate DC port connected to the transmission line between the two varactors. In this case, the two DC ports must be isolated by means of an additional blocking capacitance inserted into the gap in the line. In other words, an interlock capacitor must be provided, connected in series with the transmission line. The two varactors are independently tuned to achieve the best performance in terms of transmission loss.

In one embodiment, a phase shifter is proposed, the structure of the device being the same as in the previous embodiment, and further includes a correction resistor connected in parallel at the point where one of the varactors is connected to the transmission line between the varactors. Additionally, the structure may include a matching open loop or line connected in parallel at the point where another varactor (closest to the bridge) is connected to the transmission line between the varactors. Preferably, the correction resistor is connected in series with another blocking capacitor. The two varactors are independently tuned to achieve the best performance in terms of transmission loss.

In another aspect of the invention, there is provided a microwave emitting system including a microwave analog phase shifter according to the invention. In a preferred embodiment of the above system, the phase shifter is integrated into an 8 × 8 transmit antenna array configured to be autofocused at the receiver.

The main technical result achieved in the present invention is to reduce and correct (equalize) the loss of transmission coefficient when using the proposed device. The above result provides an increase in the data transfer rate for communication systems, an increase in the range for radar systems and an increase in the energy transfer efficiency for WPT systems. The proposed analog phase shifter has two independent control channels, and each reflective load capacitance is independently configured. Consequently, the internal resonance can be attenuated by sequentially manipulating both channels using a control program that is optimized by analyzing the phase and transmission coefficient loss of control voltages. The reduction in losses leads to an increase in the efficiency of the proposed device, namely, a higher data transfer rate and longer operating ranges can be provided. In addition, reduced ripple does not cause stray modulation. Therefore, when using the invention provides improved signal integrity.

Other advantages of the proposed invention are described and explained in the following detailed description.

BRIEF DESCRIPTION OF THE DRAWINGS

The present invention will be explained in more detail in the following description with reference to the attached drawings, in which:

FIG. 1 shows a block diagram of a phase shifter according to the present invention.

FIG. 2 schematically shows a microstrip topology of a phase shifter according to one preferred embodiment of the present invention.

FIG. 3 shows an electrical diagram of a custom open-loop reflective load according to one preferred embodiment of the proposed phase shifter.

FIG. 4 shows an electrical diagram of a tunable reflective load with shorted lines according to another preferred embodiment of the proposed phase shifter.

FIG. 5 shows an electrical circuit of a tunable reflective load with shorted lines and a correction resistor according to yet another preferred embodiment of a phase shifter.

FIG. 6 shows a two-dimensional representation of the amplitude of the reflection coefficient for an approximate reflective load as a function of C ₁ , C ₂ .

FIG. 7 shows the amplitude of the reflection coefficient for an approximate reflective load as a function of two control voltages when both capacitances are tuned simultaneously.

FIG. 8 shows the amplitude of the reflection coefficient for an approximate reflective load as a function of two control voltages when both capacitances are independently tuned using the optimal tuning path with and without a correction resistor.

FIG. 9 shows the measured amplitude of the gain for the prototype full-range phase shifter versus frequency and control voltages.

DESCRIPTION OF THE PREFERRED EMBODIMENTS OF THE INVENTION

FIG. 1 shows a block diagram of a phase shifter according to the present invention. A microwave analog phase shifter 10 is proposed, comprising a four-terminal network with a radio-frequency input port 1 and a radio-frequency output port 2, 3-dB hybrid bridge 3, configured to electrically connect the input port 1 and output port 2 with two load ports, with two identically adjustable reflective loads 4 , 5 are connected to the load ports of the 3-dB hybrid bridge 3, while two independent control channels 6, 7 are provided for independent tuning of custom reflective loads 4 , 5 by voltage applied to the corresponding DC ports 8, 9.

Each reflective load should ideally manifest itself as an ideal reflector, i.e. it should have a constant amplitude of the reflection coefficient close to 1 (0 dB), and a phase that varies from 0 to 360 degrees or from 0 to -360 degrees. To achieve such operating parameters, each reflective load contains at least two tunable capacitors. The phase of the reflection coefficient can be changed by changing the capacitance of each capacitor. Unfortunately, each capacitor (semiconductor, dielectric varactor, etc.) at microwave frequencies has some stray losses. Therefore, the amplitude of the reflection coefficient also changes during the phase shift. This spurious amplitude modulation (ripple) is well known for simple reflective structures and causes resonant losses during transmission in the phase shifter, when the reflective load with two capacitors is used to realize a phase shift of 360 degrees using only one control voltage, i.e. when both capacitors are tuned at the same time. The present invention aims to reduce the average loss of reflective load and spurious ripple by modifying the structure of the reflective load and implementing the optimal algorithm for controlling capacitors.

In FIG. 2 schematically shows a microstrip topology of a phase shifter according to one preferred embodiment of the present invention. Presented microstrip phase shifter comprises a microstrip input line 201, output microstrip line 202 to the characteristic impedance Z _0, double stub 3-dB hybrid directional bridge 203 with two pull-up ports 204, 205 configured to couple with identical reflective loads 4, 5. The characteristic impedance is equal to Z ₀ for all ports 3-dB hybrid bridge. Further, each load port 204, 205 of the bridge 203 is connected to the input of the reflective load with wave impedance Z ₀ . Thus, the load ports 204, 205 and the inputs of the reflective loads are formed so as to have a single section of the microstrip line. Each reflective load 4, 5 additionally contains one varactor 243, 244 with a capacitance C _var2 connected in parallel to the input microstrip line. This varactor 243, 244 is connected to another varactor 241, 242 with a capacitance C _var1 through a quarter-wave microstrip line, also referred to as transmission line 245, 246. Each varactor 241, 242 and 243, 244 contains corresponding tuning microstrip open lines 214, 215 and 212 213 connected in series with it to form a custom varactor segment. These lines are used to obtain the desired phase shift through each varactor segment. Typically, a 180 degree phase shift is required from each varactor segment, and the lengths of the training lines and wave impedances are selected based on the ability to configure the varactors. The impedance of the transmission lines 245, 246 is usually chosen so as to match the reflective load with the bridge ports, i.e. to achieve the smallest losses and ripples. DC filters 216, 217, 218, 219, 219 are used to provide independent control channels 6 and 7 of the varactors by applying voltage to the two DC ports 208, 209. In this preferred embodiment, the filters are implemented as a microwave bandpass filter tuned to f ₀ , where f ₀ is the center frequency of the microwave operating frequency range. It should be understood that other possible filters having the same performance characteristics may be used. To provide direct current (DC-ground) to other varactor contacts, a short-wave quarter-wave cable 220 is used to ground bridge 3 along with reflective loads 4 and 5.

The described structure can be manufactured on standard microwave substrates, for example, Teflon-based substrates or alumina substrates, using existing technologies: etching of printed circuit boards, vacuum deposition, low-temperature co-fired ceramics, etc. The microstrip topology can be directly integrated into some topology of the microwave system as part of it or it can be implemented as a separate device with its own case and connectors (for example, SMA connectors).

The presented sample topology contains two separate DC ports. In principle, the invention may include three different reflective load structures, as described in detail below.

In FIG. 3 schematically shows an electric circuit of an adjustable open-loop reflective load according to one preferred embodiment of the proposed phase shifter.

A custom reflective load 304 in accordance with the embodiment of FIG. 3 contains two similar adjustable capacitances C _var1,2 (for example, varactors 341, 343) separated by a quarter-wave transmission line 345 with a wave impedance of 2Z ₀ , where Z ₀ is the aforementioned reference (base or reference) wave impedance. Each varactor 341, 343 has a similar training open line 314, 312 with a length and impedance selected to provide a 180-degree phase shift from each varactor segment. This choice depends on the setting range (C _max and C _min ) of the varactor and the value of Z ₀ . The required calculation formula can be obtained from the condition of a 180-degree phase shift:

(1)

(one)

where X _s is the total reactivity of the tuning open line along with the parasitic inductance of the varactor. The length of the open tuning lines are usually lies between λ _0/4 and λ _0/2, where λ ₀ - wavelength transmission line at the center frequency f _0.

By connecting the two DC ports 308, 309 with the training open lines 314, 312 through the corresponding DC filters 316, 318, significant isolation can be achieved between the DC ports 308, 309. In this case, the transmission line 345 must be grounded through another filter 320 constant current to maintain a reference DC voltage level at the varactor contacts connected to the transmission line 345. The above-described electrical circuit of the reflective load 304 according to FIG. 3 corresponds to the schematic topology illustrated in FIG. 2.

FIG. 4 schematically shows an electric circuit of a tunable reflective load with shorted lines according to another preferred embodiment of the proposed phase shifter.

A custom reflective load 404 in accordance with the embodiment of FIG. 4 contains the same microwave components as in the embodiment of FIG. 3. However, the training lines 412, 414 are shorted. The length and wave resistance of the tuning shorted lines 412, 414 are determined by the formula (1), where X _s is the total reactivity of the tuning shorted line along with the parasitic inductance of the varactor. In this case, the DC voltage should be applied to the varactors 441, 443 connected to the transmission line 445 through the corresponding DC filters 418, 416. Therefore, in order to guarantee isolation of the DC ports 408, 409, an additional DC blocking capacitor 450 is connected in series with the transmission line 445. It should be noted that the reactivity of the blocking capacitor 450 at a frequency f _{0 of the} working microwave range is at least an order of magnitude less than Z ₀ .

FIG. 5 schematically shows an electrical circuit of a tunable reflective load with shorted lines and a corrective parallel resistor in accordance with yet another preferred embodiment of a phase shifter.

The electrical circuit of the reflective load 504 in accordance with the embodiment of FIG. 5 contains the same components as in the embodiment of FIG. 4. Additionally, the reflective load 504 includes a resistor 555 having a resistance R _p , the resistor 555 is connected in parallel with the varactor 541. In addition, a matching open loop or line 560 is connected (a) in parallel with the varactor 543. The value of R _p is selected for correction reflection loss in accordance with the formula:

(2)

(2)

where R _v is the active resistance of varactor 541 at the microwave frequency.

A correction resistor 555 is connected in series with an additional DC blocking capacitor 551. It should be noted that the resistance of the blocking capacitor 551 at a frequency f _{0 of the} working microwave range is at least an order of magnitude less than R _p . An open line 560 is used to further correct transmission loss by adjusting the impedance of the reflective load. The length and impedance of the open line 560 are selected from the resonance equation:

(3)

(3)

where X _p is the reactance of the open line 560.

Thus, in any of the above embodiments, each capacitance can be adjusted independently, as a result of which the reflective load can be adjusted in a more versatile manner, and the tuned capacitances can operate in a less stressed mode.

With reference to FIG. 6-8, the advantage of the proposed invention is that the internal resonance of the reflective load 4, 5 can be effectively attenuated by sequentially manipulating both control channels 6, 7. It should be understood that the reflectance of the reflective load 4, 5 is a two-dimensional function of C _var1 and C _var2 .

An example of such a two-dimensional map representation is illustrated in FIG. 6. This map was built when modeling the reflective load 404 with the following parameters: Z ₀ = 50 Ohm, C _min = 0.15 pF, C _max = 0.6 pF, R _v = 4 Ohm at f ₀ = 5.8 GHz for the chain in accordance with the embodiment illustrated in FIG. 4. The resonance region 600 is shown on the map as the darkest region.

When setting the capacitance from the maximum value to the minimum value, the reflective load 404 changes the state from the upper right corner of the map (point 601) to the lower left corner (point 602). If all parameters are selected in accordance with the above formulas (1) - (3), the total phase shift during movement along any path between these two points is 360 degrees. It should be noted that each path has its own average losses and ripples. The idea is to choose a path with minimal average losses and ripples, i.e. to avoid the resonance region 600, where the average losses and ripples are maximum.

If a known algorithm is used, i.e. Since both C _var1 and C _{var2 are} configured at the same time, the circuit is tuned along the diagonal path (dashed line 603, the direction of adjustment is shown by the arrow). In FIG. 6, it is clearly shown that the circuit, when tuned, passes through or near the resonance region 600. On the other hand, when each capacitance is independently tuned, the resonance can be circumvented, for example, by the lower right corner (point 604), as shown in FIG. 6 as an optimal path (dashed lines 605 and 606, the direction of adjustment is indicated by an arrow).

For some other cases, the adjustment path may be formed as a quarter of a circle passing through the upper right point 601 and the lower left point 602 and above the lower right point 604 of the loss map (see dash-dot curve 607 in Fig. 6).

More specifically, as a non-limiting example of a control algorithm, an algorithm can be used as described below: C _{var1 is} adjusted from C _max to C _min (corresponds to line 605 from point 601 to point 604), while C _{var2 is} supported in C _max , and then C _{var2 is} adjusted from C _max to C _min , while C _{var1 is} supported at C _min (corresponds to line 606 from point 604 to point 602). Another way is to perform a phase shift in accordance with, for example, line 607 from 601 to 602 while simultaneously configuring each of C _var1 and C _var2 independently.

It is understood that any suitable means known in the art can be used to implement the above algorithm. By way of non-limiting example, such a tool includes program code stored on a computer-readable medium, said program code including instructions for implementing DC control of a varactor.

Varactor DC control can be implemented using, for example, a standard digital-to-analog converter (DAC). In this case, the digital controller or other microprocessor unit used to control the phase shifter can be configured to send a digital code to the DAC input. The DAC can be further configured to generate an analog output voltage for application to the DC ports of the proposed phase shifter. For most applications, an 8-bit DAC per DC channel is sufficient for precise phase control.

The simulation results for the reflection coefficient amplitude as a function of two DC control voltages while simultaneously adjusting from C _max to C _min in accordance with a known algorithm (path 603 shown in FIG. 6) are illustrated in FIG. 7 of curve 703. In particular, at the initial time 701, the control voltages U _DC1 , U _DC2 applied to the control DC ports are minimal, while C _var1 , C _{var2 are} maximum, and at the final instant 702: U _DC1 , U _DC2 = max, C _var1 , C _var2 = min.

The simulation results for the control algorithm proposed above are presented in FIG. 8 by curves 803 and 830. In particular, at the initial time 801, the control voltages U _DC1 , U _{DC2 are} minimum, while C _var1 , C _{var2 are} maximum, at an intermediate time 804 (when the tuning path shown in Fig. 6, changes direction from line 605 to line 606 at point 604) U _DC1 = max, U _DC2 = min, C _var1 = min, C _var2 = max; and at the final moment of time 802: U _DC1 , U _DC2 = max, C _var1 , C _var2 = min.

The average losses for the case without a correction resistor (see curve 803) were improved by 0.7 dB, and the ripple level was reduced from 1.6 dB to 0.6 dB. In the case of the correction resistor according to the embodiment of FIG. 5 (see curve 830), the average loss was improved by 0.35 dB, and the ripple level was reduced from 1.6 dB to 0.1 dB.

Thus, in the present invention, the resonance loss is attenuated by selecting the optimal control algorithm, as a result of which the average reflection loss and transmission ripple of the reflective load are reduced.

The proposed phase shifter in accordance with the embodiment shown in FIG. 2, was tested under the following conditions: Rogers substrate RO4003C, varactor diodes MA-COM MA46H120, voltage range 1-12 DC, center frequency 5.8 GHz. Measurements were performed using an Agilent PNA-X vector network analyzer. FIG. 9 shows the measured phase shifter gain versus frequency and control voltages. Transmission loss is less than 2 dB, for the 360 ° phase range, ripple value <0.7 dB, phase shifter geometrical area is approximately 0.5 × 0.5 wavelength ² , Q factor is 211 deg / dB, these parameters are significantly improved in comparison with the corresponding parameters according to the prior art. For comparison, the simulated transmission loss for a solution according to US-679 is estimated at about 3-5 dB, the geometric area is about 1 × 0.5 wavelength ² , and the quality factor is estimated at about 100-130 deg / dB for a fairly complex design.

As a non-limiting example, the following is the use of the proposed phase shifter in a microwave emitting system. One example is the WPT system, where the proposed phase shifter is integrated into the 8 × 8 transmit antenna array with autofocus capability on the receiver. In this system, the method of generating the desired phase distribution over the lattice aperture can be implemented using the proposed phase shifter.

Another exemplary system is a satellite tracking receiver, where the proposed phase shifter is used to adjust the phase during satellite tracking by automatically controlling a four-antenna interferometer.

As described above, the proposed invention has a number of positive effects for electronic systems, in particular, providing a higher energy potential and higher signal quality. Therefore, the proposed solution can be advantageously used in radar antenna arrays, base station antennas, satellite communications systems, radio frequency integrated circuits (RFIC), etc. In addition, you can design microwave analog phase shifters based on custom capacities (with operating frequencies up to 100 GHz) for use in IoT sensors, Wi-Fi and mobile communications, long-range WPT systems, etc.

It should be understood that any of the above exemplary systems may include a storage device that is a memory, such as non-volatile and writable memory or flash memory. The storage device may include other types of storage devices, such as a magnetic storage device or a hard disk. The storage device may be write-once memory. The system may further comprise a microprocessor configured to execute program code for implementing the above algorithms and methods. The program code or software may be stored in an appropriate suitable storage device, such as a hard disk, diskette, memory, etc. The software may be sent as a signal by wire or wireless or using a data network such as the Internet.

Finally, it should be understood that the devices, systems, and methods described above are exemplary embodiments that should be considered in all respects as illustrative but not limiting.

Claims (7)

1. A microwave analog phase shifter comprising a four-terminal with a radio frequency input port and a radio frequency output port, a 3-dB hybrid bridge configured to electrically connect the input port and the output port with two load ports, with two identical adjustable reflective loads connected to the load ports 3- dB hybrid bridge and contain two controlled similar varactors each, while there are two independent analog control channels for independent us triples of each varactor of each reflective load of adjustable reflective loads by voltage applied to the corresponding DC ports. 2. The phase shifter according to claim 1, wherein two similar varactors of each adjustable reflective load are separated by a quarter-wave transmission line, the varactors having similar tuning open lines connected to them to form a varactor segment, the DC ports being connected to open lines through DC filters , and the quarter-wave transmission line is grounded through another DC filter. 3. The phase shifter according to claim 1, in which two similar varactors of each adjustable reflective load are separated by a quarter-wave transmission line, the varactors have similar tuning shorted lines connected to them to form a varactor segment, the DC ports being connected to the varactors through direct current filters, and a blocking capacitor is connected in series with the quarter-wave transmission line. 4. The phase shifter according to claim 2 or 3, in which the tuning open lines or tuning shorted lines have a length and impedance selected to obtain a 180-degree phase shift from each varactor segment. 5. The phase shifter according to claim 2 or 3, in which each adjustable reflective load further comprises a correction resistor connected in parallel with one of the varactors, and a matching open line connected in parallel with another varactor, the correction resistor connected in series with another blocking capacitor. 6. A microwave emitting system, including a microwave analog phase shifter according to any one of paragraphs. 1-5. 7. The system according to claim 6, in which the phase shifter is integrated into a transmitting antenna array of dimension 8 × 8, configured with autofocus on the receiver.

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