WO2020198170A1

WO2020198170A1 - Apparatus and systems for beam controllable patch antenna

Info

Publication number: WO2020198170A1
Application number: PCT/US2020/024329
Authority: WO
Inventors: Tatsuo Itoh; Haozhan TIAN; Kirti DHWAJ
Original assignee: The Regents Of The University Of California
Priority date: 2019-03-22
Filing date: 2020-03-23
Publication date: 2020-10-01
Also published as: WO2020197977A1

Abstract

Apparatus and systems for a beam controllable patch antenna are described. Metal vias can be introduced to create two coupled half-mode cavities in the patch antenna. The mutual coupling between the two cavities can make the phase difference as a function of the operating frequency.

Description

Apparatus and Systems for Beam Controllable Patch Antenna

STATEMENT OF FEDERALLY SPONSORED RESEARCH

[0001] This invention was made with government support under Grant Number 1610892, awarded by the National Science Foundation. The government has certain rights in the invention.

CROSS-REFERENCE TO RELATED APPLICATIONS

[0002] The current application claims the benefit of and priority under 35 U.S.C. § 1 19 (e) to U.S. Provisional Patent Application No. 62/822,421 entitled“Beam Controllable Patch Antenna” filed March 22, 2019. The disclosure of U.S. Provisional Patent Application No. 62/822,421 is hereby incorporated by reference in its entirety for all purposes.

FIELD OF THE INVENTION

[0003] The present invention generally relates to apparatus and systems for beam controllable patch antennas; and more particularly to apparatus and systems that utilize metal vias in the antennas to achieve beam scanning.

BACKGROUND

[0004] Rectangular patch antenna has been widely used in many applications due to its simplicity and low cost as well as the compact configuration. Beam-scanning antennas are desired for many applications. One example is the leaky-wave antenna. The radiation for this type of antenna is realized by using a phased array. However, the structure of an array can be bulky and complex.

BRIEF SUMMARY OF THE INVENTION

[0005] Apparatus and systems in accordance with various embodiments of the invention enable the design and/or implementation of beam controllable patch antennas. Many embodiments provide beam scanning in a single-patch antenna. The antenna design in accordance with many embodiments can be implemented by putting vias to create two coupled half-mode cavities in a rectangular patch antenna. The phase difference may be manipulated between the two equivalent magnetic currents and thus steer the beam through this design. Circuit model can be built in order to control the coupling and achieve the wide matching bandwidth. Measured and simulated results show that the beam may scan over a large range as the frequency changes while maintaining high radiation efficiency. The beam-scanning antenna in accordance with many embodiments has advantages of compact size and low cost. It can be scaled to high-frequency domain due to the simplicity. The phase manipulation within a single patch can potentially benefit many patch array designs.

[0006] Many embodiments describe a rectangular patch antenna comprising an inset feed; a rectangular flat metal patch; a flat metal ground plane; a substrate between the flat metal patch and the flat metal ground plane; and a plurality of metal vias connecting the flat metal patch to the flat metal ground plane, where the plurality of metal vias are arranged in a line that divides the patch antenna into two half-mode cavities and are positioned slightly off center so that the two half-mode cavities are not identical in size, where each half-mode cavity acts as a resonator.

[0007] In one embodiment of the invention, the rectangular patch antenna further comprises a circuit, where the circuit is configured to provide a source signal, where the patch antenna emits a certain pattern of radiation based upon the source signal.

[0008] In a further embodiment, the emitted radiation is beam scanning.

[0009] In another embodiment, the beam scanning is a function of frequency.

[0010] A still further embodiment includes the substrate comprising of a dielectric material.

[0011] In still another embodiment, the patch comprises of a conductive material.

[0012] In a yet further embodiment, the ground plane comprises of a conductive material.

[0013] In yet another embodiment, the dielectric material has a dielectric constant of about 2.2.

[0014] In a further embodiment, the dielectric material has a loss tangent of about 0.001 . [0015] In a further embodiment again, the rectangular patch antenna further comprising a gap, wherein the vias are placed in a line from one side to the other side of the antenna and the gap is an enlarged spacing omitting vias in the center of the line.

[0016] In a further additional embodiment, the gap is about 3.5 millimeter.

[0017] In another additional embodiment, the patch antenna has a width of about 28 millimeter.

[0018] In a still yet further embodiment, the patch antenna has a length of about 19.5 millimeter.

[0019] In still yet another embodiment, the patch antenna has a height of about 0.79 millimeter.

[0020] In a still further embodiment again, the metal vias have a spacing of about 1.6 millimeter.

[0021] In still another embodiment again, the metal vias have a diameter of about 0.8 millimeter.

[0022] In a still further additional embodiment, the antenna works in coupled modes.

[0023] In a still another embodiment, the coupling between the two half-mode cavities adjusts the phase difference as a function of the operating frequency.

[0024] In a yet further embodiment, the coupling takes place at the gap.

[0025] In yet another embodiment, the coupling is either even mode or odd mode.

[0026] Additional embodiments and features are set forth in part in the description that follows, and in part will become apparent to those skilled in the art upon examination of the specification or may be learned by the practice of the disclosure. A further understanding of the nature and advantages of the present disclosure may be realized by reference to the remaining portions of the specification and the drawings, which forms a part of this disclosure.

BRIEF DESCRIPTION OF THE DRAWINGS

[0027] The description will be more fully understood with reference to the following figures, which are presented as exemplary embodiments of the invention and should not be construed as a complete recitation of the scope of the invention, wherein: [0028] Fig. 1A - 1 C illustrate a beam controllable patch antenna in accordance with an embodiment of the invention.

[0029] Fig. 2A illustrates demonstration of coupled E-field distribution in symmetric structure at even mode, where the dashed line represents the via wall with an open gap for coupling in accordance with an embodiment of the invention.

[0030] Fig. 2B illustrates demonstration of coupled E-field distribution in symmetric structure at odd mode, where the dashed line represents the via wall with an open gap for coupling in accordance with an embodiment of the invention.

[0031] Fig. 3 illustrates an equivalent lumped circuit model where radiation loss of the two cavities is represented by Gi and G2, respectively, in accordance with an embodiment of the invention.

[0032] Fig. 4 illustrates simulated, measured, and circuit-model S11 response in accordance with an embodiment of the invention.

[0033] Fig. 5 illustrates simulated vector distribution of the electric field in the substrate on the cross section, where three frequencies refer to even mode, the mode at the intermediate peak frequency, and odd mode in accordance with an embodiment of the invention.

[0034] Fig. 6 illustrates simulated copolarized radiation patterns on the E-plane, where the patterns are normalized to 1 and shown in absolute value in accordance with an embodiment of the invention.

[0035] Fig. 7 illustrates realized boresight gain versus frequency, where the measured gains are at 4.975, 5.03, and 5.165 GFIz, respectively, in accordance with an embodiment of the invention.

[0036] Fig. 8 illustrates simulated and measured radiation patterns on the E-plane at (a) 4.975 GFIz (even mode), (b) 5.03 GFIz (90° out of phase), and (c) 5.12 GFIz (odd mode), and the implementation of the design is shown in (d) in accordance with an embodiment of the invention.

[0037] Fig. 9 illustrates measured realized gain patterns in absolute value on the E- plane in accordance with an embodiment of the invention. DETAILED DESCRIPTION OF THE INVENTION

[0038] Turning now to the drawings, beam controllable patch antennas in accordance with various embodiments of the invention are illustrated. Many embodiments provide the ability for beam scanning in a single-rectangular-patch antenna realized by coupled modes. In some embodiments, the beam controllable patch antennas can exhibit radiated beams that scan as a function of frequency. In several embodiments, the beam controllable patch antennas may operate in different mechanism compared to a traditional beam-scanning antenna, examples include but are not limited to a leaky-wave antenna. As can readily be appreciated, the specific features used in a beam-scanning antenna in accordance with embodiments of the invention are largely tailored to the requirements of specific applications. Many embodiments of the patch antenna may manipulate the phase difference between the fields at different radiating slots of the antenna to reform the beam. In several embodiments, metal vias can be used to create coupled half-mode cavities in the patch antenna. The mutual coupling between the cavities can make the phase difference a function of the operating frequency. In several embodiments, two radiating slots can be implemented and hence two coupled half-mode cavities can be created in the patch antenna. In many embodiments, the beam controllable patch antennas can be made in compact sizes. In several embodiments, the beam controllable patch antennas can be made with low cost. In some embodiments, the beam controllable patch antennas can be scaled to high-frequency domain. Some embodiments include that the beam controllable patch antennas can be applied to patch arrays.

[0039] In many embodiments, the simulated beam may scan from broadside to about 43 degrees within about 3 dB gain band. In several embodiments, a patch antenna may realize about 4.55% bandwidth of Sn with center frequency at about 5.05 GHz by measurement. In certain embodiments, the beam may scan from about 14 degrees to about 34 degrees within the matching band, which may be confirmed by both simulation and measurement. The measured radiation efficiency within the matching band in certain embodiments may be comparable to a regular patch antenna. [0040] Rectangular patch antennas have been used in many applications due to its simplicity and low cost as well as compact configuration. The fundamental mode of operation for this antenna is TM- f₀₀, which offers the maximum electric fields with 180 degrees phase difference at two radiating slots. The radiation mechanism of a patch antenna can be explained from two perspectives. In the transmission line model, the radiation can be from in-phase fringing fields at two radiation slots. In the cavity model, it can be from two in-phase equivalent magnetic currents. Derived from either of the explanation, the radiation pattern of the patch antenna can be independent on the frequency as long as the operation mode remains the same.

[0041] Beam-scanning antennas, on the other hand, may be desired for many applications. For example, the leaky-wave antenna has a beam pattern that can scan in the space as the operating frequency changes. The radiation for this type of antenna can be realized by leaking energy gradually as the wave propagates along the structure. The beam-scanning angle may then be dictated by the frequency dependent propagation constant. In order to maintain a high radiation efficiency, the antenna may be longer than wavelength in general. Another antenna structure for beam-scanning is the phased array. The beam scanning can be realized by properly tuning the phase and amplitude of each element. However, the structure of a beam-scanning antenna array can be bulky and complex.

[0042] Accordingly, many embodiments provide coupled modes in a patch antenna to realize beam scanning. The phase difference of two radiation slots can be manipulated by the patch antenna design in accordance with various embodiments. In several embodiments, the beam may scan from broadside to about 43 degrees within about 3 dB realized boresight gain bandwidth in simulation. The implemented sample may realize about 4.55% 10 dB bandwidth of S_1:L with center frequency at about 5.05 GHz by measurement. The patch antenna design in accordance with several embodiments may share the simple and compact configuration of a patch antenna while also maintaining a high radiation efficiency. The operation mechanism along with the results of simulation and measurement in accordance with various embodiments are discussed in detail below. Structure

[0043] Many embodiments include structures of beam scanning patch antennas. One structure of beam scanning patch antennas in accordance with various embodiments of the invention is illustrated Fig. 1A-1 C. Fig. 1A illustrates a top view of the beam scanning patch antenna. Fig. 1 B illustrates a side view of the antenna. Fig. 1 C illustrates a bottom view of the antenna. The illustrated patch antenna includes ground plane 150, substrate layer 130, and patch layer 160.

[0044] The ground plane 150 and patch 160 can be made of a conductive material. The substrate 130 can be made of a dielectric material. For example, the substrate 130 can be made of (but are not limited to) Rogers RT/Duriod 5880, with dielectric constant of 2.2, loss tangent of 0.001 , and height of 0.79 mm. In other embodiments, the substrate can be air or other non-solid substance with or without spacers acting as mounts. As can readily be appreciated, any of a variety of substrates can be utilized as appropriate to the requirements of specific applications in accordance with various embodiments of the invention. The antenna may be based on a rectangular patch antenna with an inset feedline 170. The size of the antenna can be characterized by the width w 180 and total length 190 by the equation, Z_totai = 21 + M . In some embodiments, the width of the antenna is of about 28 mm. In several embodiments, the length l is of about 9.5 mm, M is about 0.5 mm. In several embodiments, the ground plane extends beyond the edges of the patch. As can readily be appreciated, any of a variety of antenna and ground plane widths and lengths can be utilized as appropriate to the requirements of specific applications in accordance with various embodiments of the invention.

[0045] Several embodiments provide metal vias 140 around the center connecting the conductive patch 160 with the ground 150. These metal vias 140 can connect portions of the ground and patch, through the substrate 130, in places that are not normally connected in a conventional patch antenna. In some embodiments, the metal vias 140 form a line starting from one side of the patch antenna to the other opposing side and thereby bisect the patch antenna. In further embodiments, the joined sides are immediately adjacent to the feedline. This line of metal vias can behave like a metal wall for the electromagnetic field in the substrate, which divide the patch cavity into two parts. Each part can be a resonator, which supports a half-mode of TM100, such as the mode in a planar inverted-F antenna. A gap 1 1 1 can be opened on the interior wall of the line of metal vias by omitting or leaving out some metal vias in the line. In several embodiments, the gap is located in the middle of the line of metal vias. In some embodiments, the diameter of the via is of about 0.8 mm. In several embodiments, the spacing of the vias is about 1 .6 mm. As can readily be appreciated, any of a variety of metal via diameter and spacing can be utilized as appropriate to the requirements of specific applications in accordance with various embodiments of the invention. In additional embodiments, vias can be constructed of other conductive material.

[0046] The gap can function as a coupling iris. In some embodiments the width of the gap s is of about 3.5 mm. As can readily be appreciated, any of a variety of gap widths can be utilized as appropriate to the requirements of specific applications in accordance with various embodiments of the invention.

[0047] While various structures for beam scanning patch antennas are described, any variety of structures that realize beam scanning in a patch antenna can be utilized as appropriate to the requirements of specific applications in accordance with various embodiments of the invention. Processes for characterizing beam scanning patch antennas in accordance with various embodiments of the invention are discussed further below.

Operation

[0048] Processes in accordance with various embodiments of the invention may utilize the operation modes of beam scanning patch antennas. As is discussed further below, any of a variety of coupling modes can be utilized as appropriate to the requirements of specific applications in accordance with various embodiments of the invention.

[0049] In many embodiments, the mutual coupling between two cavities may result in two eigen-mode solutions, which can be called even mode, due to electric coupling, and odd mode, due to magnetic coupling. An analysis of a symmetric structure in accordance with an embodiment of the invention is illustrated in Figs. 2A and 2B, where the dashed line 210 is at the center and thus the two half-mode cavities can be identical. The E-field distributions of even and odd modes are demonstrated in Fig. 2A and Fig. 2B, respectively. The structure can then be modeled by an array of two equivalent magnetic currents with the same amplitude but different phases. In the xz plane, the array factor (AF) is given by the equation:

AF = 2 cos ( k₀dsinQ + Df) (1 )

where k₀ is the propagation constant in free space, d is the separation distance, and Df is the phase difference of the two magnetic currents. The pattern of one of the magnetic currents on the E-plane can be approximately omnidirectional, and thus, beam pattern of the array on the E-plane can be only dependent on AF. In the even-mode case where Df is of about 180°, there can be a null at broadside Q of about 0° in the far-field pattern due to the out-of-phase cancellation. Meanwhile for the odd mode where Df is of about 0°, the structure may radiate like a rectangular patch antenna where the beam peak is at the broadside. In many embodiments, as frequencies can change from even mode to odd mode, the phase difference Df can gradually shift from about 0° to about 180° due to the coupling. In consequence, the beam peak can scan away from broadside according to equation (1 ).

[0050] Flowever, in certain embodiments, far-field cancellation and two equal lobes in the radiation pattern of even mode may be undesired since one main beam scanning around the broadside is the goal. Thus, certain embodiments provide an asymmetry to the model by putting the vias slightly off the center so that the two cavities are not identical. In such embodiments, the fields coupling at the iris can pass through different electrical lengths to reach the two radiating edges, respectively. The phase difference at even mode may be close to about 180° but not exactly at 180°, which may avoid the strong broadside cancellation, and may lead to one main beam instead of two equal lobes at even mode in accordance with embodiments. In the meantime, the beam peak may also shift slightly away from the broadside at odd mode due to the asymmetry. In several embodiments, the even mode or odd mode can be defined by the type of coupling that happens at the iris, not by the phase difference between the two radiation slots. In such embodiments the even and odd modes can be still two eigenmode solutions of an asymmetric model. In several embodiments, there may still be a minor lobe and a null in the radiation pattern scanning with frequency as predicted by equation (1 ) above. In fact, certain embodiments may benefit from the scanning null in some applications such as cognitive radio. As can readily be appreciated, any of a variety of scanning null applications can be utilized as appropriate to the requirements of specific applications in accordance with various embodiments of the invention.

[0051] While various operation modes for beam scanning patch antennas are described, any variety of operation modes that realize beam scanning in a patch antenna can be utilized as appropriate to the requirements of specific applications in accordance with various embodiments of the invention. Processes for determining desired operation frequency and bandwidth based on a circuit model in accordance with various embodiments of the invention are discussed further below.

Filtering Antenna Design

[0052] Processes in accordance with various embodiments of the invention may utilize the designs of beam scanning patch antennas. As is discussed further below, any of a variety of design approaches can be utilized as appropriate to the requirements of specific applications in accordance with various embodiments of the invention.

[0053] Many embodiments may be designed such that they first generate a circuit model, as illustrated in Fig. 3, to meet the desired operation frequency and the bandwidth, as the design procedures of a two-pole passband filter. In this way, it may be determined how fast the beam scans with frequency and gets reasonable realized boresight gain within the band. With the values of the circuit model, the structure of the antenna can be designed in accordance with many embodiments of the invention. The dimension of the two half-mode cavities may be calculated following similar procedures for designing rectangular patches. In several embodiments, the inset feed may be designed to guarantee a desired input coupling which can lead to a good input matching. The gap size may be determined by the internal coupling coefficient extracted from the circuit model.

[0054] Unlike a traditional two-port filter, the model of the design in accordance with several embodiments may be built with only a source port but no load port. This is because the energy from the input source of this design may be radiated out directly to the free space instead of a load port as the traditional two-port passband filter. Flowever, load may still be necessary, so in many embodiments, the radiation conductance of the antenna may be regarded as equivalent load.

[0055] Fig. 3 illustrates an equivalent lumped circuit model in accordance with an embodiment of the invention. In Fig. 3, the equivalent lumped circuit model may be built for a fractional bandwidth of about 4.8% at the midband frequency of about 5.05 GHz. The shunt LC circuits may represent the resonance of the two half-mode cavities. G and G₂ may stand for the equivalent radiation conductances, which can be theoretically calculated by equation:

W

Gt =

120 l;

where l_ί is the resonant wavelength in each cavity, for / = 1 , 2. The interstage couplings can be modeled by the admittance inverters, whose values may be related to external quality factor and coupling coefficient k by:

= 2nfiCi for i = 1, 2. In Fig. 3, the radiation loss of the two cavities can be represented by G₁ and G₂ respectively. G₀ has a conductance of about 0.02 S. Input coupling J_{Q 1} is of about 1.035 x 10^-2 S. Inductor L₁ in the first half-mode radiating cavity has inductance of about 0.278 nH. Capacitor C₁ in the first half-mode radiating cavity has capacitance of about 3.575 pF. Radiation loss G is of about 4.040 x 10^-3 S. Internal coupling 2 is of about 3.938 x 10³ S. Inductor L₂ in the second half-mode radiating cavity has inductance of about 0.264 nH. Capacitor C₂ in the second half-mode radiating cavity has capacitance of about 3.771 pF. Radiation loss G₂ is of about 3.856 x 10^-3 S. By applying the methods of filter designs, the parameter values of the model can be calculated. Fig. 4 illustrates a response of the circuit model to compare with the one of simulation and measurement in accordance with an embodiment of the invention. [0056] In many embodiments, the structure of the antenna can be designed based on the values of the circuit model. The dimensions of the two half-mode cavities can be calculated following similar procedures for designing rectangular patches. Besides the fields of cavity mode, the fringing fields at the radiation slots can contribute to the shunt LC circuits in the circuit model as well. The inset feed can be designed to guarantee a desired external coupling, which may lead to a good input matching. The gap size can be determined by the internal coupling coefficient extracted from the circuit model by equation (4). With minor optimization to compensate the inaccuracy of the approximations, the structure may then be designed as illustrated in Fig. 1 in accordance with several embodiments of the invention.

[0057] While various approaches to design beam scanning patch antennas based on filtering antenna perspective are described, any variety of design approaches can be utilized as appropriate to the requirements of specific applications in accordance with various embodiments of the invention. Processes for characterizing beam scanning patch antennas properties by simulation and measurement in accordance with various embodiments of the invention are discussed further below.

SIMULATION AND MEASUREMENT

[0058] Processes in accordance with various embodiments of the invention rely upon the characterization of beam scanning patch antennas properties. As is discussed further below, any of a variety of simulation and measurement modes can be utilized as appropriate to the requirements of specific applications in accordance with various embodiments of the invention.

[0059] The simulated, measured, and circuit-model S_1:L responses may match well with each other, as shown in Fig. 4 in accordance with an embodiment of the invention. The solid line represents the measured response, dashed line represents the simulated response, and the dotted line represents the circuit model 5_X1 response. The reflection coefficient of the antenna can be measured by an Agilent™ 8510C vector network analyzer. The measured center frequency can be of about 5.05 GHz with about 10 dB fractional bandwidth of about 4.55%. The measured fractional bandwidth may be slightly reduced from the desired about 4.8% due to the fabrication errors. The two poles in the measured response can be at of about 4.975 and about 5.12 GHz, whereas the intermediate frequency peak can be at about 5.03 GHz.

[0060] Fig. 5 illustrates a simulated vector distribution of the electric field in the substrate on the cross section in accordance with several embodiments of the invention. In Fig. 5, the vector distribution of the E-field at these three frequencies are shown. The fields at about 4.975 and about 5.12 GHz may represent the even and odd-mode field distributions, respectively. The distributions can be apparently asymmetric because the two cavities may be designed to be nonidentical. This may be desired for the radiation purpose as mentioned earlier. At about 5.03 GHz, the fields at the two edges can be around 90° out of phase. The frequency at about 5.03 GHz refers to the mode at intermediate peak frequency. The field distributions for these three frequencies may strongly support that the phase difference at the two radiating slots changes with the operating frequency.

[0061] To demonstrate the beam-steering capability of the design in accordance with many embodiments, the simulated radiation patterns are plotted on the E-plane at three selected frequencies, as shown in Fig. 6 in accordance with various embodiments of the invention. The odd-mode frequency of about 5.12 GHz may be the frequency where maximum realized gain of about 8.12 dBi is achieved in simulation. While at about 4.92 GHz and about 5.25 GHz, the realized boresight gain may be about 3 dB lower than the maximum one, as shown in Fig. 7 in accordance with several embodiments of the invention. The beam peak may be at the broadside at about 5.25 GHz, and may be at about 43° at about 4.92 GHz, which may indicate a maximum about 43° of scanning range within about 3 dB gain band. The patterns may be normalized to 1 and plotted in absolute values to provide a clear view of the beam steering.

[0062] Fig. 7 shows the realized boresight gain versus frequency in accordance with various embodiments of the invention. The black circles represent measured results, and the empty circles represent simulated results. The measured gains are at about 4.975 GHz, about 5.03 GHz, about 5.12 GHz, and about 5.165 GHz, respectively. [0063] The measured and simulated patterns on the E-plane can compare closely for even-mode, 90-out-of-phase, and odd-mode frequencies, as shown in Figs. 8A-8D in accordance with several embodiments of the invention. The peak of the copolarized beam 810 can clearly scan from side to the center as the frequency changes. There may be a -25 dB null at 341 ° on the copolarized pattern at even mode 4.975 GHz, as shown in Fig. 8A. This may be the cancellation for the even-mode radiation as discussed previously. The null may scan to 324° as the frequency tuning to 90-out-of-phase 5.03 GHz in Fig. 8B, and may disappear at odd mode 5.12 GHz in Fig. 8C. The measured cross-polarized patterns on the E-plane 820 may all be below -20 dB, though they may be higher than the simulated ones 830 due to fabrication and measurement tolerances. The implementation of the design is shown in Fig. 8D.

[0064] The realized gain patterns at different frequencies on the E-plane can be measured and shown in Fig. 9 in accordance with an embodiment of the invention. The beam may scan from about 14° to about 34° as the frequency changes from about 5.165 GHz to about 4.975 GHz. The maximum realized gain of about 6.34 dBi can be achieved at about 5.12 GHz, the odd-mode frequency. At even-mode frequency of about 4.975 GHz, the realized gain may be about 5.98 dBi with radiation efficiency of about 70.1 %. Radiation efficiency can be the ratio of radiated power to input power, which may not account for reflection loss due to mismatch. The highest radiation efficiency within the matching band can be about 73.8% realized at about 5.165 GHz, where the realized gain of about 6.2 dBi is slightly lower than the maximum due to reflection loss.

[0065] As a comparison, a patch antenna, which has exactly the same configuration as the proposed patch antenna design but has no vias, is fabricated and measured. It matches at about 5.03 GHz with about 1 .4% matching bandwidth. The measured realized gain is about 6.85 dBi, and radiation efficiency is about 78.2% at about 5.03 GHz. In comparison to the regular-patch antenna, the proposed antenna may realize the beam steering within a much wider matching bandwidth and maintain similar radiation efficiency at the same time. [0066] Accordingly, many embodiments provide beam scanning in a single-patch antenna. The antenna design in accordance with many embodiments can be implemented by incorporating vias to create two coupled half-mode cavities in a rectangular patch antenna. The phase difference may be manipulated between the two equivalent magnetic currents and thus steer the beam through this design. Circuit model can be built in order to control the coupling and achieve the wide matching bandwidth. Measured and simulated results show that the beam may scan over a large range as the frequency changes while maintaining high radiation efficiency. The beam-scanning antenna in accordance with many embodiments has advantages of compact size and low cost. It can be scaled to high-frequency domain due to the simplicity. The phase manipulation within a single patch can potentially benefit many patch array designs.

[0067] Although specific methods and systems for generating beam scanning patch antennas are discussed above, many different designs can be implemented in accordance with many different embodiments of the invention. It is therefore to be understood that the present invention may be practiced in ways other than specifically described, without departing from the scope and spirit of the present invention. Thus, embodiments of the present invention should be considered in all respects as illustrative and not restrictive. Accordingly, the scope of the invention should be determined not by the embodiments illustrated, but by the appended claims and their equivalents.

Claims

What is claimed is:

1. A rectangular patch antenna, comprising:

an inset feed;

a rectangular flat metal patch;

a flat metal ground plane;

a substrate between the flat metal patch and the flat metal ground plane;

a plurality of metal vias connecting the flat metal patch to the flat metal ground plane, wherein the plurality of metal vias are arranged in a line that divides the patch antenna into two half-mode cavities and are positioned slightly off center so that the two half-mode cavities are not identical in size; and

wherein each half-mode cavity acts as a resonator.

2. The rectangular patch antenna of claim 1 , further comprising a circuit, wherein the circuit is configured to provide a source signal; wherein the patch antenna emits a certain pattern of radiation based upon the source signal.

3. The rectangular patch antenna of claim 2, wherein the emitted radiation is beam scanning.

4. The rectangular patch antenna of claim 3, wherein the beam scanning is a function of frequency.

5. The rectangular patch antenna of claim 1 , wherein the substrate comprises of a dielectric material.

6. The rectangular patch antenna of claim 1 , wherein the patch comprises of a conductive material.

7. The rectangular patch antenna of claim 1 , wherein the ground plane comprises of a conductive material.

8. The rectangular patch antenna of claim 5, wherein the dielectric material has a dielectric constant of about 2.2.

9. The rectangular patch antenna of claim 5, wherein the dielectric material has a loss tangent of about 0.001 .

10. The rectangular patch antenna of claim 1 , further comprising a gap, wherein the vias are placed in a line from one side to the other side of the antenna and the gap is an enlarged spacing omitting vias in the center of the line.

1 1 . The rectangular patch antenna of claim 10, wherein the gap is about 3.5 millimeter.

12. The rectangular patch antenna of claim 1 , wherein the patch antenna has a width of about 28 millimeter.

13. The rectangular patch antenna of claim 1 , wherein the patch antenna has a length of about 19.5 millimeter.

14. The rectangular patch antenna of claim 1 , wherein the patch antenna has a height of about 0.79 millimeter.

15. The rectangular patch antenna of claim 1 , wherein the metal vias have a spacing of about 1.6 millimeter.

16. The rectangular patch antenna of claim 1 , wherein the metal vias have a diameter of about 0.8 millimeter.

17. The rectangular patch antenna in claim 1 , wherein the antenna works in coupled modes.

18. The rectangular patch antenna in claim 17, wherein the coupling between the two half-mode cavities adjusts the phase difference as a function of the operating frequency.

19. The rectangular patch antenna as in claim 10 or 17, wherein the coupling takes place at the gap.

20. The rectangular patch antenna of claim 17, wherein the coupling is either even mode or odd mode.