RU2526741C1 - Radar antenna with reduced scattering cross-section - Google Patents

Abstract

FIELD: radio engineering, communication.

SUBSTANCE: radar antenna, having at least one radiator operating in a given frequency band, placed in front of radiators in one plane of the a frequency selection device with band characteristics which enable to transmit electromagnetic radiation in the operating frequency band, and reflect radiation outside said band, between the radiators at a distance therefrom in one plane, there are linear arrays of identical dipoles which form a flat two-dimensional periodic mesh structure.

EFFECT: reducing the scattering cross-section of the antenna in the operating frequency band thereof.

13 dwg

Description

The invention relates to radio engineering, namely to antenna technology, and can be used in the design of antenna devices with a reduced effective scattering area (EPR).

One of the main structural elements of modern aircraft, making a significant, up to 30% or more, contribution to their EPR in the sectors of the front hemisphere, are antennas of avionics (avionics). Of all the avionics, the largest contribution to the EPR of the aircraft is made by the bow antenna compartment with the antenna of the airborne radar station. Various measures are taken to reduce the visibility of avionics avionics, including the replacement of parabolic reflector antennas with active phased array antennas (AFAR) [Foreign Military Review. No. 11 (680), Moscow, 2003]. Due to this, the problem of reducing the levels of reflections from equipment elements located behind the opening of the antenna is solved. In addition, receiving and emitting AFAR modules can be installed on a low-reflective base (plane), where, unlike waveguide-slotted PARs, their EPR levels are mainly determined by reflection from the radiating elements of the modules. However, at present, the task of creating subtle antennas cannot be considered completely solved; therefore, original technical solutions, which make it possible to approach its solution, are of particular value.

The closest technical solution to the proposed one is an antenna with a reduced backscatter surface 1 (Fig. 1) [DE 3642072. MKI: G01S 7/38, H01Q 15/14, 1988, No. 25] containing at least one emitter 2 operating in a given operating frequency band, and frequency selection devices 3 with band characteristics located in front of the emitter in the same plane, which allow transmission of electromagnetic radiation in the working frequency band, and reflect radiation outside this band. Obviously, the main drawback of such an antenna is its “visibility” in the operating frequency band, when the antenna reflects back part of the energy coming from an external radiation source.

The objective of the present invention is to reduce the effective scattering area of the antenna in the band of its operating frequencies.

The technical result that provides a solution to this problem is an antenna with reduced radar signature in the band of its operating frequencies.

This problem is solved due to the fact that in a radar antenna with a reduced effective scattering area 1 containing at least one emitter 2 operating in a given operating frequency band and placed in front of the emitter in the same plane of a frequency selection device 3 with strip characteristics, allowing electromagnetic radiation to be transmitted in the operating frequency band, and outside this band to reflect radiation, according to the invention, between emitters, commensurate with the working wavelength λ ₀ , at a distance h from them in one plane, linear gratings 4 of the same dipoles of length l and cross-sectional diameter d are placed, forming a flat two-dimensional-periodic mesh structure with crosshairs that coincide with the middle of the gaps between the dipoles of magnitude b (figure 2), while the numerical values of h, l, d and b are selected from the relations

h≈λ _0/4, l≈0,5λ _0, d≈0,05λ _0, b≥0,1λ _0, where

λ ₀ - average working wavelength of the emitters of the antenna,

l is the length of the dipole,

d is the diameter of the cross section of the dipole,

b is the gap between the dipoles.

Let us explain this technical solution. From the theory and technology of antennas [Modern theory and practical use of antennas, ed. V.A. Neganova. M .: "Radio engineering". 2009. P.611] there are various principles of construction of combined HEADLIGHTS of the microwave range. One of the circuit solutions of combined lattices provides for their placement one above the other at a certain distance. Moreover, the upper lattice, for example, is a vibrator, and the emitters of the lower lattice can be waveguides, slots, vibrators and other elements whose transverse dimensions, depending on the type of emitter, range from 0.4λ ₀ to 1.2λ ₀ , t. e. commensurate with the working wavelength. Taking into account the directed properties of the emitters, the limiting distances between them in the grating are established. In this case, they are guided by the fact that the common-mode addition of the fields of individual emitters occurs within the main diffraction maximum (radiation pattern), and the remaining diffraction maxima of the higher orders are absent. To fulfill this condition, for example, the distance between the emitters should be no more than 0.58λ ₀ ... 0.68λ ₀ .

The combination of the gratings can be performed within one radiating aperture, while the emitters of one range are located between the emitters of another range. Such a combination is conveniently carried out for slotted, waveguide and vibrator emitters. Combined on a single radiating grating, the AFAR has broadband properties for operation both at a wavelength of λ ₀₁ and at a wavelength of λ ₀₂ (frequencies ω ₀₁ and ω ₀₂ ) (Fig. 3). However, the combination of two lattices one above the other or one in the other can lead to some deterioration in their electrodynamic characteristics. The results of studies of these antennas show that the combination leads to a decrease in gain, an increase in the level of side lobes and the restriction of the scanning sector. With a frequency ratio of ω ₀₂ / ω ₀₁ ≈2 ... 4, the characteristics of the high-frequency grating may deteriorate. For this reason, in order not to violate the functional properties of the combined AFAR, it is advisable to adhere to the ratio when λ ₀₂ ≈0.5λ ₀₁ .

Consider the combined AFAR as a radar reflector. The first grating can be represented as a conductive surface (opening the antenna), composed of flat emitters commensurate with the wavelength, between which horizontal symmetrical vibrators are uniformly located at a given height. The field reflected from such a design will be the result of interference from fields reflected directly from the aperture of the antenna and from vibrators located above the conducting plane. However, it is known that for anti-radar masking of metal objects used are radio-absorbing coatings of the interference type [Vakin SA, Shustov L.N. Fundamentals of radio counteraction and electronic intelligence M .: Sov. radio. 1968. With 347], the principle of which is based on the fact that the incident and reflected waves cancel each other out. Interference-type resonant radar absorbing coatings that are more effective in terms of weight and overall characteristics are also known. The simplest representative of the coating thickness is a two-layer structure h≈λ _0/4, consisting of dielectric and resistive film [Giants VD and others. Radio engineering systems in rocket technology. - M .: Military Publishing House, 1974. P.230] with an input resistance of the normally reflecting surface of 377 Ohms. It is known that to expand the frequency range and ensure the effectiveness of the interference coating with an increase in the angle of incidence of electromagnetic energy, the resistive film is replaced by a lattice of identical dipoles oriented parallel to the electric field vector. The equivalent circuit in this case is a constant resistance in the form of a series circuit R, L, C (Fig. 4), the loss resistance of which is determined by the ohmic resistance of the dipoles and the grid pitch, and the closed line is determined by the parallel resonant circuit R ', L', C ' . In this case, the input conductivity of such a line is determined from the relation

, где $$\frac{\omega }{{\omega }_o}=\frac{{\lambda }_o}{\lambda }$$

, $$y=-j\frac{\sqrt{\epsilon }}{z_o}c\mathrm{<figure-callout\; id="2"\; label="t\; g\; \pi "\; filenames="00000008.png,00000009.png"\; state="\{\{state\}\}">t\; </figure-callout>}g\frac{\pi }{}\frac{}{2}\frac{\omega }{{\omega }_o}$$

where $$\frac{\omega }{{\omega }_o}=\frac{{\lambda }_o}{\lambda }$$

,

where λ is the wavelength in a dielectric with a permittivity of 8.

The conductivity of the parallel resonant circuit and the reflection coefficient of the dual-circuit circuit are respectively written as

, $$y=j\left(\frac{\omega }{{\omega }_o}-\frac{{\omega }_o}{\omega }\right)\sqrt{\frac{C\text{'}\text{'}}{L\text{'}\text{'}}}$$

,

. $$r=\frac{\left[-R-\frac{\mathrm{one}}{\sqrt{\epsilon }}\sqrt{\frac{L}{C}}\left(\frac{\omega }{{\omega }_o}-\frac{{\omega }_o}{\omega }\right)\mathrm{<figure-callout\; id="2"\; label="t\; g\; \pi "\; filenames="00000008.png,00000009.png"\; state="\{\{state\}\}">t\; </figure-callout>}g\frac{\pi }{}\frac{}{2}\frac{\omega }{{\omega }_o}\right]+j\left[\frac{\mathrm{one}}{\sqrt{\epsilon }}\mathrm{<figure-callout\; id="2"\; label="t\; g\; \pi "\; filenames="00000008.png,00000009.png"\; state="\{\{state\}\}">t\; </figure-callout>}g\frac{\pi }{}\frac{}{2}\frac{\omega }{{\omega }_o}\left(R-z_o\right)-\sqrt{\frac{L}{C}}\left(\frac{\omega }{{\omega }_o}-\frac{{\omega }_o}{\omega }\right)\right]}{\left[R-\frac{\mathrm{one}}{\sqrt{\epsilon }}\sqrt{\frac{L}{C}}\left(\frac{\omega }{{\omega }_o}-\frac{{\omega }_o}{\omega }\right)\mathrm{<figure-callout\; id="2"\; label="t\; g\; \pi "\; filenames="00000008.png,00000009.png"\; state="\{\{state\}\}">t\; </figure-callout>}g\frac{\pi }{}\frac{}{2}\frac{\omega }{{\omega }_o}\right]+j\left[\frac{\mathrm{one}}{\sqrt{\epsilon }}\mathrm{<figure-callout\; id="2"\; label="t\; g\; \pi "\; filenames="00000008.png,00000009.png"\; state="\{\{state\}\}">t\; </figure-callout>}g\frac{\pi }{}\frac{}{2}\frac{\omega }{{\omega }_o}\left(R+z_o\right)-\sqrt{\frac{L}{C}}\left(\frac{\omega }{{\omega }_o}-\frac{{\omega }_o}{\omega }\right)\right]}$$

.

при условии, что $$\sqrt{\frac{L}{C}}=\frac{\pi }{4}{z}_0\sqrt{\epsilon }$$

.Figure 5 shows the frequency dependence of the reflection coefficient of a radar absorbing coating based on a one-dimensional dipole lattice. The optimal matching of the double-circuit absorber at R = R '= Z ₀ and when compensating the imaginary components of the impedance of both loops is represented by the relation $$\sqrt{\frac{L}{C}}={\mathrm{<figure-callout\; id="0"\; label="z"\; filenames="00000011.png,00000012.png"\; state="\{\{state\}\}">z</figure-callout>}}_0^2\sqrt{\frac{C\text{'}\text{'}}{L\text{'}\text{'}}}$$

provided that $$\sqrt{\frac{L}{C}}=\frac{\pi }{\mathrm{four}}{\mathrm{<figure-callout\; id="0"\; label="z"\; filenames="00000011.png,00000012.png"\; state="\{\{state\}\}">z</figure-callout>}}_0\sqrt{\epsilon }$$

.

For the practical use of this coating, it is important that its effectiveness does not depend very much on the length of the dipoles and the distance between them. Experimental results cited in V. Velikanov and others. Radio engineering systems in rocket technology. - M .: Military Publishing House, 1974. P.234-236, show that when the dipole length l changes by 10% and the distance between the linear gratings by 15%, the reflected energy increases by less than 10%. For effective absorption, dipoles must be oriented parallel to the electric field vector. In this case, the field reflection coefficient will be minimal. Figure 6 shows the dependence of the reflection coefficient of a one-dimensional dipole lattice on the angle of incidence under the condition of the optimal distribution density of dipoles and the polarization of the electric field coinciding with them. The graph shows that for a well-matched coating with a resonant dipole length l≈0.5λ _{0, the} reflected energy is less than 2% in power in the range of incidence angles from 0 to 40 °. The angular dependence of the reflection coefficient of a one-dimensional lattice of dipoles for a wave with different polarization is shown in Fig.7. So that the effective absorption of electromagnetic energy does not depend on the random orientation of the electric field vector, it is necessary to add a second linear dipole lattice to the first linear dipole lattice in the coating scheme by rotating it 90 ° relative to the first one. From the dependence shown in Fig. 8, it can be seen that the inclusion of additional dipoles deployed at 90 ° in the coating scheme does not impair the consistency of the coating. In other words, the reflection coefficient of the coating based on a cross-shaped dipole lattice is practically independent of the polarization of the irradiating signal. From the analysis of the presented dependencies [Velikanov V.D. and others. Radio engineering systems in rocket technology. - M .: Military Publishing House, 1974. P.237] it follows that the diameter of the dipoles and the gap between them in the lattice must be selected on the basis of the condition that provides the minimum reflection coefficient of the lattice, ie when d≈0.05λ ₀ , b≥0.1λ ₀ .

Thus, the choice of the parameters h, l, d, and b with respect to the wavelength allows one to minimize the reflection coefficient of the network of dipoles above the conducting plane and to ensure the absorption of the incident field without deterioration of the tactical and technical characteristics of the AFAR in the band of its operating frequencies.

To eliminate reflected from the grating with square cell signal diffraction peaks (when the distance between the dipole gratings longer λ _0/2), a second linear dipole grating can be rotated through an angle less than 90 °, forming thus a network structure with oblique cell or by constructing a mesh structure with a triangular cell. Possible options for such designs are presented in Fig.9 (a, b).

In practice, linear dipole arrays can be mounted at a given distance from the AFAR emitters using a plane-parallel dielectric plate, which does not contradict the conditions when it is necessary to ensure wide-angle matching of antenna emitters [Antennas and microwave devices (design of phased antenna arrays), ed. D.I. Voskresensky. M .: "Radio and communication". 1981. P.216]. Of course, not to the detriment of the AFAR performance, the density of the dipole distribution over the opening will be determined, first of all, by the distance between the emitters and will not always be optimal for effective absorption even in the AFAR operating frequency band.

A radar antenna with a reduced effective scattering area works as follows. A flat front of an electromagnetic wave falls on the antenna opening. Frequency selection devices with predetermined band-pass characteristics transmit electromagnetic radiation in the operating frequency band of the antenna, and outside this band they reflect radiation in different directions, excluding reverse re-reflections to the side of the radiation source. An electromagnetic wave in the working frequency band of the antenna, having passed through the selection device, is partially reflected from linear gratings from the same dipoles of resonant length and is added in antiphase with the wave passing through the gratings, the air gap and reflected from the emitters. Another part of the electromagnetic wave that has passed through the linear gratings is repeatedly reflected from the interface between “emitters - air” and “gratings - air”, loses energy when absorbed by dipoles, and eventually attenuates in the air gap. Thus, due to the addition of waves in antiphase and the loss of its energy, the reverse re-reflection of the electromagnetic wave from the antenna in the direction normal to its opening is excluded.

The essence of the proposed technical solution is illustrated by figures 1-13, which shows a radar antenna with a reduced effective scattering area, as well as the results of experimental studies of its model in the conditions of the Reference Radar Measurement Complex of the SIC (Tver) 4 CRI of the Ministry of Defense of Russia ["Reference Radar Measurement Complex Open Type (ERIC). " Weapons and technology of Russia. Encyclopedia. XXI Century. Air defense and missile defense. Volume IX. M .: "Weapons and technology." 2004. S. 385].

Figure 1 shows a diagram of a known radar antenna with a reduced effective scattering area.

Figure 2 - diagram of the proposed radar antenna with a reduced effective scattering area.

Figure 3 - diagram of a dual-frequency combined AFAR.

Figure 4 is an equivalent circuit of a resonant radar absorbing coating based on a one-dimensional dipole lattice.

Figure 5 - frequency dependence of the reflection coefficient of a radar absorbing coating based on a one-dimensional dipole lattice.

Figure 6 - dependence of the reflection coefficient of a one-dimensional dipole lattice on the angle of incidence (0 - corresponds to the fall along the normal to the lattice).

7 is an angular dependence of the reflection coefficient of a one-dimensional dipole lattice for a wave with different polarization.

On Fig - angular dependence of the reflection coefficient of the lattice of linear and cross-shaped dipoles.

Figure 9 - options mesh structure with an oblique (a) and triangular (b) cell.

Figure 10 - geometry of the model elements of the proposed radar antenna with a reduced effective scattering area, where for λ ₀ = 3.2 cm; l≈15 mm; d≈1.5 mm; b≈5 mm; h≈8 mm.

11 is a photograph of a model of the proposed radar antenna with a reduced effective scattering area.

On Fig - scheme of the experiment, where the known antenna (a) is a metal disk with a diameter of 12.5λ _0, simulating 212 emitters (2) lying in the same plane, and the proposed antenna (b) is the same disk, but with located in front of it is a two-dimensional periodic mesh structure of linear dipole lattices (4).

In Fig.13 - the results of experimental studies: backward reflection diagrams (a) of the model of the known (k) and proposed (l and m) antennas at a wavelength of λ = 3.2 cm with vertical polarization of radio emission, as well as the corresponding laws of distribution of EPR values (b) in the sector of observation angles 0 ± 5 °. Option (l), when the electric field vector is oriented along half the number of all dipoles. Option (m), when the vector is oriented at an angle of 45 ° to all dipoles.

An analysis of the results allows us to conclude that the proposed radar antenna with a reduced effective scattering area compared to the known prototype antenna has lower EPR values (in probability level 0.5) in the location sector 0 ± 5 ° relative to the normal to the antenna opening by 5 , 6 and 3.5 dB (for the most and less effective options for the position of the polarization vector of the radio emission, respectively).

Implementation of the inventive antenna with a reduced effective scattering area is not difficult. Obviously, the invention is not limited to the foregoing example of its implementation. Based on its scheme, other options for its implementation may be provided, without going beyond the scope of the invention.

The device is advisable to use in organizations involved in the design of antenna radar systems.

Claims (1)

A radar antenna with a reduced effective scattering area, containing at least one emitter operating in a given operating frequency band, placed in front of the emitter in the same plane of a frequency selection device with band characteristics that allow transmission of electromagnetic radiation in the operating frequency band, and reflecting radiation outside this band, characterized in that between the emitters, commensurate with the working wavelength λ ₀ , at a distance h from them in the same plane are placed linear gratings of about identical dipoles of length l and cross-sectional diameter d, forming a flat two-dimensional-periodic mesh structure with crosshairs that coincide with the middle of the gaps between dipoles of magnitude b, while the numerical values of h, l, d and b are selected from the relations
h≈λ _0/4, l≈0,5λ _0, d≈0,05λ _0, b≥0,1λ _0, where
λ ₀ - average working wavelength of the emitters of the antenna,
l is the length of the dipole,
d is the diameter of the cross section of the dipole,
b is the gap between the dipoles in the lattice.

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