RU2504902C9 - Receiving radio centre - Google Patents

Abstract

FIELD: radio engineering, communication.

SUBSTANCE: receiving radio centre further includes an antenna system consisting of n directional antennae which correspond to n multichannel radio receivers, n bidirectional fibre-optic communication lines, n signal processing units, a local area network (LAN), a radio reception channel control centre, wherein each multichannel radio receiver has an input device, a first multiplexer/demultiplexer, a first optoelectronic/electro-optical converter, a first optical transceiver, and each of the m analogue channels has a first tunable filter unit (1TFU), a second control and monitoring unit (2CMU), a controlled radio frequency amplifier, a second tunable filter unit (2TFU), a first controlled attenuator (1CA), a first controlled switch (1CS), a frequency converter, a controlled intermediate frequency amplifier, a second intermediate frequency filter unit (2IFFU), a second controlled switch (2CS), a second controlled attenuator (2CA) and an analogue-to-digital converter unit (ADC).

EFFECT: high noise immunity of receiving messages due to high sensitivity, intermodulation dynamic range and reliability.

5 cl, 2 dwg

Description

The present invention relates to the field of radio engineering and can be used in radio communication systems and radio monitoring in order to increase the noise immunity of receiving messages.

A well-known radio reception center (see Chelyshev V.D. Reception radio centers. - M .: Communication, 1975. - 264 p.).

This receiving radio center (PRC) includes serially connected: antenna-feeder system (AFS), switching and distribution channel (SRT), broadband radio receiving system (SRPS), a complex of signal processing and transmitting them to the final destination. At the same time: the AFS contains antenna-feeder devices (AFU) containing antennas and feeder lines, antennas with their radiation patterns (DN) overlap all the necessary directions in volume space and all the necessary polarizations of the received radio signals; SRT contains broadband amplifiers, splitters, radio signal switches and provides multiple use of each AFU; SRPS contains many separate radio receivers (RPUs) that receive radio signals in the entire operating frequency range.

PRC allows receiving radio signals at different and changing frequencies from different and changing directions.

However, due to the long distance of each RPM from the antennas, due to the attenuation of the radio signals in the feeder lines, the PRC has a low sensitivity for receiving radio signals, and with an air feeder line, the antenna effect leads to distortion of the radiation pattern and a decrease in the directional properties of the antennas. In addition, due to the presence of amplifiers, splitters and switches in the SRT, the PRC has a low dynamic range (DD).

Well-known PRC (see. Poberezhsky ES Digital short-wave radio receivers. - journal "Radio Engineering", 1978, No. 5, p.16-24), which contains a series-connected AFU containing an antenna and a feeder line, a broadband multi-channel RPU (SHMPU ), a device for demodulating and decoding signals (UDDS), wherein the SMRPU contains serially connected matching device (SU), the input of which is connected to the feeder line of the AFU, and the outputs of the SU are connected to the inputs of bandpass filters (PF), and the passband (PP) of the PF is close fit each other and in total cover the entire operating frequency range, an analog-to-digital converter (ADC) is connected to the output of each PF, the ADC outputs are connected to the inputs of a digital computer (CV), which is a matrix switch (MK) and a set of digital RPUs (CRPU) the outputs of which, when combined, are connected to the input of the UDDS, which is implemented on the basis of an electronic computer (computer).

In this PRC, higher DC, since there is no SRT. However, the well-known PRC has low noise immunity due to the wide PP of input PFs in each SMRPU, which are subjected to multi-signal external influences, as well as low sensitivity due to attenuation of radio signals in a long AFU feeder line. The low sensitivity is due to the fact that to prevent the influence of electric digital signal streams acting in the output lines of the SMRPU on the antenna, the SMRPU is removed from the antenna by increasing the length of the feeder line, and when using air feeder lines, the inevitable antenna effect leads to distortion of the beams and a decrease in directional properties antennas. In addition, the PRC has low reliability, since there is no possibility of interchangeability of PF in each SHMPU. This is due to the fact that each non-tunable PF operates in only one, rigidly fixed, frequency range of the working frequency range. The extension of the DD in this PRC is limited by the fact that one analog-to-digital converter is used in each analog receive path, adaptation to the external influence of radio signals is carried out only by means of automatic gain control (AGC), and there is no possibility of changing the structure and width of the receiver of the receive path to the ADC.

Well-known PRC (see Poberezhsky ES Digital radio receivers. - M .: Radio and communications, 1987. - 184 p.), Which contains a series-connected AFS, analog channels (AK), the outputs of which are connected to the inputs of a CV containing series-connected MK and a set of central control circuits, the outputs of which, when combined, are connected to the input of the UDDS, while the AFS contains AFUs, each of which is an antenna and a feeder line, in AK the signal amplification can be direct or with additional frequency conversions. Frequency PP AK closely adjacent to each other and in total overlap the operating frequency range, and UDS is implemented on a computer.

This PRC allows diversity reception, which increases the noise immunity of the reception. However, the PRC has low noise immunity due to wide AC AK and low sensitivity due to attenuation of radio signals in the long feeder lines between the antennas and antenna inputs of the AK, and when using air feeder lines, the arising antenna effect leads to distortion of the beam pattern and a decrease in the directional properties of the antennas. In addition, the PRC has low reliability due to the lack of interchangeability of the AK, since each AK works only in one, rigidly fixed, frequency range of the operating frequency range. The inability of the AK to tune in according to the frequency of the operating range and change the width of the PP does not allow the PRC to adapt to the changing electromagnetic environment (EMO) at the receiving site. The use of one ADC in each AK does not allow to increase the DD for intermodulation of the PRC. There is no possibility of structural and parametric optimization of the PRC by adapting the structure and operation modes of the elements of each analog channel to the EMO at the receiving location.

Known PRC (see RF patent for the invention No. 2308149, M.cl. Н04В 1/06, publ. 10.10.2007), which contains a series-connected AFU, multichannel radio receiver (MPPU) and UDS, while the AFU contains sequentially the antenna and the feeder line are connected, the control system includes an input, the input of which, being the input of the control device, is connected to the feeder line of the AFU, and the outputs of the control system are connected to the inputs of the analog channels (AK), the outputs of the AK are connected to the inputs of a digital computer (CV), the outputs of which are MPPU outputs, while each AK contains sequentially soy One single-pass filter (PF), the input of which is connected to one of the control system outputs, a controlled attenuator (UA), a first frequency converter (1PrCh), containing a first mixer (CM1), a signal conditioning unit of the first local oscillator (BFSG1) and a filter of the first intermediate frequency (FPP1), the width of the PP of which is equal to the width of the PP PF, a second frequency converter (2PrCh), containing a second mixer (SM2), a second local oscillator signal generating unit (BPSG2) and a filter of the second intermediate frequency (FPP2). FPP2 is a frequency-selective system (NIS), which contains a set of filters, the inputs of which are connected in parallel, and the outputs are connected to the inputs of a ring switch (CC), the output of which is connected to the input of the ADC, the output of which is the output of the AC and connected to the input CV, containing a series-connected matrix switch (MK) and a block of digital radio receivers (BCRPU), which is a set of digital radio receivers, with the input of the CV being the input of the MK, and the output of the CV being the output d BCRPU, which is connected to the input of the device demodulation and decoding of signals (UDDS), implemented on a computer.

In this case, the number of PFs is equal to the number of AKs in the MCI and equal to the number of frequency intervals into which the entire working frequency range is divided, so that the PP PF are closely adjacent to each other and in total overlap it, and the number of PF depends on the size of the working frequency range and the width of the PP of each PF; the control input of the UA is connected to the first output of the ADC control to ensure the AGC of the AK; BFSG1 of each AK is a crystal oscillator tuned to a single frequency; the BFSG2 signal is formed by converting the BFSG1 signal with a reference frequency signal, which provides compensation for the frequency instability of the BFSG1 signal; the set of Numerical filters with their PPs break down the PPP1 filters into narrower PPs, the number of filters in the NPS is equal to the number of frequency intervals into which the PPP1 filters are divided, and the PPs of the NPC filters are closely adjacent to each other and in total overlap the entire PPP1 filter, therefore, each ChIS filter has a central tuning frequency different from the frequency converter; the control input of the QC is connected to the second control output of the ADC to control the QC signals of the ADC's readiness to process the next signal report, thereby the QC, by the ADC commands, alternately connects the output of each NUM filter to the ADC input, that is, during the sampling period the signals located in the PP filters Numbers are alternately subjected to analog-to-digital conversion and thus provides a temporary seal of the signals from the outputs of the NUM filters to the input of the ADC, thereby ensuring the width of the AC AK to the input of the ADC, the apparent width of the PP of each filter of the Numbers, due to this, the power of the group signal at the input of the ADC is reduced; and the BCRPU contains digital channels (CC) of reception with the width of the PP necessary for receiving signals.

In this PRC, a higher signal DD was achieved by applying temporary multiplexing of the group signal, thereby reducing the power of the signal acting on the ADC input. However, the PRC has a low DD for intermodulation, since the power of the individual signals located in the PC of each filter of the Numbers and affecting the ADC does not decrease and the intermodulation interference that occurs in the ADC and arriving at the input of the CV is as large as without the use of time compression in addition, the PRC has low sensitivity due to the large attenuation of radio signals in long feeder lines, and the use of air feeder lines, in addition to signal attenuation, additionally leads to distortion of the pattern and a decrease in the directional properties of the antennas n It is impossible to place an MRPU in the immediate vicinity of the antenna due to the influence of electric digital signals of the input / output lines of the MPPU on it, therefore, to solve the problem of electromagnetic compatibility (EMC), MPPUs are placed at a large distance from the antenna. In addition, the PRC has low reliability, since there is no protection of the input circuits of each MPPA from the effects of high power signals and there is no possibility of interchangeability of the AK in each MPPU, since the AK can work only in one rigidly fixed frequency range of the operating frequency range. Low reliability, low DD on intermodulation, and low sensitivity determine the low noise immunity of receiving PRC messages. This PRC is selected as a prototype. The technical result of the invention is to increase the noise immunity of receiving messages by increasing sensitivity, dynamic range for intermodulation and reliability.

The achievement of the technical result is provided in the receiving radio center (PRC), containing at least one antenna, one multi-channel radio receiving device (MRPU), one matrix switchboard MK, one block of digital radio receiving devices (BTsRPU), one demodulation and decoding device (UDS) Moreover, the control device (CS), analogue channels (AK) are included in the MRPU, and the SS outputs are connected to the inputs of the corresponding AK, each of which contains at least one band-pass filter (PF), one controlled attenuator (UA) , a frequency converter (PF), one analog-to-digital converter (ADC), the PF input is an AK input for connecting to the corresponding output of the control unit, the PF contains a mixer (SM), a local oscillator signal generating unit (BPSH), at least one intermediate filter frequency (PLL), with the RF input being the first SM input, the second input of which is connected to the BPSG output, the SM output is connected to the PLL input, the output of which is the RF output, and the ADC output is the AC output, and the corresponding MK outputs are connected to the corresponding inputs BTsRPU, Vkho the bottom / output bus of which is connected to the corresponding output / input bus of the UDDS, characterized in that it contains an antenna system (AC) of n directional antennas, the outputs of which are directly connected to the inputs of the corresponding n MPPUs, equipped with optical inputs / outputs, which, through the input n bidirectional fiber-optic communication lines (FOCL) are connected to n optical outputs / inputs of n input signal processing units (BOC), other electrical inputs / outputs of which are connected to n outputs / inputs of input loc a computer network (LAN), the other input / output of which is connected to the output / input of the entered radio channel control center (CCAM), wherein n is selected from the condition:

, $$\frac{\mathrm{four}\pi }{\Delta \varphi }\le n\le p\frac{\mathrm{four}{\pi }^2}{\Delta \varphi \Delta \theta }$$

,

where Δφ is the width of the radiation pattern (MD) of the antennas in the azimuthal plane in radians (rad),

Δθ is the antenna beam width in the elevation plane in rad,

p is a coefficient taking into account the number of polarizations of radio waves received at the corresponding antennas,

wherein each MRPU contains an input device (VU), the input of which is the input of the MPU, am the outputs of the VU are m outputs of the said VU included in the VU, made controlled (USU), the input connected to the output of the controlled amplification unit (UBU) introduced into the VU, input which is connected to the output of the managed protection unit (UBZ) introduced into the VU, the input of which is the VU input, and their inputs / outputs of control and monitoring of the UBU, UBZ and USU are connected with the corresponding outputs / inputs of the first control and monitoring unit introduced into the VU For (1BUK), the signal outputs and the input / output control and monitoring buses of all m AK are connected to the corresponding m inputs and m output / input buses of the first multiplexer / demultiplexer introduced into the MPPU, another output / input bus connected to the input / output bus of the control unit, which is the corresponding input / output bus 1BUK, and the input / output of the first multiplexer / demultiplexer is connected to the output / input of the first optoelectronic / electron-optical converter introduced into the MPPU, the other optical output / input of which th is connected to input / output first entered the optical transceiver, the other input / output of which is input / output for connection MRPU biofeedback, wherein m is selected from the condition:

, $$\frac{\Delta f}{\Delta F_{\max }}\le m\le \mathrm{TO}s\frac{\Delta f}{\Delta F_{\min }}$$

,

where Δf is the range of working radio frequencies,

ΔF _min - the minimum width of the PP AK,

ΔF _max - the maximum width of the PP AK,

Kz is the load factor of the bottom of the antenna of this antenna, to which this MRPA is connected, by radio signals from sources of radio emission (IRI), while all m AKs of all n MRPUs are performed identically and each AK contains serially connected the first block of tunable bandpass filters (1BTPF), a controlled amplifier introduced radio frequencies (URCH), the introduced second tunable bandpass filter unit (2BPPF), the mentioned first controlled attenuator (1УА), the introduced first managed switch (1УК), the mentioned frequency converter, the introduced controlled amplifier П H (UPUCH), the second IF filter unit (2BFPCH), the second managed switch (2UK), the second managed attenuator (2UA) and the analog-to-digital conversion unit (BACP), the output of which is the signal output of the AK, and the input / output control bus and AC control is the corresponding input / output bus of the introduced second control and monitoring unit (2BUK), the other corresponding control and monitoring inputs and outputs of which are connected to the corresponding control / monitoring outputs / inputs of all units included in the AK, and the second input 2UK with it is one with the second 1UK output, while the BACP included in the AC contains a controlled signal splitter (URS), the input of which is the BACP input for connecting the 2UA output, an analog-to-digital converter unit (UACP), an analog and digital signal synthesizer (CACS) and a controlled digital signal adder (USSC), where k URS outputs are connected to the corresponding k inputs of the UARC, where k is equal to the number of ADCs included in the UARC, the other corresponding inputs of which are connected to the corresponding outputs of the UACS, and the corresponding k outputs of the UACS are connected to respective inputs USTSS k, the output of which is the output BATSP for connecting to first multiplexer / demultiplexer, wherein k is chosen from the condition:

$$\frac{P_{c\mathrm{BUT}\text{A.}\mathrm{TO}\mathrm{and}}{K}_{p\Sigma }}{R_{\mathrm{from}\mathrm{one}\mathrm{BUT}\mathrm{Ts}P\mathrm{and}}}\le k\le \{\begin{array}{c}{K}_{w\mathrm{one}\mathrm{BUT}\mathrm{Ts}PR}PR\mathrm{and}{r}_w=0r_c=\mathrm{one,}\\ \frac{{\delta }_{\left(W/\mathrm{FROM}\right)d\mathrm{about}P}{r}_c-\mathrm{one}}{r_w}PR\mathrm{and}{r}_w>0r_c<\mathrm{one,}\end{array}$$

where R _cAKi - the maximum allowable input power of one of the two signals acting on the input of the AC, in which the third-order intermodulation interference resulting from the interaction of these two signals in the AC before the input of the ADC has an acceptable value,

K _p∑ is the transmission coefficient for AC power to the input of a single ADC,

R _s1ATsPi - the maximum allowable power of one of the two signals acting on the input of a single ADC, in which the third-order intermodulation interference resulting from the interaction of these two signals within the integral ADC characteristic has an acceptable value,

To _w1ACPr - noise figure of a single ADC taking into account distortions and residuals of randomization noise,

δ _{(W / C) add} - allowable coefficient for increasing the noise / signal ratio at the output of the BACC compared to the ratio of the noise / signal at the output of a single ADC,

r _c is the cross-correlation coefficient between the signals in the BAC,

r _W - the coefficient of cross-correlation between the output noise UATSP,

and each biofeedback contains a second optical transceiver connected in series with its inputs / outputs, the input / output of which is the input / output of the biofeedback, a second optoelectronic / electron-optical converter and a second multiplexer / demultiplexer, corresponding input / output bus, m outputs and m input / output buses connected to the corresponding output / input bus, m inputs and m output / input buses of said MK, and its other corresponding input / output bus is connected to the corresponding output / input bus UDDS, the other corresponding input / output bus of which and the output bus of the mentioned BCRPU are connected to the corresponding output / input and input buses of the storage device (RAM) inserted into the biofeedback, the input / output of the UDDS is the input / output of the biofeedback L, by which the MK is connected to the L inputs of the BCRPU, is selected from the condition:

, $$L=m\frac{\Delta f_{\mathrm{BUT}\mathrm{TO}}}{\Delta F_{\mathrm{Ts}\mathrm{TO}}}$$

,

where L is the number of CRPU creating digital channels (CC) reception,

Δf _AK - the width of the PP of each AK,

ΔF _CC - the width of the PP of each Central Committee,

m is the number of AK.

In this case, the frequency converter may contain, as a local oscillator signal generating unit (BFSG), a controlled frequency synthesizer (UHF) with an output connected to the corresponding input of the above-mentioned SM, the output of which is connected to the input of the first IF filter block (1BFPC), the output of which is the RF output, and the input RF is the input of SM.

The controlled signal splitter (URS) included in the BACP may contain a block of controlled splitters of the received radio signal and a block of controlled analog phase shifters, while the input of the URS is the input of the block of controlled splitters of the received radio signal, k outputs of which are connected to the corresponding inputs of the block of controlled analog phase shifters, k outputs which are the outputs of the URS, while the input / output control and monitoring buses of the unit of controlled splitters of the received radio signal and the control unit The current analog phase shifters are connected to the corresponding output / input of 2БУК by means of control input and control URS.

The analog-to-digital converters (ADCS) unit included in the BACC may contain a signal splitter with a sampling frequency, k outputs of which are connected to the corresponding k inputs of a block of controlled phase shifters for a signal with a sampling frequency, each of k corresponding outputs of which is connected to each of k corresponding an input for a signal with a sampling frequency of each of k ADCs, the other inputs of which are connected to the outputs of the corresponding from k analog adders, one input of each of which is corresponding and k UACP inputs for connecting the outputs of URS, and the other input of each of k analog adders is each of k inputs of UACP for connecting to the corresponding of k outputs of analog signals of the CACS, and the output of each of k ADCs is connected to the input of each of k corresponding digital subtractors, the other the input of each of k digital subtractors, being each of the k inputs of the UACP, is connected to the corresponding of k outputs of digital signals of the CACS, and the k outputs of digital subtractors are k outputs of the UACP to connect to the k inputs of the controlled adder q phase signals (USTSS), while the input / output bus control and monitoring of the signal splitter with a sampling frequency and a block of controlled phase shifters for a signal with a sampling frequency through the input / output of control and control UATSP are connected to the corresponding output / input 2BUK.

The USSC, included in the BACS, may contain a block of controlled digital phase shifters, k inputs of which are k inputs of the USSC, and its k outputs are connected to the corresponding k inputs of the block of controlled digital adders, the output of which is the output of the USSC and BACP for connecting to the input of the first multiplexer / demultiplexer the input control and control buses of the block of controlled digital phase shifters and the block of controlled digital adders are connected to the corresponding input / output control and control of the USSC mu output / input 2BUK.

Introduction to the proposed PRC of an antenna system (AS) containing n highly efficient directional antennas, as well as the introduction of a LAN and CCAM, make it possible in the PRC to receive radio signals from radio emission sources (IRI) located anywhere in the surrounding space, and to selectively and spacedly receive radio signals over space, in polarization, in frequency, which makes it possible to increase the sensitivity of DD by intermodulation and, therefore, increase the noise immunity of the PRC.

To enable each speaker antenna to be connected directly to the antenna input of the corresponding MRPU without using a feeder line, optoelectronic / electron-optical converters, optical transceivers and FOCLs are introduced into each MRPU and BOS, which eliminates the attenuation of radio signals to the antenna input of the MPPU, eliminates the deterioration of the directional properties of the antennas, thereby increasing the sensitivity, spatial and polarization selectivity and thus increasing the noise immunity of the PRC.

Moreover, the possibility of connecting each speaker antenna directly to the antenna input of the corresponding MRPU led to the introduction into each MRPU of an input device (VU) containing a managed protection unit (UBZ), a controlled amplification unit (UBU), a controlled matching device (USU) and the first control unit and control (1BUK), which control, monitor, amplify and distribute the group signal from the antenna to the input of each analog channel (AK). The introduction of VU allows you to: protect the input of the MPPU from damage by high power signals and thereby increase reliability; increase the attenuation of the group signal and thereby increase the DD by intermodulation; to regulate the gain of the group signal and the number of connected AKs to the antenna, thereby increasing the sensitivity of the MRPU, that is, the introduction of VU can increase the noise immunity of the PRC.

Improving the reliability and, therefore, noise immunity of the PRC is also ensured by the use of the same type of interchangeable AK in all MRPUs, each of which, with the help of the introduced UCH, 2BUK, LAN, TsUKR, is tuned in frequency, which means that it is possible to receive signals by each AK at any frequency of the operating range, at In this case, interchangeability of AK is carried out both in automatic and in automated modes.

Introduced 2BUK and BACP, containing URS, UAATS, SACS, USTSS, allow to increase and optimize DD on intermodulation of each AK and thus increase the noise immunity of receiving PRC messages. Indeed, an URS containing a block of controlled splitters of the received radio signal and a block of controlled analog phase shifters branches the input signal into k branches with in-phase oscillations of the signals, and if necessary produces the required phase shift between the oscillations of the signals of certain branches that go to k inputs from the k outputs of the URS UACP, while the signal power at the input of each of the k ADCs is k times less than the signal power acting at the input of the URS, which allows you to increase the allowable signal power at the input de BACP and thus increase the DD on intermodulation of AK. After analog-to-digital conversion of signals, digital signals from k outputs of the UAACP are fed to the k inputs of the USSC, where the in-phase addition of the received digital signals is performed, and if necessary, the required phase shift between the oscillations of the signals of certain branches is performed before summing, while the oscillations of the correlated in-phase signals are summed arithmetically (by amplitudes), antiphase intermodulation interference is compensated (arithmetically subtracted), and uncorrelated oscillations of quantization and thermal noise the noise of each ADC are summed geometrically (effective values), so the signal-to-noise ratio at the output of the USSC does not change compared to a single ADC, and using a block of controlled phase shifters for a signal with a sampling frequency of the ADC, if necessary, a phase shift is made between signal oscillations with a sampling frequency for each ADC, which allows to increase the number of samples in the total signal at the output of the USSC and decorrelate the quantization noise of the ADC, thereby also preserving the signal / w ratio mind at the output of the USSC in comparison with a single ADC and, accordingly, maintain the sensitivity of the AK with a minimum increase in the gain in power, as well as increase the DD on intermodulation by compensating for noise and thus increase the noise immunity of the AK, MPPA and the whole PRC.

The introduced CACS allows you to synthesize an analog signal with an optimal shape and with an optimal sampling frequency, which is fed from the output of the CACC to the UAAC to the input of a signal splitter with a sampling frequency, where this signal branches out into k signals, which, through a block of controlled phase shifters, are input to the signal for sampling frequency each of k ADCs, optimizing the process of analog-to-digital conversion. In addition, in the CACS, k noise or wobble signals are synthesized, and each one separate signal is presented in digital and analog forms for randomization or wobble procedures in each ADC, and all, from 1 to k, noise or wobble signals random and independent among themselves. The introduction of noise or wobble signals allows one to reduce the level of interference arising within the differential characteristics of each ADC, to decorrelate between the output noise and interference of the ADC and thereby increase the sensitivity and DD on the intermodulation of each ADC and, accordingly, the entire ADC. The number of ADCs included in the UAAC from 1 to k is regulated using 2BUK by switching the inputs, outputs and power circuits in the URS, UATSP, SATSS, USTSS. The connection of the UAAC control input / output to 2BUK is also used for automatic adjustments in 1UA, 1BFPCh, 2BFPPCH, 2UA and, if necessary, in 1 BFPF, 2BPPF, UURCH, to optimize the AK parameters in order to increase the noise immunity of receiving messages.

The introduction of 2BUK, BACP, as well as 1BFPCh, 2BPFCH, 2UA 1BPPF, 2BPPF, UURCH allows adaptively, in automatic mode, to optimize the parameters (transmission coefficient, bandwidth) of the AK in order to increase the DC for intermodulation and, therefore, increase the noise immunity of the PRC.

The introduction of UBU, USU, 1BUK, LAN, CCA allows you to change the number of AK connected to the antenna, which is carried out in automatic mode, according to the control commands received from CCU or UDS, and in automatic mode according to the commands received from UBU to USU using 1BUK. Adjusting the number of AKs used to receive radio signals at a given time, allows optimizing the structure of the MCI according to the electromagnetic environment (EMO) at the receiving site and thereby increasing the intermodulation DD while maintaining high sensitivity and, therefore, increasing the noise immunity of receiving PRC messages.

Thus, the newly introduced blocks and devices provide an increase in the sensitivity of DDs for intermodulation and reliability, which improves the noise immunity of receiving PRC messages and obtain a technical result.

The structural diagram of the proposed receiving radio center is shown in Fig. 1, according to which the PRC contains AC 1, consisting of n directional antennas 1 ₁ ... 1 _n , the outputs of which are respectively connected to n antenna inputs of the MPU 2 ₁ ... MPU 2 _n , inputs / outputs which are connected to the corresponding n outputs / inputs of FOCL 3 ₁ ... FOCL 3 _n , the inputs / outputs of which are connected to the corresponding n outputs / inputs of BOC 4 ₁ ... BOC 4 _n , the other n inputs / outputs of which are connected to the corresponding n outputs / inputs of LAN 5 , the other input / output of which is connected to the output / input of the MCC 6, each of MRPU 2 ₁ ... MRPU 2 _P comprises RT 7, the input of which is the input of each MRPU 2 ₁ ... MRPU 2 _n, ac 1 to m outputs VU 7 connected to the corresponding 1 to m inputs AK August ₁ ... AK 8 _m , from 1 to m signal outputs and from 1 to m input / output control and monitoring buses which are connected to the corresponding 1 to m inputs and 1 to m output / input buses of the first multiplexer / demultiplexer 9, another output / input bus which is connected to the input / output bus VU 7, and the input / output of the first multiplexer / demultiplexer 9 is connected to the output / in the first optoelectronic / electron-optical converter 10, the other optical output / input of which is connected to the input / output of the first optical transceiver 11, the other input / output of which is the input / output of each MPPU 2 ₁ ... MPPU 2 _n , while VU 7 contains 1BUK 12, connected in series to UBZ 13, UBU 14, the input of UBZ 13 is the input of VU 7, and the output of UBU 14 is connected to the input of USU 15, from 1 to m of which outputs are outputs of VU 7, the control / monitoring inputs and outputs of UBZ 13, UBU 14 and USU 15 are connected to the corresponding outputs / in 1BUK 12 moves, the input / output bus of which is VU 7 input / output bus, and each AK 8 ₁ ... AK 8 _m contains 1 BPPF 16, whose input is the input of each AK 8] ... AK 8 t, control input / output and control 1BPFP 16 is connected to the corresponding output / input 2BUK 17, and the output 1BPFP 16 is connected to the input UURCH 18, the input / output of control and control of which is connected to the corresponding output / input 2BUK 17, and the output is connected to the input 2BPF 19, input / output control and monitoring which is connected to the corresponding output / input 2BUK 17, and the output is connected to 1UA 20, the control / monitoring input / output of which is connected to the corresponding output / input of 2BUK 17, and the output is connected to the 1UK 21 input, the control and monitoring input / output of which is connected to the corresponding output / input of 2BUK 17, the first 1UK 21 output is connected to the input of the frequency converter 22, containing SM 23, UChF 24 and 1 BFPC 25, the input of the RF 22 is the first input of the CM 23, the second input of which is connected to the output of the frequency band 24, and the output of the SM 23 is connected to the input 1 BFPC 25, the output of which is the output of the RF 22, corresponding I / O control and monitoring of the inverter 22, which are inputs s / outputs, respectively, USCH 24 and 1 БФПЧ 25, connected to the corresponding outputs / inputs of 2 БУК 17, the output of ПЧЧ 22 is connected to the input of УУЧ 26, the input / output of control and control of which is connected to the corresponding output / input of 2 БУК 17, and the output is connected to the input 2 ББПЧ 27, the input / output of control and monitoring 2BFPCH 27 is connected to the corresponding output / input 2BUK 17, and the output is connected to the first input 2UK 28, the input / output of control and monitoring which is connected to the corresponding output / input 2BUK 17, the second input 2UK 28 is connected to the second output 1UK 21, and the output is connected n with input 2UA 29, the control / monitoring input / output of which is connected to the corresponding output / input of 2BUK 17, and the output is connected to the BACP input 30, the input of which is the URS 31 input, the control / monitoring input / output of which is connected to the corresponding output / input 2BUK 17, ak of outputs are connected respectively to k inputs of UACP 32, other k inputs for analog signals, k inputs for digital signals and input for analog signal with sampling frequency of UACP 32 are respectively connected to corresponding outputs of CACC 33, control inputs / outputs UAACP 32 and CACS 33 are connected to the corresponding outputs / inputs 2BUK 17, k outputs of UACP 32 are connected to k inputs of USSC 34, the output of which is the output of BACP 30 and the corresponding output of each of the AC 8 ₁ ... AK 8 _m , and the input / output control and monitoring system USTSS 34 is connected to the corresponding output / input 2BUK 17, the input / output control and control bus of which is the corresponding input / output bus of each of the AK 8 ₁ ... AK 8 _m , and the input / output of each of the BF 4 ₁ ... BF 4 _n is the input / output of the second optical transceiver 35, the other input d / output of which is connected to the optical output / input of the second optoelectronic / electron-optical converter 36, another electrical input / output of which is connected to the output / input of the second multiplexer / demultiplexer 37, m outputs and m input / output buses of which are connected respectively to m inputs and m output / input buses MK 38, the L outputs of which are connected to the L inputs of the BCRPU 39, which is connected by input / output buses to the output / input buses, respectively, of the memory 40 and UDDS 41, other input / output buses U CP 41 are connected to respective output / input busses 38 and IC memory 40, and input / output UDDS 41 is an input / output of each BOS BOS 4 ₁ ... 4 _n.

The work of the proposed PRC, designed to receive radio signals at different and changing frequencies from different and changing directions, is as follows.

Using CCAM 6 or UDDS 41, the control and monitoring center is controlled, namely, the direction of the radio reception is carried out by selecting one of the MPPU 2 ₁ ... MPPU 2 _n , each of which is directly connected to one of the antennas 1 ₁ ... 1 _n AC 1, setting the number AK 8 ₁ ... AK 8n in each of the MPU 2 ₁ ... MPU 2 _n , setting the necessary operating modes in VU 7, setting the tuning frequencies, setting the structure and operating modes in each of the AK 8 ₁ ... AK 8 _n and in each of the BF 4 ₁ ... BOC 4 _n , distribution of information signals on LAN 5 between BFB 4 ₁ ... BFB 4 _n and CCAM 6. Moreover, priority management and control is assigned to CCAM 6, and UDDS 41 from any BOC 4 ₁ ... BF 4 _n is used if necessary. In the course of the PRC's operation, the corresponding control commands from CCAM 6 or UDS 41 enter the LAN 5 and then through the corresponding BOC 4 ₁ ... BOC 4 _n and the corresponding FOCL 3 ₁ ... FOCL 3 _n are transmitted to the corresponding MPPU 2 ₁ ... MPPU 2 _n , where in 1BUK 12, VU 7 and in 2BUK 17 of the corresponding AK 8 ₁ ... AK 8 _n , each of the MPU 2 ₁ ... MPU 2 _n . In 1BUK 12 and 2BUK 17 command signals are converted into executive signals, which are received in the corresponding blocks and devices for executing control commands. After executing the commands, control signals are generated in the corresponding blocks and devices, which are sent to the corresponding 1BUK 12 and 2BUK 17, where the common control signal flows are generated, which are transmitted from the outputs of the corresponding MPPU 2 ₁ ... MPPU 2 _n through the corresponding FOCL 3 ₁ ... FOCL 3 _n , corresponding to BOC 4 ₁ ... BF 4 _n in LAN 5 and further in CCAM 6 or UDS41, where in a computer they appear in the form of visual and audio signaling about the execution of commands. After making the necessary settings and settings in the PRC, the radio signals from the n 1 antennas 1 ₁ ... 1 _n entering the AS enter directly the n antenna inputs of the MPU 2 ₁ ... MPU 2 _n , while only one of the antennas 1 ₁ ... 1 _{n is} connected MRPU MRPU 2 ₁ ... 2 _n, for example, antenna 1 ₁ is connected antenna input MRPU February _1. The number n is determined by the number of served directions of the radio reception. When performing radio reception from different and changing directions of the antenna, its DDs should overlap all the necessary directions in volumetric space and all necessary polarizations of the radio waves of the received radio signals, while the adjacent antenna bottoms should overlap at half the maximum power of the received signal in the direction of the DL data, i.e. half the width of each beam at a level of 0.707 from the maximum level of the emf of the received signal.

Thus, when using antennas having the same beam width, the maximum number of antennas n _{max is} calculated by the formula:

, $$n_{\max }=p\times \frac{2\pi }{\Delta \varphi /2}\times \frac{\pi /2}{\Delta \theta /2}=\frac{p\mathrm{four}{\pi }^2}{\Delta \varphi \Delta \theta }$$

,

where Δφ (rad) is the beam width of each antenna in the azimuthal plane,

2π is the maximum angle in the azimuthal plane,

Δθ (rad) is the beam width of each antenna in the elevation plane,

π / 2 - maximum angle in elevation plane,

p is a coefficient taking into account the number of polarizations of radio waves received at the corresponding antennas.

The number of polarizations of radio waves is usually several - these are horizontal, vertical, elliptical, flat, linear, circular, right-handed, left-handed, etc. In addition, the nature of the polarization can change and become mixed in the process of propagation of radio waves.

For example, to receive horizontally polarized and vertically polarized radio waves: p = 2, then the number of antennas should be equal to:

. $$n_{\max }=\frac{2\times \mathrm{four}{\pi }^2}{\Delta \varphi \Delta \theta }=\frac{8{\pi }^2}{\Delta \varphi \Delta \theta }$$

.

Given the economic and structural-technical capabilities, as well as the a priori known location and nature of the radiation of some IRIs, it is enough to create a circular in the azimuthal plane of the beam with horizontal polarization, which is composed of separate directed beams in the azimuth plane with the corresponding polarizations, then the minimum number of antennas n _min calculate according to the formula:

, $$n_{\min }=\frac{\mathrm{four}\pi }{\Delta \varphi }$$

,

it is obvious that AC 1 should contain such a number of antennas 11 ... 1 _n that should provide the ND that covers all the necessary directions in volume space and all the necessary polarizations of the received radio signals and at the same time should take into account economic and structural-technical capabilities, therefore n is determined from the condition:

. $$\frac{\mathrm{four}\pi }{\Delta \varphi }\le n\le p\frac{\mathrm{four}{\pi }^2}{\Delta \varphi \Delta \theta }$$

.

The antenna input of each of the MPU 2 ₁ ... MPU 2 _n is the input VU 7, the input of which is the input UBZ 13, UBZ 13 is designed to protect the MPPU from the powerful input effects of the EMF, including the frequency of the received radio signal. When a radio signal from any of the antennas 1 ₁ ... 1 _n with a power exceeding the response threshold set in UBZ 13 is exposed, in UBZ 13 a bypass mode is instantly created at the input of the corresponding from MPPU 2 ₁ ... MPPU 2 _n , and then it turns off the corresponding one from antennas 1 ₁ ... 1 _n from the input circuit of the VU 7. The threshold level of the UBZ 13 is set using 1BUK 12 so as to prevent damage to the input circuit of the VU 7. The threshold is set in CCU 6 via LAN 5 corresponding to BOC 4 ₁ ... BOC 4 _n corresponding RL 3 ₁ ... 3 FOCL _n, optical transceiver 11, the optoelectronic / electro-optical converter 10, a multiplexer / demultiplexer 9 and the control bus and controls the respective slave 7 MRPU MRPU 2 ₁ ... 2 _n. When the power level of the input radio signal from the antennas 1 ₁ ... 1 _n below the threshold UBZ 13 is reduced, it is automatically reset. In addition, with the help of UBZ 13, if necessary, the level of the input radio signal is reduced by turning on the attenuators. From the output of UBZ 13, the received radio signal is fed to the input of UBU 14, in which the radio signal is amplified by power in order to compensate for the inevitable power losses in the USU 15, in the USU 15 it is possible to adjust the number of outputs from 1 to m, while m is the maximum number of AK 8 ₁ ... AK 8 _m in each of the MPPU 2 ₁ ... MPPU 2 _n and may be different, i.e. numbers corresponding to m ₁ ... m _{n, -} is the maximum number of AA in each of the respective MRPU MRPU 2 ₁ ... 2 _n.

The number of outputs of USU 15 is set, from 1 to m, and the corresponding transmission coefficient of UBU 14 is set depending on the current EMO and the number of connected AK 8 ₁ ... AK 8 _m in each of the MPU 2 ₁ ... MPU 2 _n , which depends on the number of IRI and the distribution of radio signals in the operating range of frequencies in this direction. The regulation of the number of connected AK 8 ₁ ... AK 8 _m in each MPU 1 ₁ ... MPU 2 _n , the number of outputs of the USU 15 and the transmission coefficient UBU 14 is performed using 1BUK 12 according to the commands received from the CCAM 6 or UDS 41. Since when using narrow-band AK 8 ₁ ... AK 8 _{m the} number m in each of the MPU 2 ₁ ... MPU 2 _n , which is respectively equal to m ₁ ... m _n, is determined by the width of the operating frequency range equal to Δf, the width of the PP of one narrow-band AK equal to ΔF _min , and the antenna load factor of the antenna, equal to Kz, which for each of the antennas 1 ₁ ... 1 _{n is} respectively equal to KZ ₁ ... Kz _n , therefore, the number m is calculated by the formula:

, $$m_{\mathrm{one}}...m_n=\left(K{s}_{\mathrm{one}}...K{s}_n\right)\frac{\Delta f}{\Delta F_{\min }}$$

,

where Δf is the range of working radio frequencies,

ΔF _min - the minimum width of the PP AK,

Kz ₁ ... Kz _n are the load factors of the DN of the corresponding antennas 1 ₁ ... 1 _n from AC 1.

In this case, the value of each of Kz ₁ ... Kz _n is defined as the ratio of the number of radio signals in the working range of frequencies that are in the coverage area of the beam of each of the corresponding antennas 1 ₁ ... 1 _n to the total number of radio signals in the working range of frequencies that are in full volume space.

The load factor DN (Kz) for each of the antennas 1 ₁ ... 1 _n , respectively equal to Kz ₁ ... Kz _n , is determined by the method of measurements and studies of EMO at the receiving site.

However, under conditions of uniform distribution in the IRI space and for identical values of the beam widths in the azimuthal and elevation planes of each antenna, Kz for each of the antennas are equal to each other: Kz ₁ = Kz ₂ = Kz ₃ = ... = Kz _n = Kz, then we determine analytically as the ratio of the width of the beam in the azimuthal and elevation planes of each antenna to the maximum angles in the azimuthal and elevation planes of the entire volume space according to the formula:

, $$Ks=\frac{\Delta \varphi }{2\pi }\times \frac{\Delta \theta }{\pi /2}=\frac{\Delta \varphi \Delta \theta }{{\pi }^2}$$

,

it is obvious that when Kz ₁ = Kz ₂ = Kz ₃ = ... = Kz _n = Kz,

m ₁ = m ₂ = m ₃ = ... = m _n = m, which is determined by the formula:

, $$m=Ks\frac{\Delta f}{\Delta F_{\min }}=\frac{\Delta \varphi \Delta \theta }{{\pi }^2}\times \frac{\Delta f}{\Delta F_{\min }}$$

,

however, when using the maximum width of the PP in AK 8 ₁ ... AK 8 _m equal to ΔF _max , the number m is determined without taking into account Kz by the formula:

, $$m=\frac{\Delta f}{\Delta F_{\max }}$$

,

where ΔF _max is the maximum width of the PP AK.

Since ΔF _max >> ΔF _min , the corresponding number of analog channels m ₁ ... m _n in each of the corresponding MPPU 2 ₁ ... MPPU 2 _n is determined from the condition:

, $$\frac{\Delta f}{\Delta F_{\max }}\le \left(m_{\mathrm{one}}...m_n\right)\le \left(K{s}_{\mathrm{one}}...K{s}_n\right)\frac{\Delta f}{\Delta F_{\min }}$$

,

therefore, the number of AK equal to m in this MRPU of the receiving radio center is chosen according to the formula:

, $$\frac{\Delta f}{\Delta F_{\max }}\le m\le \mathrm{TO}s\frac{\Delta f}{\Delta F_{\min }}$$

,

where m is the number of AK in this MPPU,

Kz - load factor of the bottom of the given antenna connected to this MPPU.

From the m outputs of the USU 15, the radio signals arrive at the m inputs of the corresponding AK 8 ₁ ... AK 8 _m , the input of each AK is the input 1 BPPF 16, in the controlled units 1 BPPF 16, UURCH 18, 2 BPPF 19, 1UA 20 the preliminary amplification and frequency selectivity of the radio signals are performed, at this 1BPTP 16 and 2 BPPF 19 are made identically and contain a set of switched and tunable PF with different widths of the PP, and UURCH 18 is a controlled multi-stage amplifier of radio signals with a controlled transmission coefficient for power, in these blocks the frequency tuning within the operating frequency range, the width of the PP bandpass filters is adjusted, the transmission coefficient for power corresponding to AK 8 ₁ ... AK 8 _{m is} changed in accordance with the EMO at the receiving location, and all adjustments are made using 2BUK 17, then the radio signal through 1UK 21 is supplied either to the input of PrCh 22 or to the second input of 2UK 28. With the help of 1UK 21, 2UK 28 and 2BUK 17, in order to optimize, each AK 8 ₁ ... AK 8 _{m is} structurally changed, or in a superheterodyne type AK with one frequency conversion in PrCh 22, either in direct amplification AK. In the path with frequency conversion, the radio signal after preliminary amplification and selectivity is fed to the input 1UK 21, in which its input is connected to the first output, while the second output is disconnected from the input, then the radio signal goes to the input of the RF 22, where, using SM 23, USCH 24, 1BFPCh 25 converts the radio frequency of the received radio signal into an intermediate frequency (IF), USCH 24 controlled by CCAM 6, LAN 5, UDS 41 and 2BUK 17, performs the synthesis of signals necessary for tuning the AK to any frequency of the operating range. In USCH 24, the local oscillator signal of the corresponding frequency is synthesized, while the shape, levels and frequencies of the synthesized signals in the USCH 24 are adjusted using 2BUK 17. The signal from the output of the USCH 24 is supplied to SM 23 to convert the received radio signal with a frequency equal to the tuning frequency of the corresponding AK 8 ₁ ... AK 8 _m , into the IF signal, which is filtered out at 1 BFFCH 25 and then amplified and filtered, respectively, at UPCH 26 and 2 BFFCH 27, while 1 BFFCH 25 and 2 BFFCH 27 are the same and represent a set of switched FCFs with different values of П P, and UCHF 26 is a multi-stage amplifier of intermediate frequency signals with a controlled gain. By switching the IF filters in 1BFPCH 25 and 2BFPCH 27, the width of the PCB corresponding to from AK 8 ₁ ... AK 8 _m is regulated, and in UUPCH 26 the transmission coefficient corresponding to from AK 8i ... AK8m is adjusted in accordance with the EMO at the receiving place, moreover, all adjustments are carried out using 2BUK 17. Next, the signal is supplied to the first input of 2UK 28, in which its first input is connected to the output, while the second input is disconnected from the output, from the output of 2UK 28, the signal goes to input 2UA 29, which changes the coefficient AK transmission using 2BUK 17 in the car in automatic or automated modes, from the output of 2УА 29, the signal is fed to the input of the BACP 30, the input of which is the input of the URS 31, from the k outputs of which the signal is fed to the k inputs of the UACP 32. In the direct amplification path, the received radio signal after preliminary amplification and selectivity in 1 BPPF 16 , UURCH 18, 2BPPF 19, 1UA 20 from the output 1UA 20 goes to the input 1UK 21, in which its input is connected to the second output, and the first output is disconnected from the input, then the radio signal goes to the second input 2UK 28, in which it is connected at of the input with the output, the first input is disconnected from the output, then the signal from the 2UK 28 output goes to the 2UA 29 input, with which the transmission coefficient is adjusted according to the signal power, from the 2UA 29 output the signal goes to the BACP 30 input, control of all incoming in BACP 30, blocks and devices are carried out in an automated mode according to commands received from CCAM 6 or UDS 41 using 2BUK 17, in automatic mode according to commands received from blocks and devices of BACP 30 using 2BUK 17.

The block diagram of the BACP 30 is shown in Fig.2.

The input of BACP 30 for connecting 2UA 29 is the input of URS 31, while URS 31 contains a series-connected block of controlled splitters of the received radio signal, a block of controlled analog phase shifters, and the input of URS 31 is the input of the block of controlled splitters of the received radio signal, k outputs of which are connected to the corresponding k inputs block of controlled analog phase shifters, k outputs of which are connected to k inputs of UAATs 32, whose inputs are inputs of each of k analog adders, other inputs are each about which, being k inputs for analog noise or wobbling signals of UACP 32, are connected to each of k outputs of analog signals of CACC 33, and the output of each of k analog adders is connected to the input of each of k ADCs, other inputs for a signal with sampling frequency of each of k ADCs are connected to each of the corresponding k outputs of the block of controlled phase shifters for a signal with a sampling frequency, k inputs of which are connected to k outputs of a signal splitter with a sampling frequency, the input of which, being an input for analogs of the signal with the sampling frequency of UACP 32, connected to the corresponding output of CACC 33, and the outputs of each of k ADCs are connected to the corresponding inputs of each of k digital subtractors, the other inputs of each of which, being inputs for digital noise or wobble signals of UACP 32, are connected to each of the k outputs of the corresponding digital signals of the CACC 33, the outputs of each of the k digital subtractors, being the k outputs of the ADC 32, are connected to the k inputs of the CACC 34, which are the k inputs of the block of controlled digital phase shifters, k outputs of which They are connected to k inputs of the block of controlled digital adders, the output of which is the output of the USSC 34 and BACP 30 for connecting the multiplexer / demultiplexer 9. In this case, the input / output control and monitoring buses of the block of controlled splitters of the received signal, the block of analog phase shifters through the control input / output and control URS 31; input / output buses for control and monitoring of a signal splitter with a sampling frequency, a block of controllable phase shifters for a signal with a sampling frequency by means of a control input / output and control of UAPP 32; input / output control and monitoring buses of a unit of controlled digital phase shifters, a unit of controlled digital adders by means of an input / output of control and monitoring of the USSC 34 are connected to the corresponding outputs / inputs of 2BUK 17.

Implementations of URS 31 may be different, however, the main technical requirements for URS 31 parameters are minimal dissipative losses, i.e. the noise coefficient should tend to unity and the transmission coefficient for power without branching should also tend to unity, therefore, the transmission coefficient for power for each branch of the URS 31 should be inversely proportional to the number k of branches, in addition, the URS 31 should have a high dynamic range in intermodulation.

At the input complex resistance of URS 31, equal to Z _VHURS , a signal oscillates with the power P _cURS = u ² _URS / Z _vhURS = i ² _URS × Z _vhURS , where u _URS , i _{URS are the} effective values of the voltage and current at the input of URS 31, respectively. Next, we use only u _URS and take Z _vkhURS = R _vkhURS , where R _vkhURS is the active input resistance of URS 31.

In URS 31, in the block of controlled splitters of the received radio signal, the signal is branched into k signals, then in the block of controlled analog phase shifters, either the required phase shift of the oscillations between the signals of certain branches is carried out, or the phase shift of the signal oscillations is not carried out, then the oscillation of k signals along k branches from the output of the block of controlled analog phase shifters, which are the outputs of the URS 31, go to the k inputs of UATSP 32, the inputs of which are the inputs of k analog adders, each about which is connected to the signal input of each of k ADCs, one of k analog noise or wobble signals from CACS 33 is supplied to the other input of each of k analogue adders, the corresponding digital subtractor is connected to the output of each of k ADCs one at the other input of which out of k digital noise or wobble signals from CACS 33, the outputs of k digital subtracters are k outputs of the UACP 32, in addition, the UACP 32 contains a series-connected signal splitter with a sampling frequency and a control unit of phase shifters for a signal with a sampling frequency, a signal with a sampling frequency is supplied from CACC 33 to the input of a signal splitter with a sampling frequency, and from the output of a block of controlled phase shifters for a signal with a sampling frequency, each of k signals is fed to each input for a signal with a sampling frequency of k ADC.

The power of each of k input signals for the received signal of each of k ADCs is equal to:

P _sACP = P _sURS K _rURS K _rSUM / k,

where P _surs - signal power at the input of the URS 31,

K _rURS - power transfer coefficient URS 31,

To _rSUM - transmission coefficient for the power of the analog adder at the input of each ADC,

at the same time, at the input complex resistance Z of the _{ADC of} each of the k ADCs, which is taken equal to the active resistance (Z _ADC = R _ADC ), the received signal oscillates with the effective voltage:

$$\begin{array}{l}{u}_{\mathrm{from}\mathrm{BUT}\mathrm{Ts}P}={\left[P_{\mathrm{from}\mathrm{At}R\mathrm{FROM}}{K}_{p\mathrm{At}R\mathrm{FROM}}{K}_{\mathrm{from}\mathrm{FROM}\mathrm{At}M}{R}_{\mathrm{BUT}\mathrm{Ts}P}/k\right]}^{\raisebox{1ex}{$\mathrm{one}$}\!\left/ \!\raisebox{-1ex}{$2$}\right.}={\left[u^2{}_{\mathrm{At}R\mathrm{FROM}}{K}_{p\mathrm{At}R\mathrm{FROM}}{K}_{p\mathrm{FROM}\mathrm{At}M}{R}_{\mathrm{BUT}\mathrm{Ts}P}/\left(R_{\mathrm{at}x\mathrm{At}R\mathrm{FROM}}\right)k\right]}^{\raisebox{1ex}{$\mathrm{one}$}\!\left/ \!\raisebox{-1ex}{$2$}\right.}=\\ \left[u_{\mathrm{At}R\mathrm{FROM}}{\left(K_{p\mathrm{At}R\mathrm{FROM}}{K}_{p\mathrm{FROM}\mathrm{At}M}{R}_{\mathrm{BUT}\mathrm{Ts}P}/R_{\mathrm{at}x\mathrm{At}R\mathrm{FROM}}\right)}^{\raisebox{1ex}{$\mathrm{one}$}\!\left/ \!\raisebox{-1ex}{$2$}\right.}\right]/k^{\raisebox{1ex}{$\mathrm{one}$}\!\left/ \!\raisebox{-1ex}{$2$}\right.}=u_{\mathrm{At}R\mathrm{FROM}}a/k^{\raisebox{1ex}{$\mathrm{one}$}\!\left/ \!\raisebox{-1ex}{$2$}\right.}\end{array}$$

where K _rURS K _rSUM K _ADC / R _vkhURS ) ^½ = a - coefficient of proportionality.

In this case, the voltage acting at the input of a single ADC without branching the signal is:

u _{с1АЦП =} u _УРС a.

, а так как напряжение, действующее на входе одиночного АЦП без разветвления, равно общему напряжению, действующему на входе УАЦП 32, то квадрат суммарного напряжения на входе УАЦП 32 равен:When the signal branching into URS 31 into k signals and turning on the 32 k ADC in the ADC, the square of the effective voltage at the input of each of the k ADCs is: $$u^2{}_{c\mathrm{BUT}\text{A.}\mathrm{Ts}P}=u^2{}_{\mathrm{from}\mathrm{one}\mathrm{BUT}\mathrm{Ts}P}/k$$

, and since the voltage acting at the input of a single ADC without branching is equal to the total voltage acting at the input of the ADC 32, the square of the total voltage at the input of the ADC 32 is:

. $$u^2{}_{c\mathrm{At}\text{A.}\mathrm{BUT}\text{A.}\mathrm{Ts}P}=u^2{}_{\mathrm{from}\mathrm{one}\mathrm{BUT}\mathrm{Ts}P}=k{u}^2{}_{\mathrm{from}\mathrm{BUT}\text{A.}\mathrm{Ts}P}$$

.

After analog-to-digital conversion, the signals are fed to the k inputs of the USSC 34, which contains a series-connected block of controlled digital phase shifters and a block of controlled digital adders, the k inputs of the USSC 34 are k inputs of the block of controlled digital phase shifters, and the output of the USSC 34 is the output of the block of controlled digital adders. In USSC 34, where in the block of controlled digital phase shifters, either the required phase shift between the digital signals of certain branches is carried out, or the phase shift between digital signals is not carried out, then k digital signals are transmitted along k branches to k inputs of the block of controlled digital adders, where the signals are summed in phase, while the square of the effective voltage value of the total oscillation of the useful signal at the output of the USSC 34 is determined by the formula:

$$u^2{}_{\mathrm{from}\Sigma }={\left(u_{\mathrm{from}\mathrm{BUT}\text{A.}\mathrm{Ts}P\mathrm{one}}+u_{\mathrm{from}\mathrm{BUT}\text{A.}\mathrm{Ts}P2}+u_{\mathrm{from}\mathrm{BUT}\text{A.}\mathrm{Ts}P3}+...+u_{\mathrm{from}\mathrm{BUT}\text{A.}\mathrm{Ts}Pk-\mathrm{one}}+u_{\mathrm{from}\mathrm{BUT}\text{A.}\mathrm{Ts}Pk}\right)}^2,$$

.when u _sACP1 = u _sACp2 = u _sACP3 = ... = u _{sACP k-1} = u _sACP k = u _sACP and phase- _wise signal oscillations, $$u^2{}_{\mathrm{from}\Sigma }=k^2\times u^2{}_{c\mathrm{BUT}\text{A.}\mathrm{Ts}P}=k\times u^2{}_{c\mathrm{one}\mathrm{BUT}\text{A.}\mathrm{Ts}P}=k\times u^2{}_{\mathrm{from}\mathrm{At}\mathrm{BUT}\text{A.}\mathrm{Ts}P}$$

.

, где q - высота ступени квантования, которая определяется как:At the same time, in the USSC 34 summation of the quantization noise of each of the k ADCs is carried out. The square of the effective noise voltage of quantization of a single ADC is determined by the formula: $$u^2{}_{w\mathrm{one}\mathrm{BUT}\text{A.}\mathrm{Ts}P}=q^2/12$$

where q is the height of the quantization step, which is defined as:

q = U _pp / (2 ^N -1),

where N is the digitization length of the ADC,

U _pp - the maximum permissible amplitude of the voltage acting on the input of the ADC, and U _pp = 2U _in , where U _in - the maximum allowable amplitude of the input voltage, while the maximum allowable power at the input of a single ADC is:

, таким образом: $$P_{c\mathrm{one}\mathrm{BUT}\text{A.}\mathrm{Ts}Pd\mathrm{about}P}=U^2{}_{\mathrm{at}x}/\left(2R_{\mathrm{BUT}\text{A.}\mathrm{Ts}P}\right)$$

, in this way:

, $$u^2{}_{w\mathrm{one}\mathrm{BUT}\text{A.}\mathrm{Ts}P}=U^2{}_{pp}/\left[12\times {\left(2^N-\mathrm{one}\right)}^2\right]=U^2{}_{\mathrm{at}x}/\left[3\times {\left(2^N-\mathrm{one}\right)}^2\right]$$

,

: $$u^2{}_{c1А\text{}ЦП}/u^2{}_{ш1А\text{}ЦП}=U^2{}_{вх}/\left(2u^2{}_{ш1А\text{}ЦП}\right)=\left(3/2\right)\times {\left(2^N-1\right)}^2$$

,signal-to-noise ratio at the output of a single ADC at $${\left(U^2{}_{\mathrm{at}x}/2\right)}^{\mathrm{one}/2}=u_{\mathrm{from}\mathrm{one}\mathrm{BUT}\text{A.}\mathrm{Ts}P}$$

: $$u^2{}_{c\mathrm{one}\mathrm{BUT}\text{A.}\mathrm{Ts}P}/u^2{}_{w\mathrm{one}\mathrm{BUT}\text{A.}\mathrm{Ts}P}=U^2{}_{\mathrm{at}x}/\left(2u^2{}_{w\mathrm{one}\mathrm{BUT}\text{A.}\mathrm{Ts}P}\right)=\left(3/2\right)\times {\left(2^N-\mathrm{one}\right)}^2$$

,

.on a logarithmic scale: $$101g\left(u^2{}_{c\mathrm{one}\mathrm{BUT}\text{A.}\mathrm{Ts}P}/u^2{}_{w\mathrm{one}\mathrm{BUT}\text{A.}\mathrm{Ts}P}\right)\approx 1.76+6.02N\left(dB\right)$$

.

Since the amplitudes and phases of the oscillations of the quantization noise of each ADC are random in nature, then assuming the absolute absence of mutual correlation, the square of the effective voltage value of the total quantization noise of the ADC 32 at the output of the USSC 34 is determined by the formula:

$$u^2{}_{w\Sigma }=u^2{}_{w\mathrm{one}}+u^2{}_{w2}+u^2{}_{w3}+...+u^2{}_{wk-\mathrm{one}}+u^2{}_{w\mathrm{to}}$$

, $$u^2{}_{ш\Sigma }=k\times u^2{}_{ш1А\text{}ЦП}$$

,at $$u^2{}_{w\mathrm{one}}=u^2{}_{w2}=u^2{}_{w3}=...=u^2{}_{k-\mathrm{one}}=u^2{}_{w\mathrm{to}}=u^2{}_{w\mathrm{one}\mathrm{BUT}\text{A.}\mathrm{Ts}P}$$

, $$u^2{}_{w\Sigma }=k\times u^2{}_{w\mathrm{one}\mathrm{BUT}\text{A.}\mathrm{Ts}P}$$

,

the signal-to-noise ratio at the output of USSC 34 is equal to:

$$u^2{}_{\mathrm{from}\Sigma }/u^2{}_{w\Sigma }=k{u}^2{}_{c\mathrm{At}\text{A.}\mathrm{BUT}\mathrm{Ts}P}/\left(k{u}^2{}_{w\mathrm{one}\mathrm{BUT}\text{A.}\mathrm{Ts}P}\right)=u^2{}_{c\mathrm{At}\mathrm{BUT}\text{A.}\mathrm{Ts}P}/u^2{}_{w\mathrm{one}\mathrm{BUT}\text{A.}\mathrm{Ts}P}=u^2{}_{c\mathrm{one}\mathrm{BUT}\text{A.}\mathrm{Ts}P}/u^2{}_{w\mathrm{one}\mathrm{BUT}\text{A.}\mathrm{Ts}P},$$

thus, the signal-to-noise ratio at the output of the BACC 30 does not change compared to the signal-to-noise ratio at the output of a single ADC, assuming the signals are completely in phase and the output noise is not correlated, 32. Therefore, the number k from the point of view of the permissible effective voltage of the group signal at the input of a single ADC is determined by the formula:

, $$k\ge \frac{u^2{}_{\mathrm{At}R\mathrm{FROM}\max }{a}^2}{u^2{}_{c\mathrm{BUT}\text{A.}\mathrm{Ts}Pd\mathrm{about}P}}$$

,

where u _URSmax is the maximum effective voltage at the input of URS 31,

a is the coefficient of proportionality,

u _sACPdop - permissible effective voltage at the input of a single ADC.

However, when determining the number k, it is necessary to take into account the DD by intermodulation (D) and the BACC noise figure 30. In this case, the DD by intermodulation (D) of a single ADC is equal to: D = Р _с1АЦПи / P ₀ ,

where _{Рс1АЦПи} is the maximum power of one of the two signals at the input of a single ADC, at which the intermodulation interference resulting from the interaction of these two signals within the integral characteristic of the ADC, which is approximated fairly accurately by a power series, reaches a predetermined value P ₀ . On a logarithmic scale, D is determined by the formula:

D (dB) = P _s1ACPi -P ₀ , where P ₀ is determined by the formulas:

P ₀ (dBm) = 2P _s1Atspi (dBm) -IP ₂ (dBm), P ₀ (dBm) = 3P _s1ATsPi (dBm) -2IP ₃ (dBm),

it is obvious that when the input signal power (Р _с1АЦПи ) _changes , the intermodulation interference power (Р ₀ ) at the output of a single ADC changes on a logarithmic scale in proportion to the order of intermodulation, while:

P _s1ATsPi (dBm) = (P ₀ + IP ₂ )) / 2, P _s1ATsPi = (P ₀ + 2 IP ₃ ) / 3,

where IP ₂ , IP ₃ are the constant intermodulation parameters of the nonlinearity of the second and third orders within the integral characteristics of the ADC, respectively, these parameters measure or use known values. Thus, the DD for intermodulation of the second and third orders of a single ADC are respectively equal:

D _2ACP = (IP ₂ -P ₀ ) / 2 (dB); D _3ACP = 2 (IP ₃ -P ₀ ) / 3 (dB).

As a result of the branching of the input signal into k signals, the power of each of the k signals P _sACPi , acting at the input of each ADC in the UACP 32, decreases by a factor of k and on a linear scale is equal to: P _sACPi = P _s1ATsPi / k, on a logarithmic scale:

R _sACPi (dBm) = R _s1ATsPi (dBm) -k (dB), while the intermodulation noise of the second and third orders at the output of each ADC are respectively equal, on a linear scale: P ₀₂ = P ₀ / k ² , P ₀₃ = P ₀ / k ³ , on a logarithmic scale:

P ₀₂ (dBm) = 2 (P _s1ATsPi -k) (dBm) -IP ₂ (dBm) = P ₀ -2k (dB),

P ₀₃ (dBm) = 3 (P _s1ATsPi -k) (dBm) -2IP ₃ (dBm) = P ₀ -3k (dB),

further, all the intermodulation interference oscillations of each ADC are in-phase summed in USSC 34 and at the output of the BACP 30 the power of the total second and third order intermodulation interference oscillations on a linear scale are respectively equal to:

P _02∑ = k ² P ₀₂ = k ² (P ₀ / k ² ) = P ₀ ,

P _03∑ = k ² P ₀₃ = k ² (P ₀ / k ³ ) = P ₀ / k.

The power of the total intermodulation noise of the second order at the output of the BACP 30 is equal to the power of the intermodulation interference of the single ADC, and the power of the total intermodulation interference of the third order at the output of the BACP 30 has decreased by a factor of k; ADC:

D _{2 BACP} (dB) = [IP ₂ dB- (P ₀ -k) dB] / 2 = D 2 _AACP (dB) +0 (dB),

and the third-order intermodulation DD of BACP 30 increased compared to a single ADC and is equal to:

D _3BACP (dB) = 2 [IP ₃ dB- (P ₀ -k) dB] / 3 = D _3ACP (dB) + 2k / 3 (dB).

The allowable power of the signals at the input of UACP 32, causing third-order intermodulation interference, increased by 2k / 3 (dB) or (k ² ) ^1/3 times.

Obviously, a simple division of the power of the input signal by k does not increase the DD on the second-order intermodulation. However, when the input signal is branched into an even number of k branches, second-order intermodulation interference can be compensated.

Consider an example of compensation.

Suppose at the input of the URS 31 there is a total oscillation of two signals, when branching in the URS 31 an even number of k signals at the input of each of k ADCs acts on the total voltage fluctuation:

u _in = U ₁ cosω ₁ t + U ₂ cosω ₂ t,

the value of which is within the integral characteristic of the ADC, which is fairly accurately approximated by a power series. Given the well-known trigonometric formulas:

cos ² α = (1 + cos2α) / 2; cos ³ α = (3cosα + cos3α) / 4; 2cosαcosβ = cos (α + β) + cos (α-β),

consider the relative nature of intermodulation noise of the second and third orders, while not taking into account the nonlinearity parameters of the ADC. The quadratic term of the power series is equal to:

$$\begin{array}{l}{\left(U_{\mathrm{one}}\cos {\omega }_{\mathrm{one}}t+U_{2}co{\omega }_{2}t\right)}^2=U_{\mathrm{one}}{}^2{\cos }^2{\omega }_{\mathrm{one}}t+2U_{\mathrm{one}}{U}_2\cos {\omega }_{\mathrm{one}}t\cos {\omega }_{2}t+U_2{}^2{\cos }^2{\omega }_{2}t=\\ =0.5U_{\mathrm{one}}{}^2+0.5U_{\mathrm{one}}{}^2\cos 2{\omega }_{\mathrm{one}}t+U_{\mathrm{one}}{U}_2\cos \left({\omega }_{\mathrm{one}}-{\omega }_2\right)t+U_{\mathrm{one}}{U}_2\cos \left({\omega }_{\mathrm{one}}+{\omega }_2\right)t+0.5U_2{}^2+0.5U_2{}^2\cos 2{\omega }_{2}t\end{array}$$

$$0,5U_2{}^{2}cos2{\omega }_{2}t$$

- вторые гармоники колебаний первого и второго сигнала соответственно;Where $$0,5U_{\mathrm{one}}{}^{2}cos2{\omega }_{\mathrm{one}}t,$$

$$0,5U_2{}^{2}cos2{\omega }_{2}t$$

- second harmonics of the oscillations of the first and second signal, respectively;

U ₁ U ₂ cos (ω ₁ + ω ₂ ) t, U ₁ U ₂ cos (ω ₁ -ω ₂ ) t - oscillations of intermodulation interference with the sum and difference frequencies, respectively. The cubic member of the power series is equal to:

$$\begin{array}{l}{\left(U_{\mathrm{one}}\cos {\omega }_{\mathrm{one}}t+U_2\cos {\omega }_{2}t\right)}^3=U_{\mathrm{one}}{}^3{\cos }^3{\omega }_{\mathrm{one}}t+3U_{\mathrm{one}}{}^2{U}_2{\cos }^2{\omega }_{\mathrm{one}}t\cos {\omega }_{2}t+\\ +3U_{\mathrm{one}}{U}_2{}^2\cos {\omega }_{\mathrm{one}}t{\cos }^2{\omega }_{2}t+U_2{}^3{\cos }^3{\omega }_{2}t=\left(3U_{\mathrm{one}}{}^3/\mathrm{four}\right)\cos {\omega }_{\mathrm{one}}t+\left(U_{\mathrm{one}}{}^3/\mathrm{four}\right)\cos {\omega }_{\mathrm{one}}t+\\ +\left(3/\mathrm{four}\right)U_{\mathrm{one}}{}^2{U}_2\left[2\cos {\omega }_{2}t+\cos \left(2{\omega }_{\mathrm{one}}+{\omega }_2\right)t+\cos \left(2{\omega }_{\mathrm{one}}-{\omega }_2\right)t\right]+\\ +\left(3/\mathrm{four}\right)U_2{}^2{U}_{\mathrm{one}}\left[2\cos {\omega }_{\mathrm{one}}t+\cos \left(2{\omega }_2+{\omega }_{\mathrm{one}}\right)t+\cos \left(2{\omega }_2-{\omega }_{\mathrm{one}}\right)t\right]+\\ \left(3U_2{}^3/\mathrm{four}\right)\cos {\omega }_{2}t+\left(U_2{}^3/\mathrm{four}\right)\cos 3{\omega }_{2}t;\end{array}$$

$$\left(U_2{}^3/4\right)cos3{\omega }_{2}t$$

- третьи гармоники сигналов,Where $$\left(U_{\mathrm{one}}{}^3/\mathrm{four}\right)cos3{\omega }_{\mathrm{one}}t,$$

$$\left(U_2{}^3/\mathrm{four}\right)cos3{\omega }_{2}t$$

- third harmonics of signals,

- интермодуляционные помехи вида 2ω₁±ω₂, $$\left(3/\mathrm{four}\right)U_{\mathrm{one}}{}^2{U}_2\left[\cos \left(2{\omega }_{\mathrm{one}}+{\omega }_2\right)t+\cos \left(2{\omega }_{\mathrm{one}}-{\omega }_2\right)t\right]$$

- intermodulation interference of the form 2ω ₁ ± ω ₂ ,

- интермодуляционные помехи вида 2ω₂±ω₁. $$\left(3/\mathrm{four}\right)U_2{}^2{U}_{\mathrm{one}}\left[\cos \left(2{\omega }_2+{\omega }_{\mathrm{one}}\right)t+\cos \left(2{\omega }_2-{\omega }_{\mathrm{one}}\right)t\right]$$

- intermodulation interference of the form 2ω ₂ ± ω ₁ .

When the phase shifters are turned on, we create the necessary phase shift between the oscillations of the signals of even and odd branches, for example, by k, then at the input of each ADC, for example even branches, according to the well-known formula cos (α + π) = - cosα, we have the oscillations:

U ₁ cos (ω ₁ t + π) = - U ₁ cosω ₁ t, U ₂ cos (ω ₂ t + π) = - U ₂ cosω ₂ t, after analog-to-digital conversion to UAATs 32 these signals are fed to the inputs of the block managed digital adders. In the odd branches after analog-to-digital conversion in UACP 32 using controlled digital phase shifters in the USSC 34 there is also a phase shift between the oscillations of the signals of the odd and even branches by π, then we have the oscillations at the inputs of the odd branches of the block of controlled digital adders: U ₁ cos ( ω ₁ t + π) = - U ₁ cosω ₁ t, U ₂ cosω ₂ π + π) = - U ₂ cosω ₂ t, that is, the phases of the oscillations of the signals of even and odd branches become the same, then the oscillations of the signals of these branches are summed in phase in digital adder.

However, as a result of nonlinear interaction of signals in even branches in the ADC, at the ADC output, second-order oscillations are:

$$\begin{array}{l}{\left(-U_{\mathrm{one}}\cos {\omega }_{\mathrm{one}}t-U_2\cos {\omega }_{2}t\right)}^2=U_{\mathrm{one}}{}^2{\cos }^2{\omega }_{\mathrm{one}}t+2U_{\mathrm{one}}{U}_2\cos {\omega }_{\mathrm{one}}t\cos {\omega }_{2}t+U_2{}^2{\cos }^2{\omega }_{2}t=0.5U_{\mathrm{one}}{}^2+\\ +\mathrm{,5}{U}_{\mathrm{one}}{}^2\cos 2{\omega }_{\mathrm{one}}t+U_{\mathrm{one}}{U}_2\cos \left({\omega }_{\mathrm{one}}-{\omega }_2\right)t+U_{\mathrm{one}}{U}_2\cos \left({\omega }_{\mathrm{one}}+{\omega }_2\right)t+0.5U_2{}^2+0.5U_2{}^2\cos 2{\omega }_{2}t,\end{array}$$

also as a result of nonlinear interaction of signals in odd branches in the ADC, second-order oscillations are:

$$\begin{array}{l}{\left(U_{\mathrm{one}}\cos {\omega }_{\mathrm{one}}t+U_2\cos {\omega }_{2}t\right)}^2=U_{\mathrm{one}}{}^2{\cos }^2{\omega }_{\mathrm{one}}t+2U_{\mathrm{one}}{U}_2\cos {\omega }_{\mathrm{one}}t\cos {\omega }_{2}t+U_2{}^2{\cos }^2{\omega }_{2}t=0.5U_{\mathrm{one}}{}^2+\\ +\mathrm{,5}{U}_{\mathrm{one}}{}^2\cos 2{\omega }_{\mathrm{one}}t+U_{\mathrm{one}}{U}_2\cos \left({\omega }_{\mathrm{one}}-{\omega }_2\right)t+U_{\mathrm{one}}{U}_2\cos \left({\omega }_{\mathrm{one}}+{\omega }_2\right)t+0.5U_2{}^2+0.5U_2{}^2\cos 2{\omega }_{2}t,\end{array}$$

$$0,5U_2{}^{2}cos2{\omega }_{2}t$$

- вторые гармоники колебаний сигналов;Where $$0,5U_{\mathrm{one}}{}^{2}cos2{\omega }_{\mathrm{one}}t,$$

$$0,5U_2{}^{2}cos2{\omega }_{2}t$$

- second harmonics of signal oscillations;

U ₁ U ₂ cos (ω ₁ + ω ₂ ) t, U ₁ U ₂ cos (ω ₁ -ω ₂ ) t are the second-order intermodulation noise with the total and difference frequencies of the signal oscillations, respectively.

Further, when the phase shift of the output digital signals of the ADC of the odd branches by π in the USSC 34, the second-order oscillations of the signals of the odd branches take the form:

$$-0.5U_{\mathrm{one}}{}^2\cos 2{\omega }_{\mathrm{one}}t-U_{\mathrm{one}}{U}_2\cos \left({\omega }_{\mathrm{one}}-{\omega }_2\right)t-U_{\mathrm{one}}{U}_2\cos \left({\omega }_{\mathrm{one}}+{\omega }_2\right)t-0.5U_2{}^2\cos 2{\omega }_{2}t,$$

 that is, they are in antiphase with second-order oscillations of signals in even branches.

When summing the second-order oscillations of the signals of the odd branches with the second-order vibrations of the signals of the even branches, we obtain:

$$\begin{array}{l}-0.5U_{\mathrm{one}}{}^2\cos 2{\omega }_{\mathrm{one}}t-U_{\mathrm{one}}{U}_2\cos \left({\omega }_{\mathrm{one}}-{\omega }_2\right)t-U_{\mathrm{one}}{U}_2\cos \left({\omega }_{\mathrm{one}}+{\omega }_2\right)t-0.5U_2{}^2\cos 2{\omega }_{2}t+\\ +0.5U_{\mathrm{one}}{}^2\cos 2{\omega }_{\mathrm{one}}t+U_{\mathrm{one}}{U}_2\cos \left({\omega }_{\mathrm{one}}-{\omega }_2\right)t+U_{\mathrm{one}}{U}_2\cos \left({\omega }_{\mathrm{one}}+{\omega }_2\right)t+0.5U_2{}^2\cos 2{\omega }_{2}t=0\end{array}$$

it is obvious that the second-order intermodulation noise of the form ω ₁ ± ω ₂ and the second harmonics of the signals 2ω ₁ , 2ω _{2 are} completely compensated.

With a phase shift of the signal oscillations by π in the even branches of the URS 31, third-order oscillations at the output of each ADC are:

$$\begin{array}{l}{\left(-U_{\mathrm{one}}\cos {\omega }_{\mathrm{one}}t-U_2\cos {\omega }_{2}t\right)}^3=-U_{\mathrm{one}}{}^3{\cos }^3{\omega }_{\mathrm{one}}t-3U_{\mathrm{one}}{}^2{U}_2{\cos }^2{\omega }_{\mathrm{one}}t\cos {\omega }_{2}t-\\ -3U_{\mathrm{one}}{U}_2{}^2\cos {\omega }_{\mathrm{one}}t{\cos }^2{\omega }_{2}t-U_2{}^3{\cos }^3{\omega }_{2}t=-\left(3U_{\mathrm{one}}{}^3/\mathrm{four}\right)\cos {\omega }_{\mathrm{one}}t-\left(U_{\mathrm{one}}{}^3/\mathrm{four}\right)\cos 3{\omega }_{\mathrm{one}}t-\\ -\left(3/\mathrm{four}\right)U_{\mathrm{one}}{}^2{U}_2\left[2\cos {\omega }_{2}t+\cos \left(2{\omega }_{\mathrm{one}}+{\omega }_2\right)t+\cos \left(2{\omega }_{\mathrm{one}}-{\omega }_2\right)t\right]-\\ -\left(3/\mathrm{four}\right)U_2{}^2{U}_{\mathrm{one}}\left[2\cos {\omega }_{\mathrm{one}}t+\cos \left(2{\omega }_2+{\omega }_{\mathrm{one}}\right)t+\cos \left(2{\omega }_2-{\omega }_{\mathrm{one}}\right)t\right]-\\ -\left(3U_2{}^3/\mathrm{four}\right)\cos {\omega }_{2}t-\left(U_2{}^3/\mathrm{four}\right)\cos 3{\omega }_{2}t,\end{array}$$

Where

- третьи гармоники сигналов 3ω₁, 3ω₂, $$-\left(U_{\mathrm{one}}{}^3/\mathrm{four}\right)\cos 3{\omega }_{\mathrm{one}}t,-\left(U_2{}^3/\mathrm{four}\right)\cos 3{\omega }_{2}t$$

- third harmonics of the signals 3ω ₁ , 3ω ₂ ,

$$-\left(3/\mathrm{four}\right)U_{\mathrm{one}}{}^2{U}_2\left[2\cos {\omega }_{2}t+\cos \left(2{\omega }_{\mathrm{one}}+{\omega }_2\right)t+\cos \left(2{\omega }_{\mathrm{one}}-{\omega }_2\right)t\right]$$

- интермодуляционные помехи третьего порядка видов: 2ω₁±ω₂, 2ω₂±ω₁. $$-\left(3/\mathrm{four}\right)U_2{}^2{U}_{\mathrm{one}}\left[2\cos {\omega }_{\mathrm{one}}t+\cos \left(2{\omega }_2+{\omega }_{\mathrm{one}}\right)t+\cos \left(2{\omega }_2-{\omega }_{\mathrm{one}}\right)t\right]$$

- third order intermodulation interference of the types: 2ω ₁ ± ω ₂ , 2ω ₂ ± ω ₁ .

Oscillations of the third order at the output of each ADC in the odd branches of UACP 32 are:

$$\begin{array}{l}{\left(U_{\mathrm{one}}\cos {\omega }_{\mathrm{one}}t+U_2\cos {\omega }_{2}t\right)}^3=U_{\mathrm{one}}{}^3{\cos }^3{\omega }_{\mathrm{one}}t+3U_{\mathrm{one}}{}^2{U}_2{\cos }^2{\omega }_{\mathrm{one}}t\cos {\omega }_{2}t+\\ +3U_{\mathrm{one}}{U}_2{}^2\cos {\omega }_{\mathrm{one}}t{\cos }^2{\omega }_{2}t+U_2{}^3{\cos }^3{\omega }_{2}t=\left(3U_{\mathrm{one}}{}^3/\mathrm{four}\right)\cos {\omega }_{\mathrm{one}}t+\left(U_{\mathrm{one}}{}^3/\mathrm{four}\right)\cos 3{\omega }_{\mathrm{one}}t+\\ +\left(3/\mathrm{four}\right)U_{\mathrm{one}}{}^2{U}_2\left[2\cos {\omega }_{2}t+\cos \left(2{\omega }_{\mathrm{one}}+{\omega }_2\right)t+\cos \left(2{\omega }_{\mathrm{one}}-{\omega }_2\right)t\right]+\\ +\left(3/\mathrm{four}\right)U_2{}^2{U}_{\mathrm{one}}\left[2\cos {\omega }_{\mathrm{one}}t+\cos \left(2{\omega }_2+{\omega }_{\mathrm{one}}\right)t+\cos \left(2{\omega }_2-{\omega }_{\mathrm{one}}\right)t\right]+\\ \left(3U_2{}^3/\mathrm{four}\right)\cos {\omega }_{2}t+\left(U_2{}^3/\mathrm{four}\right)\cos 3{\omega }_{2}t\end{array}$$

With a phase shift of the signal oscillations by π in the odd branches of the USSC 34, the third signal harmonics and third order intermodulation interference are, respectively:

, $$-\left(U_2{}^3/4\right)\cos 3{\omega }_{2}t$$

$$-\left(U_{\mathrm{one}}{}^3/\mathrm{four}\right)\cos 3{\omega }_{\mathrm{one}}t$$

, $$-\left(U_2{}^3/\mathrm{four}\right)\cos 3{\omega }_{2}t$$

$$-\left(3/\mathrm{four}\right)U_{\mathrm{one}}{}^2{U}_2\left[2\cos {\omega }_{2}t+\cos \left(2{\omega }_{\mathrm{one}}+{\omega }_2\right)t+\cos \left(2{\omega }_{\mathrm{one}}-{\omega }_2\right)t\right]$$

$$-\left(3/\mathrm{four}\right)U_2{}^2{U}_{\mathrm{one}}\left[2\cos {\omega }_{\mathrm{one}}t+\cos \left(2{\omega }_2+{\omega }_{\mathrm{one}}\right)t+\cos \left(2{\omega }_2-{\omega }_{\mathrm{one}}\right)t\right]$$

that is, oscillations of the third harmonics of signals and third-order intermodulation interference in even and odd branches at the inputs of the digital controlled adders block of the USSC 34 are in phase, that is, branching the signal into an even number of k branches does not compensate for third-order interference.

Thus, branching the received signal into an even number of k branches with a phase change between the oscillations of the signals of the corresponding branches in the URS 31, turning on the ADC in the UAPS 32, changing the phases between the digital signals of the corresponding branches and their in-phase summation in the USSC 34 allows by dividing the input signal power in the block of controlled splitters of the received signal URS 31 to increase the DD on intermodulation of the third order, and by compensating for intermodulation interference of the second order using the block of controlled analog phases rotators URS 31 blocks managed digital phase shifters and block controlled digital adders USTSS 34 increase DD Intermodulation second order BATSP 30 and thus increase DD Intermodulation each of AK 8 ₁ ... AK 8 _m each MRPU 2 ₁ ... MRPU 2 _n and therefore, increase the noise immunity of the PRC.

In BACP 30, according to commands from CCAM 6 or UDS 41 using 2BUK 17 or in automatic mode using 2BUK 17, the number of branches of the received signal is changed from 1 to k by changing the number of connected outputs in URS 31 to the inputs of UACP 32, changing the number of connected corresponding ADCs in UACP 32 and changes in the number of connected inputs in USSC 34 to the outputs of UACP 32. Moreover, in each output branch in URS 31 and the input branch in USSC 34, phase shifters controlled by 2 BUK 17 are created, which create shifts of the initial phases between signal oscillations (ψ) at an angle from 0 to 2π with divided Δψ step _f, by which to compensate the intermodulation interference of various orders, e.g., when the branches in each three phase shifters that create a phase shift between the oscillations in these branches of signals by an angle 2π / 3, etc. In this case, the maximum number k of branches, which will provide an increase in DD by intermodulation of BACP 30 and, accordingly, AK, is determined by the formula:

, $$k\ge \frac{P_{\mathrm{from}\mathrm{BUT}\mathrm{TO}\mathrm{and}}{K}_{R\mathrm{BUT}\mathrm{TO}}{K}_{R\text{A.}\mathrm{At}R\mathrm{FROM}}{K}_{R\text{A.}\mathrm{FROM}\mathrm{At}M}}{R_{\mathrm{from}\mathrm{one}\mathrm{BUT}\mathrm{Ts}P\mathrm{and}}}=\frac{R_{\mathrm{from}\mathrm{BUT}\mathrm{TO}\mathrm{and}}{K}_{p\Sigma }}{R_{\mathrm{from}\mathrm{one}\mathrm{BUT}\mathrm{Ts}P\mathrm{and}}}$$

,

where R _cAKi - the maximum allowable input power of one of the two signals acting on the input of the AC, in which the third-order intermodulation interference resulting from the interaction of these two signals in the AC before the input of the ADC has an acceptable value,

R _s1ATsPi - the maximum allowable power of one of the two signals acting at the input of a single ADC, in which the third-order intermodulation interference resulting from the interaction of these two signals within the integral ADC characteristic has an acceptable value;

K _p∑ is the transmission coefficient for AC power to the input of a single ADC, with K _p∑ = K _pAK K _rURS K _pSUM ,

where K _pAK - transmission coefficient for power AK to the input of the BACP 30,

K _RURS - power transfer coefficient URS 31,

K _rSUM - transmission coefficient for the power of the analog adder at the input of each ADC.

The values of P _sAKi and P _s1ATsPi measure or use known parameters. The larger the number k, the higher the allowable power of the signals at the input of each ADC from UACP 32, the higher the third-order intermodulation DD, and it is advisable to choose the number k based on possible phase combinations between the branched signals to compensate for non-linear interference.

The ability to change the number of branches from 1 to k in the BACP 30 allows you to change the DD by intermodulation and by the BACP 30 signal in accordance with the change in the input signal power, while the increase in the BACP 30 noise figure should be minimal. Therefore, the number k is also selected based on the requirements for a relative increase in the noise factor of the BACP 30 compared to the noise figure of a single ADC. The noise factor of BACP 30 depends on: from dissipative losses and power transfer coefficient of URS 31; from the violation of the in-phase oscillations of the useful signals; from inaccurate subtraction or insufficient filtering of noise or wobble signals at the output of each ADC; from the magnitude of the cross-correlation between the output noise UATSP 32.

The real noise figure of a single ADC, taking into account distortions and residuals of randomization noise, is determined by the formula:

$${\mathrm{TO}}_{w\mathrm{one}\mathrm{BUT}\mathrm{Ts}PR}=\mathrm{one}+\frac{u^2{}_{w\mathrm{one}\mathrm{BUT}\mathrm{Ts}PR}+\Delta u^2{}_{wR}}{2\mathrm{to}{T}_0{R}_{\mathrm{BUT}\mathrm{Ts}P}{F}_d}=\mathrm{one}+\frac{U^2{}_{\mathrm{at}x}+3{\Delta }^2{}_{wR}{\left(2^b-\mathrm{one}\right)}^2}{6\mathrm{to}{T}_0{R}_{\mathrm{BUT}\mathrm{Ts}P}{F}_d{\left(2^b-\mathrm{one}\right)}^2}=\mathrm{one}+\frac{u^2{}_w}{2\mathrm{to}{T}_0{R}_{\mathrm{BUT}\mathrm{Ts}P}{F}_d}$$

- квадрат эффективного напряжения шумов на выходе одиночного АЦП, причем $$u^2{}_{ш}=u^2{}_{ш1АЦП}+\Delta u^2{}_{шр},$$

гдеWhere $$u^2{}_w$$

- the square of the effective voltage noise at the output of a single ADC, and $$u^2{}_w=u^2{}_{w\mathrm{one}\mathrm{BUT}\mathrm{Ts}P}+\Delta u^2{}_{wR},$$

Where

- квадрат эффективного значения напряжения шумов (ошибок) квантования с учетом реальной разрядности квантования АЦП, $$u^2{}_{w\mathrm{one}\mathrm{BUT}\mathrm{Ts}P}$$

- the square of the effective value of the voltage of the noise (error) quantization, taking into account the real digitization capacity of the ADC,

- остаток шума рандомизации, причем $$\Delta u^2{}_{шр}=u^2{}_{шрА}-u^2{}_{шрЦ},$$

где $$\Delta u^2{}_{шрА}$$

и $$\Delta u^2{}_{шрЦ}$$

- квадраты эффективного значения напряжения шумов рандомизации соответственно в аналоговой и в цифровой формах, b - эффективная (реальная) разрядность квантования АЦП, причем $$\Delta u^2{}_{wR}$$

- the remainder of the randomization noise, and $$\Delta u^2{}_{wR}=u^2{}_{wR\mathrm{BUT}}-u^2{}_{wR\mathrm{Ts}},$$

Where $$\Delta u^2{}_{wR\mathrm{BUT}}$$

and $$\Delta u^2{}_{wR\mathrm{Ts}}$$

are squares of the effective value of the voltage of randomization noise, respectively, in analog and digital forms, b is the effective (real) bit depth of the ADC quantization, and

b = (SINAD-1.76dB) / 6.02dB,

where SINAD is the ratio of signal to noise and distortion in decibels (dB), is given in the data on the ADC,

F _d - sampling frequency,

k = 1.38 × 10 ^-23 J / K is the Boltzmann constant,

T ₀ = 293K - standard room temperature (t = 20 ° C),

kT ₀ = 4 × 10 ^-21 W / Hz - the intensity or spectral density of thermal noise at the input of the ADC, is the main unit for measuring noise intensities in blocks and devices of the PRC,

R _ADC - input resistance of the ADC,

U _in - the amplitude of the maximum allowable voltage at the input of the ADC, and U _in = U _pp / 2, where U _pp is the allowable amplitude of the voltage at the input of the ADC, is given in the data on the ADC.

Given that the threshold sensitivity, expressed as the minimum power density of the signal source, consistent with the input of a single ADC, is:

а предельно возможная минимальная спектральная плотность мощности источника теплового шума, согласованного с входом каждого из k АЦП, равна: $$2P_{{\min }_{\mathrm{one}}\mathrm{BUT}\mathrm{Ts}PR}/F_d={\mathrm{TO}}_{w\mathrm{one}\mathrm{BUT}\mathrm{Ts}PR}\mathrm{to}{\mathrm{<figure-callout\; id="0"\; label="T"\; filenames="00000081.png"\; state="\{\{state\}\}">T</figure-callout>}}_0,$$

and the maximum possible minimum spectral power density of the heat noise source, consistent with the input of each of k ADCs, is:

2P _{min AACP} / F _d = kT ₀ , then with absolute phase-matching of the signals in the BACP 30 branches and absolute uncorrelated output noise of the UACP 32, the maximum number k is determined by the formula:

, то есть k=К_ш1АЦПр. $$k=\frac{u^2{}_w}{2\mathrm{to}{T}_0{R}_{\mathrm{BUT}\text{A.}\mathrm{Ts}P}{F}_d}+\mathrm{one}={\mathrm{TO}}_{w\mathrm{one}\mathrm{BUT}\mathrm{Ts}PR}$$

, that is, k = К _ш1АЦПр .

Thus, from the point of view of the sensitivity of the ADC, the maximum number k is limited by the noise figure of a single ADC - K _W1ACPr .

However, in real devices, there are always deviations from the in-phase oscillations of signals and there is a mutual correlation between oscillations of noise signals.

Let us assume that there is a mutual correlation between the output noise in UAZP 32 and there are deviations in the in-phase oscillations of useful signals in BACP 30.

Let us denote the relationship between the fluctuations in the voltage of the output noise k ADC: u _w1 and u _w2 , u _w1 and u _w3 , ... u _w1 and u _wk , u _w2 and u _w3 , u _w2 and u _w4 , ... u _w2 and u _wk , ... u _{w (k-1)} and u _wk , cross-correlation coefficient r _1.2 ... r _{(k-1), k} , the range of which lies in the range from -1 to +1 and which is determined using measurements.

Express: $$\begin{array}{l}{u}_{w\mathrm{one}}\times u_{w2}=u^2{}_{\mathrm{<figure-callout\; id="1"\; label="w"\; filenames="00000081.png"\; state="\{\{state\}\}">w</figure-callout>}\mathrm{1,2}},u_{w\mathrm{one}}\times u_{w3}=u^2{}_{w1.3},...u_{w\mathrm{one}}\times u_{wk}=u^2{}_{w\mathrm{one,}k},u_{w2}\times u_{w3}=u^2{}_{w\mathrm{2,3}},\\ u_{w2}\times u_{w\mathrm{four}}=u^2{}_{w\mathrm{2,4}},...u_{w2}\times u_{wk}=u^2{}_{w2k},...u_{w(k-\mathrm{one})}\times u_{wk}=u^2{}_{w(k-\mathrm{one}),k},\end{array}$$

define the square of the total voltage of the output noise UAZP 32 at the output of the BACP 30 in the general form:

$$u^2{}_{w\Sigma }={\displaystyle \sum _{i=\mathrm{one}}^k{u}^2{}_{wi}}+2\times \left({\displaystyle \sum _{i=2}^k{u}^2{}_{w\mathrm{one,}i}{r}_{\mathrm{one,}i}}+{\displaystyle \sum _{i=3}^k{u}^2{}_{w2i}{r}_{2i}}+...+{\displaystyle \sum _{i=k-\mathrm{one}}^k{u}^2{}_{w\left(k-2\right),i}{r}_{\left(k-2\right),i}}+u^2{}_{w\left(k-\mathrm{one}\right),k}{r}_{\left(k-\mathrm{one}\right),k}\right)$$

where k is the number of ADCs included in UAOC 32,

u _w1 ... u _wk are the effective voltage of the output noise of each ADC.

Provided that the same type of ADCs are included, we assume

that u _w1 = u _w2 = u _w3 = u _w4 = ... = u _wk-1 = u _wk = u _w , hence:

$$u^2{}_{\mathrm{<figure-callout\; id="1"\; label="w"\; filenames="00000081.png"\; state="\{\{state\}\}">w</figure-callout>}\mathrm{1,2}}=...=u^2{}_{w\mathrm{one,}k}=u^2{}_{w\mathrm{2,3}}=...=u^2{}_{w2k}=...=u^2{}_{w\left(k-\mathrm{one}\right),k}=u^2{}_{w\mathrm{one}}=u^2{}_{w2}=u^2{}_{w3}...=u^2{}_{wk}=u^2{}_w$$

If r _1,2 = r _1,3 = ... = r _2,3 = ... = r _{(k-1), k} = r _w , then the square of the total voltage of the output noise UAZP 32 according to the generalizing formula is equal to:

$$u^2{}_{w\Sigma }=k{u}^2{}_w+k\left(k-\mathrm{one}\right)u^2{}_w{r}_w$$

Let us denote the relationship between the signal voltage fluctuations in k branches: u _c1 and u _c2 , u _c1 and u _c3 , ... u _c1 and u _ck , u _c2 and u _c3 ... u _c2 and u _ck , u _c3 u _c4 ... u _c3 and u _ck , ... u _{c (k-1)} and u _ck , the cross-correlation or common mode coefficient r _c1,2 ... r _{c (k-1), k} , the range of which lies in the range from -1 to +1 and which is determined for help measurements.

Express $$\begin{array}{l}{u}_{c\mathrm{one}}\times u_{c2}=u^2{}_{\mathrm{<figure-callout\; id="1"\; label="c"\; filenames="00000081.png"\; state="\{\{state\}\}">c</figure-callout>}\mathrm{1,2}},u_{c\mathrm{one}}\times u_{c3}=u^2{}_{c1.3},...u_{c\mathrm{one}}\times u_{ck}=u^2{}_{c\mathrm{one,}k},u_{c2}\times u_{c3}=u^2{}_{c\mathrm{2,3}},\\ ...u_{c2}\times u_{ck}=u^2{}_{c2k},...u_{c(k-\mathrm{one})}\times u_{ck}=u^2{}_{c(k-\mathrm{one}),k},\end{array}$$

then the square of the effective voltage of the total oscillation of the signals at the output of the USSC 34 in general form is:

$$u^2{}_{c\Sigma }={\displaystyle \sum _{i=\mathrm{one}}^k{u}^2{}_{ci}}+2\times \left({\displaystyle \sum _{i=2}^k{u}^2{}_{c\mathrm{one,}i}\times r_{c\mathrm{one,}i}}+{\displaystyle \sum _{i=3}^k{u}^2{}_{c2i}\times r_{c2i}}+...+{\displaystyle \sum _{i=k-\mathrm{one}}^k{u}^2{}_{c\left(k-2\right),i}{r}_{c\left(k-2\right),i}}+u^2{}_{c\left(k-\mathrm{one}\right),k}{r}_{c\left(k-\mathrm{one}\right),k}\right)$$

where k is the number of branches of the URS 31, the number of ADCs included in UACP 32, the number of summed signals in USSC 34.

With a uniform division of the input signal power in the URS 31 by k signals of equal power, the use of the same type of ADC and summing devices in the USSC 34, we will allow:

$$u_{c\mathrm{one}}=u_{c2}=...=u_{c(k-\mathrm{one})}=u_{ck}=u_c,$$

$$u^2{}_{\mathrm{<figure-callout\; id="1"\; label="c"\; filenames="00000081.png"\; state="\{\{state\}\}">c</figure-callout>}\mathrm{1,2}}=u^2{}_{c1.3}...=u^2{}_{c\mathrm{one,}k}=u^2{}_{c\mathrm{2,3}}=...=u^2{}_{c2k}=...=u^2{}_{c\left(k-\mathrm{one}\right)k}=u^2{}_{c\mathrm{one}}=u^2{}_{c2}=...=u^2{}_{ck-\mathrm{one}}=u^2{}_{ck}=u^2{}_c,$$

We express the coefficients of mutual correlation (in phase) between the oscillations of the received signals through the angle of shift between the oscillations:

r _c1,2 = cos (ψ ₁ -ψ ₂ ); rc _1.3 = cos (ψ ₁ -ψ ₃ ); ... r _{c1, k} = cos (ψ ₁ -ψ _k );

r _c2,3 = cos (ψ ₂ -ψ ₃ ); ... r _{c2, k} = cos (ψ ₂ -ψ _k ); ... r _{c (k-1), k} = cos (ψ _k-1 -ψ _k ) ,

where ψ ₁ , ψ ₂ , ψ _3, ... ψ _k-1 , ψ _k are the initial phases of the summed oscillations of the signals, and ψ ₁ -ψ ₂ = Δψ _1,2 , ... ψ ₂ -ψ _k = Δψ _{2, k} , ... ψ _k-1 -ψ _k = Δψ _{(k-1), k} are the angles of phase shifts between the corresponding oscillations of the signals, which can be determined using measurements.

If r _с1,2 = r _c1,3 = ... = r _{с1, k} = r _с2,3 = ... = 1c _{2, k} = ... = r _{c (k-1), k} = r _s ,

then Δψ _1,2 = Δψ _1,3 = ... = Δψ _{1, k} = Δψ _2,3 = ... = Δψ _{2, k} = ... Δψ _{(k-1), k} = Δψ, then the square of the effective voltage of the total oscillation of the useful signal at the output of USSC 34, according to the generalizing formula, is equal to:

$$\begin{array}{l}{u}^2{}_{c\Sigma }=k{u}^2{}_c+k{u}^2{}_c\left(k-\mathrm{one}\right)r_c=k{u}^2{}_c+k{u}^2{}_c\left(k-\mathrm{one}\right)\cos \Delta \psi =k{u}^2{}_{\mathrm{from}\mathrm{BUT}\mathrm{Ts}P}+k{u}^2{}_{\mathrm{from}\mathrm{BUT}\mathrm{Ts}P}\left(k-\mathrm{one}\right)r_{\mathrm{from}}=\\ =k{u}^2{}_{\mathrm{from}\mathrm{BUT}\mathrm{Ts}P}\left[\mathrm{one}+\left(k-\mathrm{one}\right)r_c\right].\end{array}$$

Noise coefficient UAZP 32 is determined by the formula:

. $${\mathrm{TO}}_{w\mathrm{At}\mathrm{BUT}\mathrm{Ts}P}=\mathrm{one}+\frac{\left({\mathrm{TO}}_{w\mathrm{one}\mathrm{BUT}\mathrm{Ts}PR}-\mathrm{one}\right)k\left[\mathrm{one}+r_w\left(k-\mathrm{one}\right)\right]}{\left[\mathrm{one}+r_c\left(k-\mathrm{one}\right)\right]}$$

.

We find the degree of increase in the noise / signal ratio of the UACP 32 at the output of the BACP 30 compared to the ratio of the noise / signal at the output of a single ADC according to the formula:

, $${\delta }_{\left(W/\mathrm{FROM}\right)}=\frac{{\mathrm{TO}}_{w\mathrm{At}\mathrm{BUT}\mathrm{Ts}P}-\mathrm{one}}{{\mathrm{TO}}_{w\mathrm{one}\mathrm{BUT}\mathrm{Ts}PR}-\mathrm{one}}=\frac{k\left[\mathrm{one}+r_w\left(k-\mathrm{one}\right)\right]}{\mathrm{one}+r_c\left(k-\mathrm{one}\right)}$$

,

where δ _{(W / C)} is the coefficient of increase in the noise / signal ratio of the UACP 32 at the output of the BACC 30 compared to the ratio of the noise / signal at the output of a single ADC. Provided r _u> 0, r _<1:

, $${\delta }_{\left(W/\mathrm{FROM}\right)}\le \frac{\mathrm{one}+r_{w}k}{r_c}$$

,

to select the number k, it is necessary to set a permissible coefficient of increasing the noise / signal ratio at the output of the BACP 30, equal to δ _{(W / Sdop)} , and determine the number k by the formula:

. $$k\le \frac{{\delta }_{\left(W/\mathrm{FROM}\right)d\mathrm{about}P}{r}_c-\mathrm{one}}{r_w}$$

.

Thus, the number k of branches in BACP 30 is selected from the condition:

$$\frac{P_{c\mathrm{BUT}\text{A.}\mathrm{TO}\mathrm{and}}{K}_{p\Sigma }}{R_{\mathrm{from}\mathrm{one}\mathrm{BUT}\mathrm{Ts}P\mathrm{and}}}\le k\le \{\begin{array}{c}{K}_{w\mathrm{one}\mathrm{BUT}\mathrm{Ts}PR}PR\mathrm{and}{r}_w=0r_c=\mathrm{one,}\\ \frac{{\delta }_{\left(W/\mathrm{FROM}\right)d\mathrm{about}P}{r}_c-\mathrm{one}}{r_w}PR\mathrm{and}{r}_w>0r_c<\mathrm{one,}\end{array}$$

where R _cAKi is the maximum allowable input power of one of the two signals acting on the AK input, at which the third-order intermodulation interference arising from the interaction of these two signals in the AK before the input of a single ADC has an acceptable value,

K _p∑ - transmission coefficient of power AK to the input of a single ADC,

R _s1ATsPi - the maximum allowable power of one of the two signals acting on the input of a single ADC, in which the third-order intermodulation interference resulting from the interaction of these two signals within the integral ADC characteristic has an acceptable value,

To _w1ACPr - noise figure of a single ADC taking into account distortions and residuals of randomization noise,

δ _{(W / C) add} - allowable coefficient for increasing the noise / signal ratio at the output of the BACP 30 compared to the ratio of the noise / signal at the output of a single ADC,

r _c is the cross-correlation coefficient between the signals in BACP 30,

r _W - the coefficient of cross-correlation between the output noise UATSP 32.

In CACS 33, which is included in the BACP 30, using 2BUK 17 the level, shape and frequency of the signal with a sampling frequency are regulated in order to optimize the analog-to-digital conversion of signals to UACP 32, this signal is fed to the input of a signal splitter with a sampling frequency in UACP 32, s k outputs of which k signals with a sampling frequency through a block of 17 phase shifters controlled by 2BUK for a signal with a sampling frequency are fed to the input for a signal with a sampling frequency of each ADC, while the controlled phase shifters by commands and TSUKR 6 or UDDS 42 change the phase (ψ) oscillations sampling signal by an angle of 0 to 2π incrementally Δψ _p furthermore in SATSS 33 formed k noise or wobbling signals, each of which is synthesized simultaneously in digital and analog form, all synthesized signals are not correlated with each other. All k noise or wobble signals in analog and digital forms are input to UACP 32, where each of k analog signals is supplied to each of k analog adders included at the input of each of k ADCs, and each of k digital signals is supplied to each of k digital subtractors included at the output of each of k ADCs. In the adder, the signal is summed with the received signal, after analog-to-digital conversion, the same signal in digital form is subtracted from the total signal in the subtractor. Subtraction is performed to eliminate or attenuate the negative effect of these signals on the noise level and noise at the ADC output, in addition, noise or wobble signals can have a frequency band outside the frequency band of the received signal, and then their negative effect decreases to a vanishingly small value.

The digital signal from the output of the BACP 30 of each of the AK 8 ₁ ... AK 8 _{m is} supplied to the corresponding input of the first multiplexer / demultiplexer 9. Management and control of the controlled units included in each of the AK 8 ₁ ... AK 8 _m are carried out in automatic and automated modes. In automatic mode, when adapting control commands through 2BUK 17 come from the AK units. In automated mode, control commands are received from CCU 6 or UDDS 41 via 2BUK 17, and control response signals through 2BUK 17 in automatic mode are sent to AK, UDDS 41 or CCU 6 units, and in automated mode, control signals are sent to UDDS 41 or CCU 6 In this case, all control commands and control signals in the AK are generated using 2BUK 17. The control and control buses AK 8i ... AK 8щ are connected to the corresponding buses of the first multiplexer / demultiplexer 9, after multiplexing the generated signal is converted is transmitted in the first optoelectronic / electron-optical converter 10 into an optical signal, the optical signal is amplified by the first optical transceiver 11 and fed to the output of the MPU 2 ₁ ... MPU 2 _n and then through the FOCL 3 ₁ ... FOCL 3 _{n the} optical signal is transmitted over a long distance in BFB 4 ₁ ... BFB 4 _n , where after receiving in the second transceiver 35 and converting from an optical signal to an electric in the second optoelectronic / electron-optical converter 36 and demultiplexing in the second multiplexer / demultiplex An axis of 37 m signals is fed to the m inputs of the MK 38, which commutes m signals to the L inputs of the BCRPU 39, containing L CRPUs, in each of which the main selectivity is achieved in the necessary frequency band of each received signal and digital reception channels (CC) are created, while L is determined by the formula:

, где L - число ЦРПУ, создающих ЦК приема, $$L=m\frac{\Delta f_{\mathrm{BUT}\mathrm{TO}}}{\Delta F_{\mathrm{Ts}\mathrm{TO}}}$$

where L is the number of central distribution centers creating the reception CC,

Δf _AK - AK each PP, ΔF _CC - PP each CC, m - the number of AK.

From the output of the BCRPU 39, the signals are received simultaneously in the memory 40 and in the UDDS 41. In the memory 40, the signals are recorded and stored, in the UDDS 41 the signals are demodulated and decoded, which, after demodulation and decoding, enter the memory 40 and are sent to LAN 5 via LAN 5 UDCS 41 also demodulates and decodes the signals that are extracted from the memory 40. In addition, the CCAM 6 receives, upon request, demodulated and decoded signals extracted from the memory 40, which are transmitted through the UDCS 41 and LAN 5. In the case of control one UDDS 41, any of the BFB 4 ₁ ... BFB 4 _n , which is assigned as the manager, all signals are sent via LAN 5 to the control of the UDDS 41.

The choice of the direction of reception in space, the implementation of diversity reception of radio signals, the spatial selectivity of radio signals, the optimization of the structure and operating modes of each MPPU 2 ₁ ... MPPU 2 _n , AK 8 ₁ ... AK 8 _m , BOS 4 ₁ ... BOS 4 _n in the proposed PRC are made in automated mode with the participation of the operator in CCAM 6 or in UDDS 41 of any of the BFB 4 ₁ ... BFB 4 _n by appropriate control. In addition, the optimization of the structure and operating modes of AK 8 ₁ ... AK 8 _m , MK 38, BTsRPU 39, UDDS 41 can be performed in automatic mode, without operator intervention, which is carried out using 2BUK 17 and AK 8 ₁ ... AK units and devices 8 _m , MK 38, BTsRPU 39, UDDS 41, in which the corresponding signal measuring elements, elements of automatic adjustments and switching are installed in order to change transmission coefficients, bandwidths and structure of AK 8 ₁ ... AK 8 _m , as well as to change the values of the bands transmitting digital channels BTsRPU 39, mode o demodulation and decoding in UDDS 41. Automatic optimization of the structure and operating modes of the receiving paths is carried out in accordance with the EMO at the receiving location.

Information signals after processing from the outputs of the BFB 4 ₁ ... BFB 4 _n via LAN 5 are sent to the CCAM 6 or to the control UDDS 41 and then to peripheral devices: monitors, audio devices, printers, ROMs, etc.

Consider the possible implementation of the proposed PRC.

In AC 1, directional antennas such as BS 2, ZBS 2, BSh 2, ABV, BSVN, BSVN 2, BE 2, log-periodic, rhombic and other directional high-performance antennas, as well as loop antennas, can be used as antennas 1 ₁ ... 1 _n antennas such as VGRD, VGDSH, horn antennas, specular parabolic, vibrator, slotted and other antennas necessary for a given range of radio frequencies. All antennas used must be with horizontal, vertical and mixed polarizations, taking into account the requirements.

1BUK 12, UBZ 13, UBU 14, USU 15, included in the control unit 7 of each of the MPPU 2 ₁ ... MPPU 2 _n , can be performed both on the domestic and imported components, for example: 1BUK 12 can be executed on FPGA ( FPGA), such as EP2C 50 or Cyclone I, Stratix II GX from Altera. UBZ 13 may contain diodes of type 2A537A or other high-power pin diodes or high-power transistors, relays REV-18, REV-20, ARA200A05 made by Nais or other high-frequency relays, comparators 521 SAZ. UBU 14 can be performed on discrete transistors with large DD or amplifiers type ARJ109 firm "Teledyne Congar". USU 15 can be performed on managed splitters, Mini-Circuites switches and other similar elements.

Filter blocks 1BPPF 16, 2BPPF 19, 1BPPCH 25, 2BPPCH 27 can be performed on L-, C-elements with switching elements. 2BUK 17 can be performed on FPGAs such as EP2C 50 or Cyclone I, Stratix II GX from Altera and others.

UURCH 18, UUPCH 26 can be performed on discrete transistors with large DD or on integrated amplifiers with large DD from Teledyne Congar, for example, type ARJ109, as well as amplifiers from Mini-Circuites and others.

USCH 24 can be performed using FPGA EP2C 50, DAC AD9726, ADC AD9446, AD7725 and others. SM 23 can be performed on discrete diodes, transistors on the corresponding microcircuit assemblies, on ready-made mixers from Mini-Circuites and others.

1UA 20,1UK 21, 2UK 28, 2UA 29 can be performed using non-inductive resistors and switching diodes or ARA200A05 relays from Nais, controlled attenuators and switches of Mini-Circuites and others.

URS 31, UACP 32, CACS 33, USSC 34, which are part of the BACP 30, respectively, can be made using: controlled splitters, phase shifters from Mini-Circuites and others, ADCs AD7725, AD9446, DAC AD9726 from Analog Devices , ADC ASD5500 of Texas Instruments and others, FPGAs of Cyclone II and EP2C 20-50 of Altera, XC4VFX100-10FF1571, XC9572XL-10VQ441 of Xilinx and others.

Multiplexers / demultiplexers 9 and 37, optoelectronic / electron-optical converters 10 and 36, optical transceivers 11 and 35 contained in each of the MPU 2 ₁ ... MPU 2 _n and BOS 4 ₁ ... BOS 4 _n , can be performed on the HEBR- microcircuits 5921 from Avago, FPGA EP2C 50 from Altera, ADSP-21065L processors, AD9761 DAC and AD9201 ADC from Analog Devices and others.

As FOCL 3 ₁ ... FOCL 3 _n , fiber optic cables of the ST / PC-ST / PC-SS-3 type and others can be used.

CCAM 6, BOS 4 ₁ ... BOS 4 _n and their constituent MK 38, BTsRPU 39, UDDS 41 can be performed using universal computers, using processors such as TMS 320C6414 from Texas Instruments, ADSP-TS001 TigerSHARC from Analog Devices "And others, FPGAs of Cyclone II and EP2C 20-50 from Altera, XC4VFX100-10FF1571, XC9572XL-10VQ441 from Xilinx and others, AD9726, AD9761, AD9201 ADCs from Analog Devices and others.

The memory 40 can be performed using hard magnetic disks (HDD), such as: IBM 81Y9774; Iomega 34786; Iomega 35448; Dell 400-23135 and others. In addition, in CCAM 6 and BOC 4 ₁ ... BF 4 _n various audio, video and other peripheral devices should be used.

LAN 5 can be of the type “Ethernet”, “Fast Ethernet” or another type.

Claims (5)

1. The receiving radio center (PRC), containing at least one antenna, one multi-channel radio receiving device (MRPU), one matrix switchboard MK, one block of digital radio receiving devices (BCRPU), one demodulation and decoding device (UDS), while MRPU includes a matching device (SU), analog channels (AK), and the outputs of the SU are connected to the inputs of the corresponding AK, each of which contains at least one band-pass filter (PF), one controlled attenuator (UA), a frequency converter ( PrCh), one analog-digital a converter (ADC), the PF input is an AK input for connecting to the corresponding output of the control system, the frequency converter contains a mixer (SM), a local oscillator signal generating unit (BPSG), at least one intermediate-frequency filter (PPC), while the input of the frequency converter is the first SM input, the second input of which is connected to the BFSG output, the SM output is connected to the PLL input, the output of which is the RF output, and the ADC output is the AK output, and the corresponding MK outputs are connected to the corresponding inputs of the BCRPU, the input / output bus of which is connected to the corresponding UDDS output / input bus, characterized in that it contains an antenna system (AS) of n directional antennas, the outputs of which are directly connected to the inputs of the corresponding n MRPUs, equipped with optical inputs / outputs, which, through n bi-directional fiber-optic communication lines (FOCLs) introduced ) are connected to n optical outputs / inputs of n input signal processing units (BOS), other electrical inputs / outputs of which are connected to n outputs / inputs of a local area network (LAN), another input / output to the second is connected to the output / input of the entered control center of the channels of the radio reception (CCAM), and n is selected from the condition:
$$\frac{\mathrm{four}\pi }{\Delta \varphi }\le n\le p\frac{\mathrm{four}{\pi }^2}{\Delta \varphi \Delta \theta }$$

,
where Δφ is the width of the radiation pattern (MD) of the antennas in the azimuthal plane in radians (rad),
Δθ is the antenna beam width in the elevation plane in rad,
p is a coefficient that takes into account the number of polarizations of the radio waves received at the respective antennas, with each MPU containing an input device (VU), the input of which is the input of the MPU, am the outputs of the VU are m outputs of the said VN included in the VU, made by a controlled (USU), input connected to the output of the managed amplification unit (UBE) introduced into the VU, the input of which is connected to the output of the managed protection unit (UBZ) entered into the VU, the input of which is the VU input, with its inputs / outputs of control and monitoring UBU, UBZ and U The control systems are connected with the corresponding outputs / inputs of the first control and monitoring unit (1BUK) introduced into the control unit, the signal outputs and input / output control and monitoring buses of all m AK are connected to the corresponding m inputs and m output / input buses of the first multiplexer / demultiplexer introduced into the MPPU , another output / input bus connected to the input / output bus of the control unit, which is the corresponding input / output bus 1BUK, and the input / output of the first multiplexer / demultiplexer is connected to the output / input of the first optoelectronic / electron-optical converter, the other optical output / input of which is connected to the input / output of the introduced first optical transceiver, the other input / output of which is the input / output of the MCI for connecting the biofeedback, and m is selected from the condition:
$$\frac{\Delta f}{\Delta F_{\max }}\le m\le \mathrm{TO}s\frac{\Delta f}{\Delta F_{\min }}$$

,
where Δf is the range of working radio frequencies,
ΔF _min - the minimum width of the PP AK,
ΔF _max - the maximum width of the PP AK,
Kz - load factor of the bottom of the given antenna, to which this MRPU is connected, by radio signals from sources of radio emissions (IRI),
at the same time, all m AKs of all n MPPUs are performed identically and each AK contains serially connected the first tunable bandpass filter unit (1BPPF), the introduced controlled radio frequency amplifier (UURCH), the introduced second tunable bandpass filter unit (2BPPF), and the first controlled attenuator (1УА) , introduced the first managed switch (1UK), the above-mentioned frequency converter, introduced the controlled IF amplifier (UPC), the second IF filter block (2BFPCH), the second managed switch (2UK), the second controlled attenuator (2UA) and the analog-to-qi unit A single conversion (BACP), the output of which is the signal output of the AK, and the input / output bus of the control and control of the AK is the corresponding input / output bus of the introduced second control and control unit (2BUK), the other corresponding inputs and outputs of which are connected to the corresponding the outputs / inputs of control and monitoring of all units included in the AK, and the second input 2UK is connected to the second output 1UK, while the BACP included in the AK contains a controlled signal splitter (URS), the input of which is I have a BACP input for connecting the 2UA output, an analog-to-digital converter unit (UACP), an analog and digital signal synthesizer (CACS), and a controlled digital signal adder (CSCS), and k URS outputs are connected to the corresponding k inputs of the UACP, where k is the number of ADCs included in the UACP, the other corresponding inputs of which are connected to the corresponding outputs of the CACS, and the corresponding k outputs of the UACC are connected to the corresponding k inputs of the ACAC, the output of which is the BACC output for connection to the first multiplexer / demultiplexer , Where k is chosen from the condition:
$$\frac{P_{c\mathrm{BUT}\text{A.}\mathrm{TO}\mathrm{and}}{K}_{p\Sigma }}{R_{\mathrm{from}\mathrm{one}\mathrm{BUT}\mathrm{Ts}P\mathrm{and}}}\le k\le \{\begin{array}{c}{K}_{w\mathrm{one}\mathrm{BUT}\mathrm{Ts}PR}PR\mathrm{and}{r}_w=0r_c=\mathrm{one,}\\ \frac{{\delta }_{\left(W/\mathrm{FROM}\right)d\mathrm{about}P}{r}_c-\mathrm{one}}{r_w}PR\mathrm{and}{r}_w>0r_c<\mathrm{one.}\end{array}$$

where R _sAKi is the maximum allowable input power of one of the two signals acting on the input of the AC, in which the third-order intermodulation interference resulting from the interaction of these two signals in the AC before the input of the ADC has an acceptable value,
K _p∑ - transmission coefficient of power AK to the input of a single ADC,
R _s1ATsPi - the maximum allowable power of one of the two signals acting on the input of a single ADC, in which the third-order intermodulation interference resulting from the interaction of these two signals within the integral ADC characteristic has an acceptable value,
To _w1ACPr - noise figure of a single ADC taking into account distortions and residuals of randomization noise,
δ _{(W / C) add} - allowable coefficient for increasing the noise / signal ratio at the output of the BACC compared to the ratio of the noise / signal at the output of a single ADC,
r _c is the cross-correlation coefficient between the signals in the BAC,
r _W - the coefficient of cross-correlation between the output noise UATSP,
and each biofeedback contains a second optical transceiver connected in series with its inputs / outputs, the input / output of which is the input / output of the biofeedback, a second optoelectronic / electron-optical converter and a second multiplexer / demultiplexer, corresponding input / output bus, m outputs and m input / output buses connected to the corresponding output / input bus, m inputs and m other output / input buses of the above-mentioned MK, and its other corresponding input / output bus is connected to the corresponding the output / input bus of the UDDS, the other corresponding input / output bus of which and the output bus of the aforementioned BCRPU are connected to the corresponding output / input and input buses of the storage device (RAM) inserted into the BOS, the input / output of the UDDS is the input / output of the BOS the outputs of L, which MK is connected to the L inputs of the BCRPU, choose from the conditions:
$$L=m\frac{\Delta f_{\mathrm{BUT}\mathrm{TO}}}{\Delta F_{\mathrm{Ts}\mathrm{TO}}}$$

,
where L is the number of CRPU creating digital channels (CC) reception,
Δf _AK - the width of the PP of each AK,
ΔF _CC - the width of the PP of each Central Committee,
m is the number of AK. 2. PRC according to claim 1, characterized in that the frequency converter contains, as a local oscillator signal generating unit (BPSH), a controlled frequency synthesizer (USCH), with an output connected to the corresponding input of the above-mentioned SM, the output of which is connected to the input of the first IF filter block (1BFPC) whose output is the output of the inverter, and the input of the inverter is the input of SM. 3. PRC according to claim 1, characterized in that the URS included in the BACP contains a block of controlled splitters of the received radio signal and a block of controlled analog phase shifters, while the input of the URS is the input of the block of controlled splitters of the received radio signal, k outputs of which are connected to the corresponding inputs of the block controlled analog phase shifters, k outputs of which are URS outputs, while the input / output control and monitoring buses of the block of controlled splitters of the received radio signal and the block of controlled analogs log phase shifters through the input / output of control and monitoring of the URS are connected to the corresponding output / input 2BUK. 4. PRC according to claim 1, characterized in that the UAOC included in the BACC contains a signal splitter with a sampling frequency, k outputs of which are connected to the corresponding k inputs of a block of controlled phase shifters for a signal with a sampling frequency, each of k corresponding outputs of which is connected to each of k corresponding input for a signal with a sampling frequency of each of k ADCs, the other inputs of which are connected to the outputs of the corresponding of k analog adders, one input of each of which is the corresponding of k inputs of ADC To connect the outputs of the URS, and the other input of each of the k analog adders is each of the k inputs of the ADC to connect to the corresponding of the k outputs of the analog signals of the CACS, and the output of each of the k ADCs is connected to the input of each of the k corresponding digital subtractors, the other input of each of k digital subtractors, being each of the k inputs of the UACP, is connected to the corresponding of k outputs of digital signals of the CACS, and k outputs of the digital subtractors are k outputs of the UACP to connect to the k inputs of the controlled adder of the digital signal (USTSS), the input / output management and control bus coupler signal with a sampling rate controlled phase shifters and the block signal with a sampling frequency by input / UATSP management and control outputs connected to respective output / input 2BUK. 5. PRTS according to claim 1, characterized in that the USSC, included in the BACS, contains a block of controlled digital phase shifters, k inputs of which are k inputs of the USSC, and its k outputs are connected to the corresponding k inputs of the block of controlled digital adders, the output of which is the output USTSS and BACP for connecting to the input of the first multiplexer / demultiplexer, while the input / output control and monitoring buses of the block of controlled digital phase shifters and the block of controlled digital adders through the input / output of control and monitoring connected to the corresponding output / input 2BUK.

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