RU2476989C1 - Method of generating, measuring parameters and processing signals for double-frequency heading-glide path landing system of aircraft - Google Patents

Abstract

FIELD: radio engineering, communication.

SUBSTANCE: operating frequency of direct digital synthesis (DDS) microchips is programmed, modulation signals are generated in digital form based on reduced dependencies on narrow channel signal modulation, wide channel signal modulation, narrow channel carrier frequency and frequency offset of the narrow and wide channels. Further, analogue-to-digital conversion of the obtained signal is performed; the digital signal is filtered; narrow and wide channel signals are separated by shifting channel frequency to "zero" frequency by multiplying with a quadrature signal of the corresponding frequency; the obtained quadrature signals are filtered and demixed. Signal parameters are then measured and gain constants and modulation factors of each transmitter channel are corrected based on measurement results and transmitter output signals are equalised according to phase.

EFFECT: simple realisation of amplitude-phase distribution for different antenna arrays and enabling correction of amplitude-phase distribution when tuning a radio beacon on-site.

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Description

The invention relates to communication technology, and in particular to radio equipment, and can be used in systems and means of air traffic control.

Course-glide path system (CGS) (Eng. ILS from Instrument Landing System) is the most common radio navigation instrument approach approach system in aviation.

KGS consists of two beacons: heading beacon (CRM) and glide path beacon (GRM).

The KRM antenna system is a multi-element antenna array consisting of a linear row of directional antennas of a meter frequency range with horizontal polarization. To expand the working sector of the beacon to angles of ± 35 °, an additional antenna array is often used.

The range of working frequencies of the CRM is 108-112 MHz (a 40-channel frequency grid is used, where each CRM frequency is associated with a certain timing frequency).

KPM is placed outside the runway at the continuation of its center line. Its antenna system forms in space simultaneously two horizontal radiation patterns. The first diagram has one wide lobe directed along the center line, in which the carrier frequency is modulated in amplitude by the signal of the sum of frequencies of 90 and 150 Hz. The second diagram has two narrow antiphase lobes on the left and right side of the center line, in which the radio frequency is modulated in amplitude by a frequency difference signal of 90 and 150 Hz, and the carrier is suppressed.

As a result of addition, the signal is distributed in space in such a way that, when flying along the center line, the modulation depth of the signals of 90 and 150 Hz is the same, which means that the difference in modulation depths (RGM) is zero. With a deviation from the centerline, the modulation depth of the signal of one frequency increases and the other decreases, therefore, the RGM increases in the positive or negative direction. At the same time, the sum of the modulation depths (CGM) in the beacon coverage area is maintained at a constant level. On-board flight and navigation equipment measures the value of the RGM, determining the side and angle of deviation of the aircraft from the landing course.

The timing antenna system is in the simplest case a grating of two horizontally polarized decimetric antennas spaced apart in height (grating "0"). The operating frequency range of the timing is 329-335 MHz. The timing is placed from the side opposite to the building site and taxiways, at a distance of 120-180 m from the axis of the runway opposite the landing zone. The timing distance from the runway threshold is determined so that for a given glide path angle, the reference point (the point above the runway end through which the rectilinear part of the glide path) passes is at an altitude of 15 ± 3 m for radio beacon landing systems of categories I and II and 15 + 3- 1 m for category III systems.

The directional diagram of the timing antenna system is formed as a result of the reflection of radio waves from the earth's surface, therefore, special requirements are imposed on the cleanliness of the area immediately adjacent to the timing antenna system. To reduce the influence of irregularities of the underlying surface on the radiation pattern, and consequently, the curvature of the glide path, an antenna array of three vertically spaced antennas is used ("M" array). It provides reduced radiation power at small angles to the horizon. The timing uses the same operating principle as the timing. Its antenna system forms simultaneously in space two vertical radiation patterns, with one wide lobe and with two narrow ones - above and below the glide path plane (plane of the zero value of the RGM), normalized for each category of the landing system.

In foreign and domestic landing systems, passive combiners are used to form the amplitude-phase distribution (AFR) of the two-frequency antenna system of the directional and glide path beacons. As a rule, these devices perform the summation of four signals, which are generated by four transmitters:

- carrier + side frequencies of the narrow channel,

- side frequencies of a narrow channel,

- carrier + side frequencies of the wide channel,

- lateral frequencies of a wide channel,

and distribution in certain amplitude and phase ratios to the antenna emitters.

, N - число столбцов ФАР, а

, М - число строк ФАР), амплитуды токов возбуждения которых снизились, и координаты неисправных р-разрядных фазовращателей, номера отказавших в них переключающих элементов и виды отказов (обрывы или короткие замыкания), а также снижение амплитуд токов возбуждения поврежденных излучателей ФАР и осуществляется расчет нормированного значения уровня сигнала, излучаемого ФАР в заданном направлении, для 2Р случаев одновременного изменения состояний всех ее р-разрядных фазовращателей на величину дискрета их переключения Δφ с учетом данных контроля о фазе тока возбуждения каждого из m, n излучателей ФАР - ⌀изл (m, n), после чего фазовращатели ФАР устанавливаются в состояния, соответствующие максимальному из 2Р рассчитанных значений уровня излучаемого сигнала, отличающийся тем, что при контроле амплитудно-фазовых характеристик i-x излучателей ФАР определяются значения величин снижения амплитуд токов их возбуждения ΔAi, на основании которых определяются коды команд управления амплитудами токов возбуждения i-x излучателей ФАР Aупрi, осуществляемого с помощью соединенных с ними активных каналов управления, подключаемых между i-ми фазовращателями и i-ми излучателями ФАР при снижении амплитуд их токов возбуждения на величину, превышающую ΔA/2 (ΔА - дискрет управления амплитудой тока возбуждения i-го излучателя с помощью подключаемого к нему активного канала управления), а при расчете нормированного значения уровня сигнала, излучаемого ФАР в заданном направлении, наряду с фазами токов возбуждения излучателей ⌀изл (m, n) учитываются величины компенсации снижения амплитуд i-x излучателей Aупрi=Ent[(ΔAi/ΔA)+0,5]ΔА (см. патент РФ №2333578, МПК H01Q 3/26, опубл. 10.09.2008 г.).The prior art method for controlling the amplitude-phase distribution at the aperture of a phased array antenna (PAR), which is the closest analogue of the invention. In this known method, p-bit phase shifters of the phased array are set to states corresponding to the required position of the main maximum of the phased array pattern, phase quantization errors are decoded at the phased array by introducing a random or nonlinear distribution of the initial phase of the emitter excitation currents with its subsequent compensation when generating phase shifter control codes , control of the amplitude-phase characteristics of the HEADLIGHTS emitters, as a result of which the ix coordinates are determined emitters m, n (where

, N is the number of PAR columns, and

, M is the number of rows of the PAR), the amplitudes of the excitation currents of which have decreased, and the coordinates of the faulty p-bit phase shifters, the numbers of the switching elements that failed in them and the types of failures (breaks or short circuits), as well as the decrease in the amplitudes of the excitation currents of the damaged radiators of the PAR and the normalized value of the signal level emitted by the PAR in a given direction, for 2P cases of simultaneous changes in the states of all its p-bit phase shifters by the value of their switching discrete Δφ taking into account the data the role of the phase of the excitation current of each of the m, n HEADLIGHTER emitters is Isl (m, n), after which the HEADLOCK phase shifters are set to states corresponding to the maximum of the 2P calculated values of the emitted signal level, characterized in that when monitoring the amplitude-phase characteristics ix of the HEADLIGHTER emitters, the values of decreasing the amplitudes of the currents of their excitation ΔAi are determined, on the basis of which the codes of the commands for controlling the amplitudes of the excitation currents ix of the HEADLIGHTER radiators Aupr, which are carried out using active control channels connected between the i-th phase shifters and the i-th HEADLIGHTER emitters when the amplitudes of their excitation currents decrease by an amount exceeding ΔA / 2 (ΔA is the discrete control of the amplitude of the excitation current of the i-th emitter using the active control channel connected to it), and when calculating the normalized value of the signal level emitted by the PAR in a given direction, along with the phases of the excitation currents of the emitters izl (m, n), the compensation values for the decrease in the amplitudes ix of the emitters Aupi = Ent [(ΔAi / ΔA) +0.5] ΔА ( cm. RF patent No. 2333578, IPC H01Q 3/26, publ. September 10, 2008).

In the method defined as the closest analogue of the invention, adders-dividers are used, which are a combination of directional couplers, attenuators and phase-shifting elements. The disadvantages of devices that use combiners-dividers should include:

- the great complexity of manufacturing,

- for each PRA a unique adder divider is required,

- the inability to adjust the amplitude and phase distribution in the process of complex adjustment of the beacon,

- the need to adjust the phase to a specific frequency channel, i.e. the equipment should be regulated for the specific location of the beacon,

- devices are not redundant, which reduces the reliability of the system.

An object of the present invention is to remedy the above disadvantages.

The stated technical problem is solved by the proposed method for generating, measuring parameters and processing signal signals for a two-frequency course-glide path landing system for an aircraft, the method comprising the steps of:

- program the operating frequency of direct digital synthesis chips (DDS);

- form in digital form modulation signals according to the formula:

Where

a (t) is the modulation of the narrow channel signal,

b (t) is the modulation of the wide channel signal,

ω is the carrier frequency of the narrow channel,

Δω is the frequency spacing of the narrow and wide channels,

- perform analog-to-digital conversion of the received signal;

- filter the digitized signal;

- divide the signals of the narrow and wide channels by transferring the channel frequency to the “zero” frequency by multiplying by the quadrature signal of the corresponding frequency;

- filter and decimate the obtained quadrature signals;

- measure the following parameters: signal levels of narrow and wide channels, modulation coefficients for narrow and wide channels with frequencies of 90 and 150 Hz, phase difference of the output signals of the transmitter;

- phase out the output signals of the transmitter;

- correct the gain and modulation coefficients of each channel of the transmitter according to the measurement results.

A method is proposed for generating a signal for a dual-frequency landing system, in which an active phased array antenna (PAR) is controlled, for each emitter of which a complete signal is generated from a separate transmitter channel. This signal contains the sum of the signals of the narrow and wide channels.

According to ICAO requirements, the operating frequency of the heading beacon is in the range (108,000 ... 111.975) MHz, and the glide path beacon (328.6 ... 335.4) MHz. The narrow and wide channels are spaced in frequency symmetrically with respect to the assigned beacon frequency. In the directional beacon they are spaced at (5 ... 14) kHz, and in the glide path at (4 ... 32) kHz. Therefore, each channel of the transmitter must generate a two-frequency signal.

Since the relative frequency spacing is relatively small (4 orders of magnitude less than the frequencies themselves), such a signal can be generated using direct digital synthesis (DDS).

The technical result of the proposed method is the ease of implementation of AFR for various lattices (performed by software methods).

Another technical result of the proposed method is the ability to adjust AFR when setting up the beacon at the place of operation, as well as providing the ability to tune to a given frequency channel using a software method.

In addition, another technical result of the proposed method is the lack of the need for the use of adders-dividers.

Technical results are achieved due to the proposed method for generating, measuring parameters and processing signal signals for a two-frequency course-glide path landing system for an aircraft, which consists in the fact that

- program the operating frequency of direct digital synthesis chips (DDS);

- form in digital form modulation signals according to the formula:

Where

a (t) is the modulation of the narrow channel signal,

b (t) is the modulation of the wide channel signal,

ω is the carrier frequency of the narrow channel,

Δω is the frequency spacing of the narrow and wide channels,

- perform analog-to-digital conversion of the received signal;

- filter the digitized signal;

- divide the signals of the narrow and wide channels by transferring the channel frequency to the “zero” frequency by multiplying by the quadrature signal of the corresponding frequency;

- filter and decimate the obtained quadrature signals;

- measure the following parameters: signal levels of narrow and wide channels, modulation coefficients for narrow and wide channels with frequencies of 90 and 150 Hz, phase difference of the output signals of the transmitter;

- phase out the output signals of the transmitter;

- correct the gain and modulation coefficients of each channel of the transmitter according to the measurement results.

The formation principle becomes clear if the sum of 2 signals with spaced frequencies is represented by a signal of one frequency, modulated in amplitude and phase:

Where

a (t) is the modulation of the narrow channel signal,

b (t) is the modulation of the wide channel signal,

ω is the carrier frequency of the narrow channel,

Δω is the frequency spacing of the narrow and wide channels,

In this case, a two-frequency signal can be obtained by tuning DDS, for example AD9910 from Analog Devises, to the narrow channel frequency and modulating the amplitude and phase in accordance with formula (1).

The DDS modulation method consists in representing a two-frequency signal as the sum of two quadrature components with a variable amplitude:

where I (t) = a (t) -b (t) × cosΔωt,

Q (t) = b (t) × sinΔωt.

In this case, a two-frequency signal can be obtained by tuning DDS, for example AD9957 by Analog Devises, to the frequency of the narrow channel and modulating the amplitude of the quadrature components in accordance with formula (2).

Consider the implementation of this method on the example of a 4-channel transmitter.

The processor programs the operating frequency of direct digital synthesis chips (DDS), the carrier level, and also loads the required modulation coefficients into the modulator. The modulator, which is a programmable logic circuit (FPGA), digitally generates DDS modulation signals according to the formula (1) or (2). Modulation signals are generated with a frequency of 5 MHz, i.e. the quantization frequency is 3 orders of magnitude higher than the separation frequency Δω. This ensures accurate reproduction of the dual-frequency signal at the DDS output. After filtering, the DDS signal is amplified to the required level by an amplifier with an adjustable gain.

Part of the power of the output signals through directional couplers (BUT) is fed to a 4-channel analog-to-digital converter (ADC).

Processing is carried out at a carrier frequency with a sampling frequency of 100 MHz. The output signals of the ADC are supplied to the FPGA, where circuitry is used to produce:

- digital filtering of input signals to prevent interference from non-working areas of Nyquist,

- separation of narrow and wide channel signals by transferring the channel frequency to the “zero” frequency by multiplying by a quadrature signal of the corresponding frequency,

- filtering and decimation of the obtained quadrature signals.

Further information processing is performed by the processor, which measures the following parameters:

- signal levels of narrow and wide channels,

- modulation factors for narrow and wide channels with frequencies of 90 and 150 Hz (operating modulation frequencies of ILS),

- phase difference of the output signals of the transmitter.

Based on the measurement results, the gain and modulation coefficients for each transmitter channel are adjusted. And also phase alignment of the transmitter output signals is performed.

If several such transmitters are necessary for the formation of an AFR, then the phase matching of the signals of different transmitters is achieved by using a single clock signal for all transmitters.

Claims (1)

A method for generating, measuring parameters and processing signal signals for a two-frequency course-glide path landing system of an aircraft, which consists in the fact that
- program the operating frequency of direct digital synthesis chips (DDS);
- form in digital form modulation signals according to the formula:

where a (t) is the modulation of the narrow channel signal,
b (t) is the modulation of the wide channel signal,
ω is the carrier frequency of the narrow channel,
Δω is the frequency spacing of the narrow and wide channels,

- perform analog-to-digital conversion of the received signal;
- filter the digitized signal;
- divide the signals of the narrow and wide channels by transferring the channel frequency to the “zero” frequency by multiplying by the quadrature signal of the corresponding frequency;
- filter and decimate the obtained quadrature signals;
- measure the following parameters: signal levels of narrow and wide channels, modulation coefficients for narrow and wide channels with frequencies of 90 and 150 Hz, phase difference of the output signals of the transmitter;
- phase out the output signals of the transmitter;
- correct the gain and modulation coefficients of each channel of the transmitter according to the measurement results.