RU2472261C1 - Dipole emitter - Google Patents

Abstract

FIELD: radio engineering.

SUBSTANCE: dipole emitter comprises two collinear identical cylindrical conductors (1) and (2), a feed coaxial cable (3) and a balancing device (4) with input (5) and two antiphase output arms (6) and (7). At the same time adjacent ends (8) and (9) of collinear conductors (1) and (2) are arranged in close proximity. The feed coaxial cable (3) is connected with an input arm (5) of the balancing device (4), antiphase output arms (6) and (7) of which are connected with remote ends (10) and (11) of collinear conductors (1) and (2).

EFFECT: improved manufacturability of assembly, mounting and building works.

12 dwg

Description

The proposed dipole emitter belongs to the field of antenna technology and can be used as an independent (freestanding) antenna, as well as the pathogen of director antennas, as well as the basic radiating element of turnstile and multi-element log-periodic antennas.

The relevance of improving these types of antennas is due to the ever-increasing requirements for antenna systems in the microwave range in terms of their compactness, manufacturability and ease of layout (connection to a microwave signal source and to each other). The devices currently being developed require small-sized emitters of linearly and circularly polarized radio waves, which could be easily interfaced with various types of baluns without unnecessary connecting coaxial, strip or microstrip transmission lines.

The implementation of dipole emitters, the power terminals of which are not adjacent (i.e., located in close proximity), but are located at a noticeable distance from each other, will expand the scope of use of such emitters and will allow to use a wide range of balun devices that are antiphase when excited. whose outputs do not have to be adjacent. In particular, space is freed up in the central part of the emitter, which in current dipole antennas is occupied by supply lines coming from a balancing device with many bends and turns, which complicates miniaturization and worsens the processability of antennas, not to mention an increase in reflection coefficient and dissipative losses due to expense of the leading lines.

A classic dipole emitter is known, containing two collinear identical cylindrical conductors, the adjacent ends of which are located in close proximity, described as far back as the late 19th and early 20th centuries, as well as in the numerous literature of recent decades, in particular in the work: Markov G.T., Sazonov D.M., "Antennas", Moscow: Energia, 1975, chapter 2. The emitter also contains a balancing device with input and two out-of-phase output arms, which are connected to adjacent ends of collinear conductors. The emitter is powered by a coaxial cable connected to the input arm of the balancing device. Structurally, both collinear conductors and a balancing device are made in the form of a complete assembly unit, which is an antenna. In this case, two options for the layout of the antenna are possible.

In the first embodiment, the balancing device should have antiphase outputs that are spaced apart from each other in space at a distance that is negligible compared to the length of the collinear conductors. Examples of such balancing devices are: a quarter-wave “cup”, quarter-wave slots in the braid of a supply coaxial cable at its end connected to conductors, a balancing prefix, a “dovetail” in a grounded foil layer for printing, which is widely used in television director antennas to balance a U-elbow and others.

In the second embodiment, the balancing device may have antiphase outputs spaced from each other in space at a distance equal to part of the wavelength of the emitted / received radio signal. In this case, the spatially separated outputs of the balancing device are connected to the adjacent ends of the collinear conductors by pieces of coaxial, strip or microstrip transmission lines. Examples of this group of balancing devices are directional couplers on connected lines, stub bridges, ring power dividers and several others.

Both versions of the design of classical dipole emitters with both groups of balancing devices are well known (see the aforementioned work "Antennas", sections 9-2, 9-3, 9-4, 13-5, Fig. 14-4) and are widely used in antenna technology.

However, this arrangement of the dipole emitter, when only the adjacent ends of the collinear radiating conductors are involved, leads to the need to place in the immediate vicinity of these conductors in the central region of the emitter either the upper part of the design of the balun device, or the supply lines coming from the balun device. At the same time, structural and layout difficulties arise, especially with a large number of dipoles, which impede the rational placement of emitters at the installation site (aircraft board, car chassis or tractor, etc.) and their connection with a coaxial cable to the signal source. For many decades, they have come to terms with these difficulties, although the search for overcoming them deserves all attention and is a worthy scientific and technical task.

Also known is a dipole emitter (a band shunt vibrator), described in the work: Eisenberg GZ, "VHF Antennas", M .: State Publishing House of Literature on Communications and Radio, 1957, p. 258. In this emitter, a signal is applied to the adjacent ends of two identical collinear cylindrical conductors mounted on a metal bracket at their midpoints. This embodiment of the emitter allows you to adjust its input impedance by changing the position of the mounting points.

However, the aforementioned emitter is powered by a symmetrical (balanced) two-wire line and cannot be powered by a coaxial cable without a balancing device. The latter again creates difficulties of a structural and layout nature that impede compact placement at the installation site.

Also known is a dipole emitter (delta transformer) formed by taps of a symmetrical two-wire line connected symmetrically to two points of a solid cylindrical conductor, described in the work: A. Dorokhov, “Calculation and design of antenna-feeder devices”, Kharkov, publishing house Kharkov University, 1960, p. 69, fig. 30.IIa. In this radiator, matching with the signal source is ensured by the choice of connection points for the diverging taps of the symmetric feeder.

However, this emitter cannot be powered by a coaxial cable without a balancing device, which also creates difficulties of a structural-layout nature.

The prototype of the invention is the first-mentioned dipole emitter described in the aforementioned work "Antennas", chapter 2. As already noted, the out-of-phase output arms of the balancing device are connected to the adjacent ends of two collinear identical cylindrical conductors, which leads to structural difficulties, some of which can be overcome (and even then not always) only by complicating the design of the balancing device or by sealing wiring suitable for the terminals of adjacent ends of the conductors of the connecting transmission lines. All these constructive solutions taken together represent an extensive way to solve the problem, which is characterized by a noticeable decrease in the manufacturability of installation and assembly works during the assembly of multi-element antennas and their placement at the installation site.

The task of the invention is the creation of a dipole emitter having a higher manufacturability of assembly, installation and layout works.

The solution to this problem is ensured by the fact that in the known dipole radiator containing two collinear identical cylindrical conductors, a coaxial cable and a balancing device with input and two out-of-phase output arms, while the adjacent ends of the collinear conductors are located in close proximity, the supply coaxial cable is connected to the input the shoulder of the balancing device, the output shoulders of which are connected to the said conductors, the output shoulders of the balancing device Twa connected to the distal ends of said conductors.

от относительной длины цилиндрических проводников, на фиг.4 изображены зависимости активных составляющих входного сопротивления излучающих цилиндрических проводников от их относительной длины, на фиг.5 показано расположение текущей точки интегрирования в ближней зоне, на фиг.6 представлены зависимости реактивных составляющих входного сопротивления излучающих цилиндрических проводников от их относительной длины, на фиг.7 изображен вариант реализации компенсирующих неизлучающих индуктивностей, на фиг.8 представлены расчетные и экспериментальные значения возвратных потерь (Return loss) и развязки опытного образца излучателя, на фиг.9 показаны расчетные и экспериментальные диаграммы направленности опытного образца в плоскости электрического вектора

для основной поляризации, на фиг.10 - те же величины для кросс-поляризации, на фиг.11 изображены расчетные и экспериментальные диаграммы направленности опытного образца в плоскости магнитного вектора

для основной поляризации, на фиг.12 - те же величины для кросс-поляризации.Figure 1 shows the proposed dipole emitter, figure 2 shows the distribution of current along its cylindrical conductors, figure 3 shows the dependence of the width of the radiation pattern in the plane of the electric vector

on the relative length of the cylindrical conductors, Fig. 4 shows the dependences of the active components of the input resistance of the radiating cylindrical conductors on their relative length, Fig. 5 shows the location of the current integration point in the near zone, Fig. 6 shows the dependences of the reactive components of the input resistance of the radiating cylindrical conductors from their relative lengths, Fig. 7 shows an embodiment of compensating non-radiating inductances, Fig. 8 shows the calculated and experimental imentalnye return loss value (Return loss) and decoupling the prototype radiator 9 shows the calculated and the experimental pattern of the test sample in the plane of the electric vector

for the main polarization, figure 10 - the same values for cross-polarization, figure 11 shows the calculated and experimental radiation patterns of the prototype in the plane of the magnetic vector

for the main polarization, in Fig.12 - the same values for cross-polarization.

The proposed dipole emitter (figure 1) contains two collinear identical cylindrical conductors 1 and 2, supplying a coaxial cable 3 and a balancing device 4 with input 5 and two out-of-phase output 6, 7 arms. In this case, the adjacent ends 8 and 9 of the collinear conductors 1 and 2 are located in close proximity. This means that the distance between the ends 8 and 9 does not exceed 0.01 · λ _c , where λ _c is the average wavelength of the working frequency range f _L ... f _{R of the} emitter:

;

;

The mentioned restriction corresponds to the classification of distances, gaps and diameters of the cylindrical conductors of the emitters adopted in the field of antennas and indicated in the aforementioned work "Antennas", chapter 2.

The coaxial supply cable 3 is connected to the input arm 5 of the balancing device 4, the out-of-phase output arms 6 and 7 of which are connected to the remote ends 10 and 11 of the collinear conductors 1 and 2. It is assumed that the dipole emitter is located in unlimited free space with relative dielectric and magnetic permeabilities

and the Cartesian coordinate system shown in Fig. 1 is connected with this emitter, so that the origin is on the axis of the conductors 1 and 2 in the center of the gap between their adjacent ends 8 and 9. The emitter is fixed in space by the corresponding mounting system (in Fig. 1, the elements fasteners are not shown conditionally). It can also be printed using microelectronics technology (vacuum deposition of copper on ceramics) or strip printed circuit boards (etching of copper foil from “blank” sections of initially foil blanks).

The principle of operation of the proposed dipole emitter is as follows.

Suppose that a harmonic signal with a frequency f _{s is} received from a microwave oscillator through a supplying coaxial cable 3 to input 5 of a balancing device 4

where φ _c is the initial phase of the signal. The supplied signal is divided between the out-of-phase outputs 6 and 7 of the balancing device 4 in a 1: 1 ratio, and the phase incursions φ ₆ and φ ₇ formed in the balancing device provide at the frequency f _{c the} phase-out of the output signals u ₆ (t) and u ₇ (t):

Under the influence of bipolar voltages (4) applied to the ends 10 and 11, electric currents appear on the conductive surface of conductors 1 and 2, which are distributed over their surface so that the electromagnetic field excited by them in the surrounding free space satisfies the Maxwell equations and boundary conditions on the surface of the conductors 1 and 2 (FIG. 1). In accordance with the generally accepted methodology for the analysis of any emitters, the internal problem is first solved, allowing to find the current distribution among the radiating elements, and then all the characteristics of the emitter are found in the process of solving the external problem (see the aforementioned work “Antennas”, page 50), including resistance radiation, input impedance, radiation pattern, etc.

In the process of solving the internal problem, conductors 1 and 2 are taken (Fig. 1), satisfying the "thin-cylinder" requirements and the condition of maximum proximity of the adjacent ends 8 and 9 of each of the conductors:

When these conditions are met, and also taking into account the axial symmetry of conductors 1 and 2, the following statements are valid (see the aforementioned work “Antennas”, p. 50, 51).

First, surface electric currents on conductors 1 and 2 are characterized only by a longitudinal component with a complex

. Торцевые токи проводников 1 и 2 на смежных концах 8 и 9 при этом игнорируются. По известной плотности тока определяется комплексная амплитуда продольного электрического тока

, который мыслится как бесконечно тонкая токовая нить, совпадающая с осью Z в пределах -l≤z≤l. В этих пределах ток

считается непрерывной функцией координаты Z и обращается в нуль на смежных концах 8 и 9. Если в соответствии с (5) пренебречь размером b, то должно соблюдаться условие:current density amplitude

. The end currents of conductors 1 and 2 at adjacent ends 8 and 9 are ignored. The known current density determines the complex amplitude of the longitudinal electric current

, which is thought of as an infinitely thin current thread, coinciding with the Z axis in the range of -l≤z≤l. Within these limits, the current

is considered a continuous function of the Z coordinate and vanishes at adjacent ends 8 and 9. If, in accordance with (5), size b is neglected, then the condition must be satisfied:

на боковой поверхности идеально проводящих проводников 1 и 2, охватывающих нить тока (т.е., при ρ=а), обращается в нуль:Secondly, the tangent component E _kac (z) of the electric field vector created by the current thread

on the side surface of perfectly conducting conductors 1 and 2, covering the current thread (i.e., at ρ = a ), it vanishes:

where ρ is the distance from the Z axis to the lateral surface of conductors 1 and 2.

создает на боковых поверхностях проводников 1 и 2 векторный потенциал только с продольной составляющей

. Эта составляющая определяет продольную составляющую вектора напряженности электрического поля E_кас(z), которая является одновременно составляющей, касательной к боковой поверхности цилиндрических проводников 1 и 2, в виде:The formulated statements allow us to give a mathematical formulation of the internal problem of the claimed dipole emitter, namely: unknown current distribution

creates on the side surfaces of conductors 1 and 2 a vector potential with only a longitudinal component

. This component determines the longitudinal component of the electric field vector E _cas (z), which is simultaneously a component tangent to the side surface of the cylindrical conductors 1 and 2, in the form:

where ε ₀ , μ ₀ are the electric and magnetic constants of vacuum, respectively:

- орт оси Z.ω is the circular frequency;

- unit of the Z axis.

(x, y, z) в произвольной точке P(x, y, z) окружающего пространства, определяемой по векторному потенциалу

электрического тока и векторному потенциалу

магнитного тока в элементах излучателя (см. вышеупомянутую работу «Антенны», стр.15):Formula (8) is derived from the general formula for the electric field strength

(x, y, z) at an arbitrary point P (x, y, z) of the surrounding space, determined by the vector potential

electric current and vector potential

magnetic current in the elements of the emitter (see the aforementioned work "Antennas", page 15):

.moreover, since b << l [conditions (5)], the contribution of the ring magnetic current in the gap between the adjacent ends 8 and 9 of conductors 1 and 2, whatever it may be, can be neglected. This corresponds to the fact that in the formula (10)

.

связан с плотностью электрического тока

в каждой точке Q(x', y', z'), принадлежащей излучающим элементам, соотношением, приведенном в вышеупомянутой работе «Антенны» на стр.17:In turn, the vector potential

associated with electric current density

at each point Q (x ', y', z ') belonging to the radiating elements, by the ratio given in the aforementioned work “Antennas” on page 17:

- волновое число окружающего свободного пространства;Where

- wave number of the surrounding free space;

- расстояние между точками наблюдения P(x, y, z) и интегрирования Q(x', y', z'); V' - объем пространства, занимаемого токами проводимости с плотностью

; интегрирование в (11) ведется только по «штрихованным» координатам x', y', z' в пределах объема V'.

- the distance between the observation points P (x, y, z) and the integration of Q (x ', y', z '); V '- the amount of space occupied by conduction currents with a density

; integration in (11) is carried out only along the “hatched” coordinates x ', y', z 'within the volume V'.

Given that according to the first statement in the formulation of the internal problem, the conduction current in the inventive emitter has only a longitudinal component

and also once again neglecting the distance 2b between adjacent ends 8 and 9 of the cylindrical conductors 1 and 2 (Fig. 1), from (11) we obtain:

электрического тока заявляемого дипольного излучателя имеет только проекцию на ось Z (иными словами: только продольную составляющую

, касательную к боковой поверхности цилиндрических проводников 1 и 2), то это позволяет, исходя из уравнений (7) и (8), получить интегро-дифференциальное уравнение относительно неизвестного пока еще закона изменения (распределения) «нитевидного» электрического тока

, текущего вдоль проводников 1 и 2 по их оси z' (совпадающей с осью z):Since the vector potential

electric current of the inventive dipole emitter has only a projection onto the Z axis (in other words: only the longitudinal component

tangent to the lateral surface of cylindrical conductors 1 and 2), this allows, based on equations (7) and (8), to obtain an integro-differential equation with respect to the yet unknown law of change (distribution) of the “filamentary” electric current

flowing along conductors 1 and 2 along their z axis (coinciding with the z axis):

After a series of transformations, the last equation is reduced to the form (it is advisable to return from the "hatched" z 'coordinate to the "unshaded" z based on the technique described in the work: Kocherzhevsky GN, “Antenna-feeder devices”, M .: Communication, 1972 p. 57):

- функционал тока вдоль излучателя; χ - малый параметр (параметр «тонкоцилиндровости» проводников 1 и 2):here C is an arbitrary constant;

- current functional along the emitter; χ is a small parameter (the "thin cylinder" parameter of conductors 1 and 2):

и уравнение (15) примет вид:If the radius of conductors 1 and 2 is small, χ tends to zero

and equation (15) takes the form:

The resulting differential equation (17) is a truncated version of a homogeneous linear differential equation of the nth order:

; x=z. При этом предполагается, что как первая, так и вторая производные тока I

(z) непрерывны на отрезке [-l≤z≤l]. Как известно из курса высшей математики, фундаментальная система решений общего уравнения (18) формируется из линейной комбинации n любых линейно-независимых частных решений. Упомянутую фундаментальную систему принято формировать по методу Эйлера, согласно которомуwhere n = 2; p ₁ (x) = 0; p ₂ (x) = k ² ; y = I

; x = z. It is assumed that both the first and second derivatives of the current I

(z) are continuous on the interval [-l≤z≤l]. As is known from the course of higher mathematics, the fundamental system of solutions to the general equation (18) is formed from a linear combination of n any linearly independent particular solutions. It is customary to form the aforementioned fundamental system according to the Euler method, according to which

that for n = 2 gives:

Next, the characteristic equation is solved:

having purely imaginary roots:

giving a general solution (20) of differential equation (17)

(z), текущего по оси проводников 1 и 2 (фиг.1), причем далее верхний индекс «э» в обозначениях тока и других величин с целью сокращения записи опускается:relative to the "filiform" conduction current I

(z), current along the axis of the conductors 1 and 2 (figure 1), and then the superscript "e" in the designations of the current and other quantities in order to reduce the record is omitted:

Here C ₁ and C ₂ are still arbitrary constants, which are concretized on the basis of the following boundary conditions imposed on the current distribution I _z (z):

a) at the adjacent ends of 8 and 9 of conductors 1 and 2 (Fig. 1), the conductivity current becomes equal to zero, which under the condition b << 1 is formulated as:

b) at the remote ends 10 and 11 of the conductors 1 and 2 connected to the antiphase outputs 6 and 7 of the balancing device 4 (figure 1), the current amplitude is I _l :

Thus, the internal problem with respect to the considered dipole emitter is solved, which allows taking into account (26) and (27) to write the expressions for the "filamentary" conduction current I _z (z) flowing along the axis of conductors 1 and 2 (Fig. 1) in form:

and also begin to solve the external problem. It should be noted that the conductivity current I _z (z) is distributed along the z axis according to a sinusoidal law with the amplitude of the sine wave (in other words: with the antinode of the current) I _m = I _l / sin (k · l), where I _l is the current amplitude at the remote ends 10 and 11 of conductors 1 and 2 [Fig. 1, see also condition (25)].

,

- есть радиус - вектор точки наблюдения. В соответствии с общепринятой методикой, совместим, согласно вышеупомянутой работы «Антенны», раздел 2.4, начало сферической системы координат (R, θ, φ) с началом декартовой системы (x, y, z), изображенном на фиг.1 посредине между смежными концами 8 и 9 проводников 1 и 2. Поскольку ток в заявляемом излучателе течет только в направлении оси z [см. формулу (28)], то векторный потенциал

в дальней зоне Фраунгофера будет иметь также только z - составляющую

, равную согласно (13):The solution of the external problem begins with obtaining the radiation pattern equation F _{E of the} inventive dipole emitter (Fig. 1), which characterizes the electromagnetic field at an arbitrary observation point P (x, y, z) located in the far Fraunhofer zone, where

,

- there is a radius - the vector of the observation point. In accordance with the generally accepted methodology, compatible, according to the aforementioned work of "Antennas", section 2.4, the beginning of the spherical coordinate system (R, θ, φ) with the beginning of the Cartesian system (x, y, z), shown in Fig. 1 in the middle between adjacent ends 8 and 9 of conductors 1 and 2. Since the current in the inventive emitter flows only in the direction of the z axis [see formula (28)], then the vector potential

in the far zone of Fraunhofer will also have only z - component

equal according to (13):

есть в данном случае разностный вектор между радиусом-вектором

(x, y, z) точки наблюдения P(x, y, z) и текущим радиусом-вектором

точки интегрирования Q(x', y', z'), перемещающейся по оси z' проводников 1 и 2 от точки z'=-l до точки z'=l. Согласно теореме косинусов и последующего разложения радикала в ряд Тейлора для дальней зоны получим (в дальней зоне используются только два элемента ряда):Where

in this case there is a difference vector between the radius vector

(x, y, z) of the observation point P (x, y, z) and the current radius vector

the integration point Q (x ', y', z ') moving along the z' axis of conductors 1 and 2 from the point z '= - l to the point z' = l. According to the theorem of cosines and the subsequent expansion of the radical in a Taylor series for the far zone, we obtain (in the far zone only two elements of the series are used):

where z'cosθ is the difference in the path of the rays drawn from the origin and from the current point of integration z 'to the observation point P (x, y, z), and the current point of integration z' is thought of as the middle of an infinitesimal portion dz 'of the split of conductors 1 and 2 (FIG. 2). This infinitesimal section is considered as a Hertz elementary electric dipole, the field structure of which is well known (see the aforementioned work "Antennas", section 1-3, pp. 24-26), which allows us to write the following equations for the section dz 'in a spherical coordinate system partitions (figure 2):

и осью z в положительном ее направлении,

- волновое (характеристическое) сопротивление окружающего безграничного пространства. В дальней зоне Фраунгофера θ₁≈θ; для знаменателей формул (31) и (32)

. Поэтому для суммарного электрического поля заявляемого излучателя в терминах соответствующих сферических проекций с учетом (28), (30)-(32), имеем:where θ ₁ is the angle between the difference vector

and the z axis in its positive direction,

- wave (characteristic) resistance of the surrounding infinite space. In the far Fraunhofer zone, θ ₁ ≈θ; for the denominators of formulas (31) and (32)

. Therefore, for the total electric field of the inventive emitter in terms of the corresponding spherical projections, taking into account (28), (30) - (32), we have:

The integrals I ₁ and I ₂ in (33) are calculated by double integration by parts [see the aforementioned work of "Antenna", p. 62], which gives the result:

электрического поля заявляемого излучателя в дальней зоне Фраунгофера выражение:Substituting (35) and (36) into (33), we obtain for tension

the electric field of the inventive emitter in the far zone of Fraunhofer expression:

Where

магнитного поля заявляемого излучателя запишется:Given (32), tension

the magnetic field of the inventive emitter is written:

магнитного поля. Свойство всенаправленности означает, что напряженность электрического поля

не зависит от угла φ, изменяющегося от 0° до 360° и отсчитываемого в плоскости XOY (фиг.1) при θ=π/2 от положительного направления оси x в сторону положительного направления оси y. Другими словами, угол φ не фигурирует в уравнениях (37)-(39). В то же время в плоскости поляризации (плоскости вектора

заявляемый излучатель обладает направленным свойством, что принято характеризовать нормированной диаграммой направленности F_E излучателя по полю [см. вышеупомянутую работу «Антенны», стр.21]. С учетом (37) получаем:The obtained equations indicate that the inventive dipole emitter, powered from a balancing device 4 at the remote ends 10 and 11 of the conductors 1 and 2 (Fig. 1), is a linearly polarized emitter [formula (37)] having an omnidirectional property in the vector plane

magnetic field. The omnidirectionality property means that the electric field

does not depend on the angle φ, varying from 0 ° to 360 ° and counted in the XOY plane (Fig. 1) at θ = π / 2 from the positive direction of the x axis towards the positive direction of the y axis. In other words, the angle φ does not appear in equations (37) - (39). At the same time, in the plane of polarization (the plane of the vector

the inventive emitter has a directional property, which is customary to characterize the normalized radiation pattern F _{E of the} emitter in the field [see the aforementioned work of "Antenna", p.21]. In view of (37), we obtain

An analysis of the radiation pattern (40) indicates that in the range of normalized lengths of conductors 1 and 2

по уровню половинной мощности (или по уровню

поля

) зависит от l/λ согласно фиг.3, позиция 12. При дальнейшем увеличении l/λ диаграмма направленности становится двухвершинной (т.е. раздваивается), а затем в ней появляются боковые лепестки, что объясняется появлением противофазных участков в распределении тока I_z(z) вдоль проводников 1 и 2 (фиг.1). Кроме того, при всех значениях l/λ излучение вдоль оси Z излучателя отсутствует, а вследствие осевой симметрии [угол φ отсутствует в формулах (37) и (39)] диаграмма направленности F_E в плоскости XOY равномерна и в полярной системе координат представляет собой окружность единичного радиуса. Существенно также, что фаза напряженности поля в дальней зоне Фраунгофера не зависит от углов наблюдения, и поэтому заявляемый дипольный излучатель имеет фазовый центр, совпадающий с началом координат [геометрическим центром излучающих проводников 1 и 2 (фиг.1)].the radiation maximum is oriented in the direction θ = π / 2 (XOY plane in Fig. 1). In this case, the side lobes in the radiation pattern are absent, and its width

by half power level (or by level

fields

) depends on l / λ according to Fig. 3, position 12. With a further increase in l / λ, the radiation pattern becomes two-vertex (i.e., bifurcates), and then side lobes appear in it, which is explained by the appearance of antiphase sections in the current distribution I _z (z) along conductors 1 and 2 (FIG. 1). In addition, for all values of l / λ, radiation along the Z axis of the emitter is absent, and due to axial symmetry [the angle φ is absent in formulas (37) and (39)] the radiation pattern F _E in the XOY plane is uniform and in the polar coordinate system it is a circle unit radius. It is also significant that the phase of the field intensity in the far Fraunhofer zone does not depend on the viewing angles, and therefore, the inventive dipole emitter has a phase center that coincides with the origin [geometric center of the radiating conductors 1 and 2 (figure 1)].

и

ортогональны радиальному орту

, то среднее за период значение вектора Пойнтинга

имеет только радиальную составляющую. На основании материалов работы: Никольский В.В. «Математический аппарат электродинамики», М.: МИРЭА, 1973, стр.79-81, величина

определяется с использованием векторных произведений как:The next step is to calculate the radiation resistance R _m , referred to the amplitude I _{m of the} current in the antinode, with the subsequent determination of the active component R _{in of the} input impedance of the emitter. In accordance with the aforementioned work of "Antennas", p. 64, the Poynting vector method is used for this purpose, which consists in integrating the power flux density determined by the radial component of the Poynting vector over the surface of a sphere located in the far zone in the center of which the claimed emitter . Since in the far zone both vectors

and

orthogonal to radial unit vector

, then the period average value of the Poynting vector

has only a radial component. Based on the materials of the work: Nikolsky V.V. "Mathematical apparatus of electrodynamics", Moscow: MIREA, 1973, pp. 79-81, value

defined using vector products as:

и

- мгновенные значения векторных гармонических функций напряженности электрического и магнитного полей;

- есть комплексная амплитуда комплексного представления

, характеризующего векторную функцию

, т.е.

- есть комплексно-сопряженная амплитуда комплексного представления

, определяющего векторную гармоническую функцию

, то есть

;

связан с ортами

и

как:

, при этом начало сферической системы координат находится по-прежнему посредине между смежными концами 8 и 9 проводников 1 и 2 (фиг.1); Re - оператор вычисления реальной части комплексного числа; ω_с=2πf_c. Тогда поток dP_Σ вектора

через дифференциально малую площадку dS=R²·sinθ·dθ·dφ сферы определяется скалярным произведением:where T _with - period of high-frequency oscillations at a frequency f _{c of the} input signal [see formula (3)],

and

- instantaneous values of vector harmonic functions of electric and magnetic fields;

- there is a complex amplitude of the complex representation

characterizing a vector function

, i.e.

- there is a complex conjugate amplitude of the complex representation

defining a vector harmonic function

, i.e

;

associated with orts

and

as:

while the beginning of the spherical coordinate system is still in the middle between the adjacent ends 8 and 9 of the conductors 1 and 2 (figure 1); Re is the operator of calculating the real part of the complex number; ω _c = 2πf _c . Then the flow dP _{Σ of the} vector

through a differentially small area dS = R ² · sinθ · dθ · dφ of the sphere is determined by the scalar product:

- направленный вектор элементарной площадки dS.Where

- directional vector of the elementary site dS.

As a result, the power P _Σ radiated by the claimed emitter is calculated as the surface integral over the closed surface S of a sphere of radius R:

:Performing the substitution, taking into account (37), (42) and (43), we successively obtain for free space, where

:

As a result, the radiation resistance R _m related to the current amplitude I _m in antinodes is found (see the definition of R _m in the aforementioned work “Antennas”, p. 65):

Despite the fact that under the current conditions of widespread use of computers, any integral can be calculated by numerical methods, the applicant received the following formula for radiation resistance R _m :

- интегральный косинус. График зависимости (47) представлен на фиг.4 сплошной линией (позиция 13). Осциллирующий характер зависимости (47) при l/λ>0.5 объясняется наличием вдоль проводников 1 и 2 (фиг.1) противофазных участков тока I_z(z).where γ = 0.5772 ... is the Euler constant;

- integral cosine. The dependence graph (47) is shown in Fig. 4 by a solid line (position 13). The oscillating nature of dependence (47) for l / λ> 0.5 is explained by the presence of antiphase current sections I _z (z) along conductors 1 and 2 (Fig. 1).

When determining the geometric dimensions of the inventive emitter in order to match the radiating conductors 1 and 2 with the characteristic (wave) resistance ρ _{0 of the} output arms 6 and 7 of the balancing device 4 (Fig. 1), the radiation resistance is not related to the antinode I _{m of} current in the conductors 1 and 2, and to the amplitude I _{l of the} current at their remote ends 10 and 11 (FIG. 2). Such radiation resistance referred to I _l is nothing more than the active component R _{in of the} complex input impedance Z _in = R _in + j · X _{in of the} radiating conductors 1 and 2. In accordance with the definition of R _in [see job: Balanis S.A. "Antenna theory. Analysis and Design ”, 3rd Edition, John Wiley & Sons, 2005, page 465], you can write:

The graphical dependence (48) of the active component R _{in of the} input impedance Z _{in of the} inventive dipole emitter is shown in Fig. 4 by a dashed line (position 14). It indicates the presence of areas of sharp (resonant) increase in input resistance R _in , which must be taken into account when designing the inventive emitter. In this case, both quantities, both R _m (k · l) and R _in (k · l) are independent of the radius a of the cylindrical conductors 1 and 2 (Fig. 1). This is because when considering the proposed dipole emitter from the far Fraunhofer zone, it is impossible to distinguish the z axis from the lateral surface of collinear cylindrical conductors 1 and 2 [in other words: the radius a of the conductors does not appear in formulas (46) - (48)].

The reactive component X _{in of the} complex input impedance Z _{in of the} radiating conductors 1 and 2 substantially depends on the aforementioned radius a. This component must be compensated for when choosing the geometric dimensions (tuning) of the inventive dipole emitter. In this case, it is necessary to choose a value of R _in such that it is consistent with the wave impedance ρ _{0 of the} supply coaxial cable 3 and the output arms 6 and 7 of the balancing device 4 (Fig. 1). Since the input impedance Z _in = R _in + j · X _{in is} included between the antiphase arms 6 and 7, the value of R _in should be 2 times greater than ρ ₀ :

In the case of a 50-ohm cable 3 (ρ ₀ = 50 Ohms) from the graph of R _in (figure 4, position 14) it follows that the value equal to:

As a result, at a given frequency f _s [see ratio (1)] the length of the collinear conductors 1 and 2 (figure 1) is calculated as:

определяется поверхностью цилиндра длиной 2·l и радиуса а, внутри которого расположены проводники 1 и 2. Сам вектор Пойнтинга

имеет две нормальные к поверхности цилиндра составляющие (радиальная составляющая P_r перпендикулярна боковой поверхности цилиндра, осевая составляющая Р_z перпендикулярна поверхностям его оснований), которые в цилиндрической системе координат (r, φ, z) имеют выражения:and then the reactive component X _in is compensated, which requires finding the dependence of X _in on the sizes l and a of the conductors 1 and 2. To obtain this dependence, the general methodology for calculating the radiation intensity in the near zone of the emitter is used, described in the aforementioned work “Antennas”, section 2.6 . According to it, to determine the power generated by conductors 1 and 2 (Fig. 1), it is necessary to take the product of the “filamentary" current I _z (z) by the longitudinal component E _z (z) of the electric field vector on the cylindrical surface of conductors 1 and 2 and integrate this the product along the length of the conductors is from -l to + l. In this case, the flow of the Poynting vector

is determined by the surface of a cylinder 2 · l long and of radius a , inside which conductors 1 and 2 are located. The Poynting vector itself

has two components normal to the cylinder surface (the radial component P _{r is} perpendicular to the side surface of the cylinder, the axial component P _{z is} perpendicular to the surfaces of its bases), which in the cylindrical coordinate system (r, φ, z) have the expressions:

where the asterisk (*) is the sign of complex conjugacy.

и вектор

в ближней зоне не находятся в фазе, то создаваемая в этой зоне проводниками 1 и 2 (фиг.1) мощность получается комплексной, т.е. имеет активную Re[P_CMPLX] и реактивную Im[P_CMPLX] составляющие. Если радиус проводников 1 и 2 мал (а/l<<1), то поверхностные интегралы по основаниям цилиндра (площадь основания π·a ²) будут пренебрежимо малы и создаваемая в ближней зоне мощность определиться интегрированием только по боковой цилиндрической поверхности:Since the vector

and vector

in the near zone are not in phase, then the power created in this zone by conductors 1 and 2 (Fig. 1) is complex, i.e. has active Re [P _CMPLX ] and reactive Im [P _CMPLX ] components. If the radius of conductors 1 and 2 is small ( a / l << 1), then the surface integrals over the cylinder bases (base area π · a ² ) will be negligible and the power generated in the near zone can be determined by integration only over the lateral cylindrical surface:

where, due to the symmetry of the current distribution I _z (z) relative to the origin (Fig. 2), integration is carried out along the half of the cylinder from z = 0 to z = l.

Substituting P _r from (52) and taking into account that the electromagnetic field does not depend on the coordinate φ, we obtain:

по замкнутому контуру окружности цилиндра 2π·а равна охватываемому контуром току проводимости

:According to Ampere’s law, vector circulation

along the closed loop circumference of the cylinder 2π · a is equal to the conduction current covered by the loop

:

therefore, the result is:

Similarly to the far zone, the integration result (56) can be represented as:

whence it follows that the quantity

can be thought of as a complex impedance related to the antinode I _{m of} current I _z (z), which, by analogy with (48), allows us to determine the active R _in and reactive X _in components of the input impedance Z _in = R _in + j · X _{in of the} claimed emitter:

. Другими словами, излучаемая активная мощность не зависит от формы окружающей излучатель поверхности интегрирования: то ли это цилиндр в ближней зоне, то ли сфера в дальней зоне. Поэтому величина R_m (а значит, и R_in), определяемая интегралом (58) при радиусе а, стремящемся к нулю, должна быть равна значению, определяемому интегралом (46). И это обстоятельство следует рассматривать как вариант взаимной проверки результатов интегрирования по (46) и (58), а также как фактор, способствующий повышению степени достоверности оценки также и реактивных составляющих Х_m и X_in [на сегодняшний день не известен альтернативный метод анализа, дающий уравнения для реактивных составляющих, чтобы сравнить их с (58) и (59)].It should be emphasized that the material part Re [P _CMPLX ] is equal to the active power (ie, energy per unit time) emitted by conductors 1 and 2 (Fig. 1) and leaving the radiator in accordance with its radiation pattern (40 ) in all directions to the surrounding free infinite space with characteristic resistance

. In other words, the radiated active power does not depend on the shape of the integration surface surrounding the emitter: either it is a cylinder in the near zone, or a sphere in the far zone. Therefore, the value R _m (and hence the R _in), defined by the integral (58) with radius a, tending to zero, must be equal to the value determined by the integral (46). And this circumstance should be considered as a variant of mutual verification of the integration results according to (46) and (58), and also as a factor contributing to an increase in the degree of reliability of the assessment of the reactive components X _m and X _{in as well} [an alternative analysis method is not known that gives equations for reactive components to compare them with (58) and (59)].

To continue the formation of the desired dependence of X _m on the sizes l and a of conductors 1 and 2 (Fig. 1), the general formulas given in the aforementioned work "Antennas" section 2.5 should be specified. First of all, we find the component E _z that appears in the complex impedance (58):

where A _z is the component of the vector potential at an arbitrary observation point M in the near field (Fig. 5, position 15):

Integration in (61) is performed from z '= 0 to z' = l, since the currents in both conductors 1 and 2 (Fig. 1) are equal at points symmetrically spaced from the origin and located at a distance from point M (Fig. 5):

Substituting (62) into (60) and taking into account the obvious equalities

;

;

we get:

Double integrating the last expression in parts, we find:

We consider the components of (65) separately. By direct substitution, we see that the first non-integral term in the lower limit z '= 0 is equal to zero:

and in the upper limit z '= l is defined as:

. Принимая во внимание ранее найденное распределение тока I_z(z')=±I_msin(k·z') [формула (28)], получаем выражение для второго слагаемого в (65):The second nonintegral term in (65) contains the first derivative

. Taking into account the previously found current distribution I _z (z ') = ± _{Im m} sin (k · z') [formula (28)], we obtain the expression for the second term in (65):

.Where

.

The third term in (65), which is a definite integral, disappears due to vanishing according to condition (17) of the first factor in square brackets.

At the final stage of the calculations, formula (58) is used, where (65), (67) and (68) should be substituted for r = a [ a is the radius of conductors 1 and 2 (Fig. 1)], as well as the current distribution I _z ( z) according to (28). Having performed the transformations taking into account k / ω · ε ₀ · ε _r = W = 120π [Ohm], we find that the components of the complex impedances of conductors 1 and 2 are equal to:

It is advisable to calculate the above integrals by a numerical method, for example, by numerical integration for equally spaced nodes using the Simpson rule described in: Korn G., Korn T., “A Handbook of Mathematics for Scientists and Engineers”, M .: Nauka, 1974, section 20.7-2, table 20.7-1, the second line is the Simpson rule. As a result, we can verify that formula (69) for the value of R _m with a sufficiently small radius a (for example, a = 0.0001 · l) gives the same values as formula (47). This indicates the adequacy of the approach used and provides the necessary degree of reliability for calculating the reactive components X _m and X _{in of the} input impedance [formulas (70) and (71)], the results of which are shown in Fig. 6 as solid (position 16, value X _m ) and dashed (position 17, value X _in ) by lines. The oscillating nature of the dependence X _m (kl) for l / λ> 0.5 is explained by the presence of antiphase current sections along conductors 1 and 2 (Fig. 1) (Fig. 2, Fig. 5). In addition, with the optimal value of l / λ [see condition (50)] there is a significant capacitive component of the input impedance, which must be compensated.

As a result, the required matching of the emitter with coaxial cables of standard wave impedance ρ ₀ = 50 or 75 Ohms is provided. The structure of the inventive dipole emitter, the distinguishing feature of which is the complete absence of any conductors or dielectrics near the emitting collinear cylindrical conductors 1 and 2 (Fig. 1), is such that, when the matching condition (49) is fulfilled, the unhindered possibility of using any balancing devices in the mode of traveling waves, and also increases the level of manufacturability of assembly, installation and layout works.

For experimental studies, according to the graphs of Figures 4 and 6, a dipole emitter was manufactured for operation at a frequency f _c = 880 MHz with a supply coaxial cable having a wave impedance ρ ₀ = 50 Ohms. As a balancing device 4 (FIG. 1), a classic 3-dB quarter-wave strip directional coupler was used. For its implementation, a FAF-4 sheet foil insulator (ε _r = 2.5) with a thickness of 2 mm was placed in a metal case of the corresponding standard size. The procedure for designing and manufacturing quarter-wave couplers is well known (for example, it is described in the work: edited by A. L. Feldstein, “A Guide to Elements of Strip Technology,” Moscow: Svyaz, 1979, section 3) and therefore is not given here.

As a result, the body of the balancing device 4, remote from the z axis of the radiating conductors 1 and 2 (Fig. 1) by the distance S _R , plays the role of a reflector and affects the formation of the radiation pattern as a whole of the emitter. It is advisable to choose the distance S _R so that the reactance X ₄ introduced by the metal body of the balancing device 4 is inductive and compensates for part of the capacitive reactance X _{in of} conductors 1 and 2. The remaining reactance X _in can be compensated by using the recommendation of the aforementioned work "Antennas" , p. 117, 3rd paragraph, according to which at the ends 10 and 11 of the collinear conductors 1 and 2 (Fig. 1), serial reactive resistances are switched on for compensation and adjustment Inductive nature, not related to the radiation process. The elimination of compensating series inductive reactances from the radiation process is ensured by the implementation of these inductances in the form of a short-circuited coaxial line, placed, according to the recommendations of the work: Kraus JD, "Antennas", Mc. Graw-Hill Book Co., Inc., N.-Y., Toronto, London, 1950, pp. 426-428, inside cylindrical conductors 1 and 2 (Fig. 7). Given the notation shown in Fig.7, it is possible to determine the input impedance Z * of a coaxial line of length l * as:

Thus, choosing (adjusting the emitter) the distance S _R (Fig. 1), the length l * (Fig. 7), the diameter D = 2 a of the cylindrical conductors 1 and 2 (Fig. 1), as well as the diameters D * and d * internal coaxial line, you can fully compensate for the component X _in [formula (71)]:

and taking into account the matching of the active component [formula (49)], it is possible to provide an input reflection coefficient of the balancing device 4 very close to zero. The specified selection (adjustment) of sizes is provided by numerical optimization using the WIPL-D three-dimensional electrodynamic modeling software package, freely available on the software market as an application on a CD to work: V.M. Kolundzja, JSOgnjanovic, and T. K. Sarkar, "WIPL-D: Microwave circuit and 3D EM simulation for RF and microwave applications. Software and User's manual", Norwood, MA: Artech House, 2005.

As a result of solving the optimization problem for the frequency f _c = 880 MHz and wave impedance ρ ₀ = 50 Ohm, the following optimal dimensions of the inventive radiator were found (in millimeters):

для основной поляризации (фиг.9, сплошная линия поз.20); б) в плоскости yoz для кросс-поляризации (фиг.10, сплошная линия поз.21); в) в плоскости хоу магнитного вектора

для основной поляризации (фиг.11, сплошная линия поз.22); г) в плоскости хоу для кросс-поляризации (фиг.12, сплошная линия поз.23).the length l of the collinear conductors 1 and 2 (Fig. 1) according to formula (51) is equal to: l = 0.289λ _s = 98.5 mm. The results of electrodynamic modeling with these dimensions are presented in Fig. 8 [line pos. 18 - return loss S ₁₁ (dB), line pos. 19 - isolation S ₁₄ (dB) of the inoperative diagonal arm of the strip directional coupler used as balancing device 4]. The following figures show radiation patterns: a) in the yoz plane of the electric vector

for the main polarization (Fig. 9, solid line, item 20); b) in the yoz plane for cross-polarization (Fig. 10, solid line, item 21); c) in the plane of the magnetic vector

for the main polarization (Fig. 11, solid line, item 22); d) in the hou plane for cross-polarization (Fig. 12, solid line, item 23).

The experimental studies of radiation patterns were carried out in an antenna laboratory with a reduced level of reflection of radio waves from its walls and ceiling, lined with a radio absorber, using a microwave generator, measuring receiver SMV 8.5 [Selectives Microvoltmeter 26 ... 1000 MHz, RPT, VEB Messeelektronic, Berlin, Fabr. - Nr. 07227] and a linearly polarized receiving horn antenna. In the same laboratory, the return loss S ₁₁ (dB) and the isolation S ₁₄ (dB) were measured using a complex transmission coefficient meter P4-11 (factory No. 02699, 1984), the generator unit of which was used as a microwave signal source for measuring the diagrams directivity in the far zone of Fraunhofer.

The results of experimental studies are shown in Fig.8-Fig.12. On Fig circles (pos.24) shows the return loss S ₁₁ (dB), and crosses (pos.25) - isolation S ₁₄ (dB). Further, the circles show: in Fig. 9 (pos. 26) - the main polarization in the yoz plane, in Fig. 10 (pos. 27) - the cross-polarization in the yoz plane, in Fig. 11 (pos. 28) - the main polarization in the plane of hou, Fig (pos. 29) - cross-polarization in the plane of hou.

The results obtained indicate the adequacy of the calculation methodology with the data of a full-scale experiment and the prospects of the inventive dipole emitter for practical use both as an independent antenna and in multi-element director, log-periodic and turnstile antennas. In this case, the remote ends 10 and 11 of the collinear cylindrical conductors 1 and 2 are connected to the antiphase outputs 6 and 7 of the balancing device 4 (Fig. 1), which does not create difficulties when using any balancing devices with spatially separated outputs and eliminates the need to develop specialized (modifications existing) balancing devices with adjacent outputs. The overall layout of the inventive dipole emitter (figure 1) indicates its good adaptation to the seats at the installation site and allows us to conclude about the increased, compared with the prototype, adaptability of assembly, installation and layout works.

(фиг.10). Представляется, что эти преимущества заявляемого дипольного излучателя будут все более весомы по мере увеличения рабочей частоты f_с, достигающей в современных системах связи и телекоммуникаций десятков гигагерц.In addition, the absence in the central region near the conductors 1 and 2 (Fig. 1) of any conductors helps to reduce the level of cross-polarization radiation, especially in the plane of the electric vector

(figure 10). It seems that these advantages of the inventive dipole emitter will be more significant as the operating frequency f _s increases, reaching tens of gigahertz in modern communication and telecommunication systems.

Claims (1)

A dipole emitter comprising two collinear identical cylindrical conductors, supplying a coaxial cable and a balancing device with input and two out-of-phase output arms, while adjacent ends of the collinear conductors are located in close proximity, the supplying coaxial cable is connected to the input arm of the balancing device, the output arms of which are connected to said conductors, characterized in that the output shoulders of the balancing device are connected to the remote ends of the specified x conductors.

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