RU2439588C1 - Method for electric power measurement in two-conductor networks with antitheft protection and measurement device - Google Patents

Abstract

FIELD: electricity. ^ SUBSTANCE: device contains sensor 1 of phase conductor current, sensor 2 of neutral lead current, sensor 3 of phase conductor current which is connected to phase conductor before entry into building, sensor 4 of load voltage connected in parallel to load 5, multipliers 6, 7, 8, mathematic treatment unit 9, indicator 10 and data remote transfer unit 11 and control centre. At that data remote transfer unit 11 contains modulating current generator 12, high frequency generator 13, phase manipulator 14, power amplifier 15 and transmission antenna 16. Control centre contains receiving antenna 17, high frequency amplifier 18, mixers 19 and 47, retrieval unit 20, heterodyne 21, IF amplifiers 22 and 48, detector (selector) 23 of phase manipulated (Pmn) signal, frequency analysers 24 and 26, phase doubler 25, comparator 27, threshold switch 28, delay circuit 29, switches 30 and 54, demodulator 31 of Pmn-signal, multipliers 32, 33 and 51, narrow-band filters 34, 37 and 52, low-pass filter 35, recording unit 36, phase-invertors 38, 41 and 44, summators 39, 42, 45 and 50, 90-phase shifters 46 and 49, amplitude detector 53. ^ EFFECT: improvement of interference resistance and reliability of data transfer about consumed power. ^ 2 cl, 7 dwg, 1 tbl

Description

The proposed method and device relate to electrical engineering and can be used to measure electrical energy in AC circuits for the purposes of commercial metering and detection of the fact and type of theft of electricity.

To date, the main means of measuring electric energy in single-phase consumers are induction meters containing a power converter into the rotational movement of the mobile system and a speed counter, and the current and voltage circuits of the converter are connected in series with a direct (phase) wire and parallel to the load circuit. These counters are sensitive to the direction of power and, in the absence of a stopper, have the property of reversibility. The readings of such counters can be reduced by forcing the disk to rotate in the opposite direction (Sedov P.G. Electric energy counters. - M.-L.: SEI, 1933).

A large number of static energy counters of a static system are also known, comprising current and voltage converters, power converters in the frequency of a pulse signal, and a pulse counter with a reading device. The load current converter of these counters is connected to the current of one usually direct (phase) load wire, and the power to pulse signal frequency converter is operable only for one direction of the load power (ed. Certificate of the USSR No. 1129526, G01R 11/00, 1984) .

The disadvantage of this method and devices that implement it is the low security of the measurement result. Since a unipolar conversion of power into the frequency of a pulsed electrical signal is used to construct non-reversible electronic electric energy meters, when the current direction changes, this conversion stops, and the direction of rotational motion changes in induction meters. The energy consumer has the opportunity to create such meter operating modes by connecting induction short circuits in order to suspend the accumulation of readings of the measured energy or even reduce this indication.

For individual homeowners, the so-called otmotchiki, which change the direction of the bill, are in high gear. In addition, in three-phase circuits with a grounded neutral, it is possible to use artificially or naturally grounded objects as a return wire. In this case, the commercially available meter does not take this part of the single-phase load into account.

There is also a known method and device for measuring electrical energy in two wire networks (RF patent No. 2077062, G01R 22/00, 2002), in which an instantaneous current signal of a phase wire is generated, an instantaneous current signal of a neutral wire is generated, a difference module of these signals is determined, determine the modulus of the sum of these signals, form the load power signal in the form of the sum of the modules of the difference in instantaneous powers of the indicated currents, the resulting signal is converted into a pulse frequency and served on the display.

The disadvantage of the data of the method and device is the poor protection of the measurement result from theft of electricity, since they do not provide for the possibility of establishing the type of theft used by an unscrupulous consumer, and they do not protect against the type of theft common among individual developers, such as connecting the load between the meter and commissioning.

There are also known methods and devices for measuring electrical energy in two wire networks with theft protection (ed. Certificate of the USSR No. 1580259; RF patents No. 2087918, 2212673, 2280256; German patent No. 2227846 and others).

Of the known methods and devices closest to the proposed are the "Method of measuring electrical energy in two-wire networks with theft protection and a device for its implementation" (RF patent No. 2280256, G01R 11/24, 2005), which are selected as the base objects.

The aforementioned method and device are based on the formation of instantaneous power signals of the neutral conductor currents and phase conductor currents measured before and after input into the structure of the supply line, the mathematical processing of these signals, comparing the values of the received signals, and the largest of them is selected to calculate the consumed electricity, and the ratio of all received signals is judged on the form of theft.

Basic technical solutions provide further improvement of the protection of the result of measuring the consumed electricity from an unscrupulous consumer, provide the measurement of the consumed electricity in full, and also allow controlling organizations to identify the presence and type of theft.

However, at the control point of known technical solutions, the same value of the intermediate frequency ω _pr can be obtained by receiving signals at two frequencies: ω _s and ω _s , i.e.

ω _ol = ω _s -ω _g and ω _ol = ω _g -ω _s ,

where ω _g is the frequency of the local oscillator.

Consequently, if ω tuning frequency _to take over the main receiving channel, along with it will be a mirror receiving channel frequency ω _of which is spaced from the main channel receiving at twice the value of the intermediate frequency 2ω _straight and located symmetrically (mirror) relative to frequency ω _g local oscillator (figure 5).

Conversion on the mirror channel of the reception occurs with the same conversion coefficient K _ol as on the main channel of the reception, so it most significantly affects the selectivity and noise immunity of the receiver.

In addition to the mirror, there are other additional (direct channel, Raman and intermodulation channels) receiving channels.

If the carrier frequency of the interference is equal to the intermediate frequency ω _CR , then a direct channel is formed (ω _p = ω _CR ). For this interference, the elements of the receiver act as simple transmission links.

If two or more powerful signals, for example, at frequencies ω ₁ and ω ₂ , simultaneously fall into the frequency band Δω _p1 located “to the left” of the passband Δω _{p of the} receiver, then they form a lot of false signals (intermodulation interference) on nonlinear elements, some of which fall into the passband Δω _{p of the} receiver (Fig.6).

If two or more powerful signals, for example, at frequencies ω ₃ and ω ₄ , simultaneously fall into the frequency band Δω _p2 located “to the right” of the passband Δω _{p of the} receiver, then they also form a lot of false signals (intermodulation interference) on nonlinear elements , some of which fall into the passband Δω _{p of the} receiver (Fig. 7).

In the general case, intermodulation frequencies are determined by the expressions:

m ω ₁ ± n ω ₂ = ω _mn , ω ₃ ± n ω ₄ = ω _mn .

The use of highly selective quartz filters at an intermediate frequency ω _pr , improving selectivity on adjacent channels, can help in suppressing interference from one powerful out-of-band signal, but help powerlessly in suppressing intermodulation interference.

The presence of false signals (interference) received via additional channels leads to a decrease in noise immunity and reliability of the transmission of information about the consumed electricity, the presence, type and time of its theft.

An object of the invention is to increase the noise immunity and reliability of the transmission of information about the consumed electricity, the presence, type and time of its theft by suppressing false signals (interference) received via additional channels.

The problem is solved in that the method of measuring electrical energy in two-wire networks with theft protection, consisting in accordance with the closest analogue, in the formation of signals of instantaneous power currents of phase and zero wires, mathematical processing of these signals and comparing the processing results by which the fact is judged theft and the amount of energy consumed, while forming a signal of the instantaneous current power of the phase wire, measured before entering the structure, compare the values of the received signals, you they select the largest of them and use them to calculate the consumed electricity, and according to the state of all the obtained instantaneous power values, they judge the type of theft, form a modulating code containing the consumer code, the amount of electricity consumed, the type and time of theft, manipulate the phase harmonic oscillation in accordance with modulating code, amplify the generated complex signal with phase manipulation in power, radiate it on the air, at the control point, search for signals in a given range of hours frequency by tuning the local oscillator frequency, the received signal is converted in frequency using the local oscillator frequency, the first voltage of the intermediate frequency is isolated, its phase is doubled, the spectrum width of the received signal is measured at the fundamental and doubled intermediate frequencies, they are compared with each other and, if there is a significant difference, the fact of detection is established a complex signal with phase shift keying, multiply the detected complex signal with phase shift keying at an intermediate frequency with a voltage reference between of the current frequency, a low-frequency voltage proportional to the modulating code is isolated, it is registered and simultaneously multiplied with the received complex signal with phase shift keying at the intermediate frequency, harmonic oscillation is isolated and used as a reference voltage, differs from the closest analogue in that the useful signal received by the main channel is converted in frequency using the voltage of the local oscillator phase-shifted by 90 °, the second voltage of the intermediate frequency is isolated, it is shifted about the phase by 90 °, summarize with the first intermediate frequency voltage, multiply the first total voltage obtained with the received useful signal, isolate the harmonic voltage at the local oscillator frequency, detect it and use the detected voltage to allow further processing of the received useful signal, false signal (interference), received on the mirror channel, is converted in frequency using the local oscillator voltage and the local oscillator voltage, 90 ° out of phase, the third and fourth n intermediate frequency voltages, respectively, the fourth intermediate frequency voltage is phase shifted by 90 ° and summed with the third intermediate frequency voltage, compensating for it, the spurious signal (interference) received on the first combinational channel is converted in frequency using the local oscillator voltage and the local oscillator voltage shifted 90 ° phase, the fifth and sixth intermediate frequency voltages are isolated, respectively, the sixth intermediate frequency voltage is shifted 90 ° phase and summed with the fifth voltage m of the intermediate frequency, compensating for it, the false signal (interference) received through the second combinational channel is converted in frequency using the local oscillator voltage and the local oscillator voltage 90 °, the seventh and eighth intermediate frequency voltages are isolated, the eighth voltage is shifted in phase 90 ° and summed with the seventh voltage of the intermediate frequency, the resulting second total voltage is multiplied with the received false signal (interference), form a harmonic voltage at the second frequency harmonic the local oscillator, which does not fall into the passband of the narrow-band filter, thereby suppressing the false signal (interference) received via the second combination channel, emit a false signal (interference) received via the direct channel, invert it by 180 ° phase and sum with the received false signal (interference), offsetting it allocate two or more spurious signals (interference) contained in a frequency band Δω _n1 disposed "left" by Δω _n receiver bandwidth and ingress which can lead to the formation of intermetallic ulyatsionnyh interference invert their phase by 180 ° and summed with the received spurious signals (interference), offsetting them allocate two or more spurious signals (interference) fed to band Δω _n2 frequencies located 'right' of Δω bandwidth receiver _n and falling into which can lead to the formation of intermodulation interference, invert them in phase by 180 ° and summarize with the received false signals (interference), compensating for them.

The problem is solved in that a device for measuring electrical energy in two-wire networks with theft protection, containing, in accordance with the closest analogue, a zero-wire current sensor, the output of which is connected to the first input of the first multiplier, a phase wire current sensor, the output of which is connected to the first input of the second multiplier, an additional phase wire current sensor, included in the phase wire before the two-wire power network is introduced into the building, an additional multiplier and a control point, the second inputs of the first and second multipliers are connected to the output of the load voltage sensor, the outputs of the multipliers are connected to the corresponding inputs of the mathematical processing unit of the sensor signals, the first output of which is connected to the indicator of the measurement result, the output of the additional phase wire current sensor is connected to the first input of the additional multiplier, the second input of which connected to the output of the load voltage sensor, the output of the additional multiplier is connected to the corresponding input of the mathematical unit Sensor signal processing, the second output of which is connected to the input of the additional information transmission unit, the control point consists of a series-connected receiving antenna and a high-frequency amplifier, series-connected search block, a local oscillator, a first mixer and a first intermediate frequency amplifier, series-connected phase doubler, second spectrum analyzer, a comparison unit, the second input of which is connected to the output of the first spectrum analyzer, a threshold block, the second input of which through The delay is connected to its output, the first key, the first multiplier, the second input of which is connected to the output of the low-pass filter, the first narrow-band filter, the second multiplier, the second input of which is connected to the output of the first key, low-pass filter and registration unit, differs from the closest analogue the fact that it is equipped with a second and third narrow-band filters, three phase inverters, four adders, two 90 ° phase shifters, a second mixer, a second intermediate frequency amplifier, and a third multiplier m, an amplitude detector and a second key, and the second narrow-band filter, the first phase inverter, the first adder, the second input of which is connected to the output of the high-frequency amplifier, the first bandpass filter, the second phase inverter, the second adder, the second input of which is connected to the output of the high-frequency amplifier with the output of the first adder, the second bandpass filter, the third phase inverter, the third adder, the second input of which is connected to the output of the second adder, the second mixer, the second input of which is through the first 90 ° phase shifter connected to the second output of the local oscillator, second intermediate frequency amplifier, second 90 ° phase shifter, fourth adder, the second input of which is connected to the output of the first intermediate frequency amplifier, third multiplier, the second input of which is connected to the output of the third adder, third narrow-band filter , an amplitude detector and a second switch, the second input of which is connected to the output of the fourth adder, and the output is connected to the inputs of the first spectrum analyzer and phase doubler and to the second input of the first switch, the second input of the first mixer is connected to the output of the third adder.

The structural diagram of the device that implements the proposed method is presented in figures 1 and 2. The structural diagram of the control point is presented in figure 3. Timing diagrams explaining the operation of the control point are shown in Fig.4. Frequency diagrams explaining the principle of formation of additional receiving channels are presented in FIGS. 5, 6 and 7. The types of theft of electricity are illustrated in the table.

A device that implements the proposed method includes a phase wire current sensor 1, a neutral wire current sensor 2, a phase wire current sensor 3 included in the phase wire before being put into the structure, a load voltage sensor 4 connected in parallel with the load 5. Sensor 3 is installed on the overhead line so that it can be seen from the street. The outputs of the sensors 1, 2, 3 are connected to the first inputs of the multipliers 6, 7, 8, the second inputs of which have a signal from the output of the voltage sensor 4. The outputs of the multipliers 6, 7, 8 are connected to the corresponding inputs of the mathematical processing unit 9, the outputs of which are connected to the indicator 10 and the remote information transmission unit 11.

At the same time, the remote information transmission unit 11 is made in the form of a modulator code generator 12, a phase manipulator 14, the second input of which is connected to the output of a high-frequency generator 13, a power amplifier 15 and a transmitting antenna 16 connected in series to the second output of the mathematical processing unit 9.

The control point includes a series-connected receiving antenna 17, a high-frequency amplifier 18, a second narrow-band filter 37, a first phase inverter 38, a first adder 39, the second input of which is connected to the output of a high-frequency amplifier 18, a first band-pass filter 40, a second phase inverter 41, and a second adder 42 the second input of which is connected to the output of the first adder 39, the second bandpass filter 43, the third phase inverter 44, the third adder 45, the second input of which is connected to the output of the second adder, the first mixer 19, the second input of which the local oscillator 21 is connected to the output of the search unit 20, the first intermediate frequency amplifier 22, the fourth adder 50, the third multiplier 51, the second input of which is connected to the output of the third adder 45, the third narrow-band filter 51, the amplitude detector 53, the second key 54, the second input of which is connected with the output of the fourth adder 50, a phase doubler 25, a second spectrum analyzer 26, a comparison unit 27, the second input of which is connected through the first spectrum analyzer 24 to the output of the second key 54, a threshold unit 28, whose second input is through the delay line 29 connected to its output, the first key 30, the second input of which is connected to the output of the second key 54, the first multiplier 32, the second input of which is connected to the output of the low-pass filter 37, the first narrow-band filter 34, the second multiplier 33, the second input of which is connected to the output of the first a key 30, a low pass filter 35, and a registration unit 36. The second mixer 47 is connected in series to the output of the third adder 45, the second input through the first phase shifter 46 is 90 ° connected to the second output of the local oscillator 21, the second intermediate frequency amplifier 48 and the second phase shifter 49 are 90 °, the output of which is connected to the second input of the fourth adder 50.

Spectrum analyzers 24 and 26, a phase doubler 25, a comparison unit 27, a threshold unit 28, and a delay line 29 form a detector (selector) 23 of the phase-shift (PSK) signal.

Multipliers 32 and 33, a narrow-band filter 34 and a low-pass filter 35 form a PSK signal demodulator 31.

A device that implements the proposed method works as follows.

The current sensors 1, 2, 3 generate signals proportional to the current of the neutral wire and the currents in the phase wire before entering the structure and after entering the structure. From these signals, as well as from the output signal of the voltage sensor 4 with the help of multipliers 6, 7, 8, signals of the instantaneous power of the currents of the neutral wire and the phase wire measured in different places are formed. Block 9 of mathematical processing, which can be designed on the basis of any microprocessor, converts the signals from multipliers 6, 7, 8 into a digital code, analyzes the values of instantaneous powers, selects the multiplier from which the signal with the highest modulus is being received, sums the signals from selected multiplier to determine the amount of electricity consumed. At the same time, the mathematical processing unit 9 analyzes the ratios of the magnitudes of the signals from the multipliers 6, 7, 8 and, in accordance with one or another combination value of these signals, outputs a signal about one or another form of theft to the corresponding output. Signals about the consumed electricity, as well as signals about the type of theft and time taken to record the fact of theft, are sent to indicator 10, which displays the above information, as well as to block 11 for the remote transmission of information.

Various theft schemes are shown in figure 2. The combination of signals at the input of block 9 of the mathematical processing corresponding to the types of theft are shown in the table.

Position 1 of the table corresponds to the case when there are no thefts, the currents in the phase and neutral wires correspond to the norm, the signals from all sensors are equal.

Position 2 of the table corresponds to the case when there is a theft of electricity due to a hidden connection R _n1 between the input to the building and the meter. In this case, the current flows through the circuit: phase wire, sensor 3, R _n1 , sensor 1, load 5, sensor 2, neutral wire. At the same time, a larger current flows through the sensor 3 than through the sensors 1 and 2. Therefore, the mathematical processing unit 9 selects the channel of the sensor 3 with a multiplier 6 to calculate the consumed energy, and the code of the theft type mentioned is set at the corresponding output.

Position 3 of the table corresponds to the case when theft by grounding the load _occurs (connection R _n2 , _figure 2), or a leakage current to the ground of more than 25 mA appeared as a result of a fault in the wiring. It can be shown, similarly to the previous case, that a larger current flows through sensor 3 than through sensors 1 and 2. To calculate the consumed electricity, the signal from the sensor channel 3 is used. The combination of the values of the signals from sensors 1, 2, 3 will be different and an indication of this type of theft or a malfunction will appear on the other output of the mathematical processing unit 9.

Position 4 of the table corresponds to the case when there is a theft of electricity with a phase change to the meter and grounding the load by connecting R _n3 ( _figure 2). Now the currents in the sensor circuit 2 and 3 will have a large value, which will be used by the mathematical processing unit 9 to calculate the consumed electricity, and the combination of the values of the sensor signals 1, 2, 3 will activate the corresponding output of the mathematical processing unit 9.

Position 5 of the table corresponds to the case when the connected electrical appliances are faulty and there is a leakage current (conditionally R _n4 , _figure 2). To account for the consumed power, a sensor signal 1 or 3 is used, an indication of the type of theft at the corresponding output of the mathematical processing unit 9.

Position 6 of the table corresponds to the case when there is a bypass, open, short circuit of the supply wires from the sensor 3. Now the signal from the sensor 3 is smaller in magnitude of the signals of the sensors 1 and 2. This combination of sensor signals causes a signal at the corresponding output of the mathematical processing unit 9, for metering the consumed energy, the signal of sensors 1 or 2 is used.

In addition, when using a device of the “unwinder” type, the amount of “coiled” electricity is automatically taken into account as consumed, since the summation takes place modulo instantaneous power and does not depend on the direction of the current.

Thus, all the most common types of theft are not only counted by the counter, but also determined by species. The amount of electricity consumed and the type of theft, as well as the theft time, are displayed on indicator 10.

With the help of the block 11 remote transmission of information, it is possible to read the above data regardless of the desire of the consumer and without entering the room. Shaper 12 generates a modulating code M (t) (Fig. 4, b) containing the consumer code, the amount of electricity consumed, the type and time of theft. This modulating code is fed to the first input of the phase manipulator 14, to the second input of which harmonic oscillation is output from the output of the high-frequency generator 13 (Fig. 4, a)

U _c (t) = υ _c Cos ((ω _c t + φ _s ), 0≤t≤T _C ,

where υ _c , ω _s , φ _s , T _C - amplitude, carrier frequency, initial phase and duration of harmonic oscillation,

At the output of the phase manipulator 14, a complex signal with phase shift keying (QPSK) is generated (Fig. 4, c)

U ₁ (t) = υ _c · Cos [ω _c t + φ _к (t) + φ _с ], 0≤t≤Т _C ,

where φ _к (t) = {0, π} is the manipulated phase component that displays the phase manipulation law in accordance with the modulating code M (t), and φ _к (t) = const for k · τ ₃ <t <(k + l) · τ ₃ and may change abruptly at t = k · τ _e , i.e. at the borders between elementary premises (k = 1, 2, ..., N-1);

τ _e , N is the duration and number of chips that make up a signal of duration T _C (T _C = N · τ _e ), N = g + l + m + n;

g is the number of chips that display the consumer code;

l is the number of elementary premises that reflect the amount of energy consumed;

m is the number of elementary premises that display the theft code;

n is the number of elementary packages that display the theft time,

which, after amplification in the power amplifier 15, is radiated by the transmitting antenna 16 to the air.

At the control point, the search for PSK signals in a given frequency range Df is carried out using the search unit 20, which periodically with a period T _P according to a sawtooth law changes the frequency of the local oscillator 21

U _G (t) = υ _g Cos (ω _g t + πγt ² + φ _g ), 0≤t≤T _P ,

where υ _r , ω _g , φ _g - amplitude, initial frequency and initial phase of the local oscillator voltage;

γ = D · f / T _P - rate of tuning the frequency of the local oscillator.

The received complex QPSK signal U ₁ (t) (Fig. 4, c) from the output of the receiving antenna 17 through the high-frequency amplifier 18 and adders 39, 42, 45, for which only one arm operates, is fed to the first inputs of the mixers 19 and 47 , to the second inputs of which the voltage of the local oscillator 21 is applied:

U _Г (t) = υ _g Cos (ω _g t + πγt ² + φ _g ),

U _Г ^' (t) = υ _g · Cos (ω _g t + πγt ² + φ _g + 90 °), 0≤t≤T _P.

At the output of the mixers 19 and 47, voltages of combination frequencies are generated. Amplifiers 22 and 48 distinguish the first and second voltages of the intermediate (difference) frequency:

U _PR1 (t) = υ _pr1 · Cos [ω _pr t + φ _к (t) -πγt ² + φ _pr1 ],

U _PR2 (t) = υ _pr1 · Cos [ω _pr t + φ _к (t) -πγt ² + φ _pr1 -90 °], 0≤t≤T _C ,

where υ _np1 = 1/2 · υ _c · υ _g ;

ω _CR = ω _with -ω _g is the intermediate frequency;

ω _pr1 = φ _s -φ _g ,

which are complex signals with combined phase shift keying and linear frequency modulation (PSK - LFM).

The voltage U _PR2 (t) from the output of the intermediate frequency amplifier 48 is supplied to the input of the phase shifter 49 by 90 °, at the output of which a voltage is generated

U _PR2e '(t) = υ _pr1 · Cos [ω _pr t + φ _к (t) -πγt ² + φ _pr1 -90 ° + 90 °] =

= υ _pr1 · Cos [ω _pr t + φ _k (t) -πγt ² + φ _pr ], 0≤t≤T _C ,

Voltages U _PR1 (t) and U _PR2 '(t) are supplied to two inputs of the adder 50, the output of which forms the first total voltage

U _Σ1 (t) = υ _Σ1 · Cos [ω _pr t + φ _к (t) -πγt ² + φ _pr1 ], 0≤t≤T _C ,

where υ _Σ1 = 2υ _pr1 .

This voltage is supplied to the second input of the multiplier 51, the first input of which receives the received signal U ₁ (t). At the output of the multiplier 51, a harmonic voltage is generated

U ₂ (t) = υ ₂ Cos (ω _g t + φ _g ), 0≤t≤T _C ,

where υ ₂ = 1/2 * υ _c · υ _Σ1 .

Since the tuning frequency ω _{H1 of the} narrow-band filter 52 is chosen equal to the initial frequency ω _{g of the} local oscillator 21 (ω _H1 = ω _g ), the voltage U ₂ (t) is emitted by the narrow-band filter 52, is detected by the amplitude detector 53 and fed to the control input of the key 54, opening him. In the initial state, keys 30 and 54 are always closed. The first total voltage U _Σ1 (t) from the output of the adder 50 through the public key 54 is fed to the input of the detector (selector) 23, which consists of spectrum analyzers 24 and 26, a phase doubler 25, a comparison unit 27, a threshold unit 28, and a line 29 delays.

At the output of the phase doubler 25, which can be used as a multiplier, two inputs of which the same voltage U _Σ1 (t) is supplied, a voltage is generated

U ₃ (t) = υ ₃ Cos (2ω _pr t + πγt ² + 2φ _pr1 ), 0≤t≤T _С ,

where υ ₃ = 1/2 · υ _Σ1 ² .

Since 2φ _K (t) = {0.2π}, phase manipulation is already absent in the indicated voltage.

The width of the spectrum Δf _{2 of the} second harmonic of the signal is determined by the duration T _{C of the} signal (Δf ₂ = 1 / T _C ), while the width of the spectrum Δf _{C of the} input QPSK signal is determined by the duration τ _{e of} its elementary premises (Δf _C = 1 / τ _e ), t .e. the width of the spectrum Δf _{2 of the} second harmonic of the signal is N times smaller than the width of the spectrum Af _{c of the} input signal (Δf _C / Δf ₂ = N).

Therefore, when the phase of the QPSK signal is doubled, its spectrum “folds” N times. This makes it possible to detect (select) the QPSK signal even when its power at the receiver input is less than the power of interference and noise.

The width of the spectrum Δf _{C of the} input QPSK signal is measured using the spectrum analyzer 24, and the spectrum width Δf _{2 of the} second harmonic of the signal is measured using the spectrum analyzer 26.

Voltages U _I and U _II , proportional to Δf _C and Δf _2, respectively, from the outputs of the spectrum analyzers 24 and 26 are fed to two inputs of the comparison unit 27, the output of which is a positive voltage that exceeds the threshold level U _POR in the threshold block 28. The threshold level U _POR is chosen so that it does not exceed random interference.

When the threshold level U _POR is exceeded, a constant voltage is generated in the threshold block 28, which is supplied to the control input of the search unit 20, turning it off, to the control input of the key 30, opening it.

Upon termination of the frequency tuning of the local oscillator 21 at the output of the adder 50, the following voltage is generated (figure 4, g)

U _{Σ '1 (t) = υ Σ1 · Cos} [ω t + φ _{forth in} (t) -πγt _pr1 + φ ^2], 0≤t≤T _C,

which through the public key 30 is fed to the input of the demodulator FMN signal 31, which consists of multipliers 32 and 33, a narrow-band filter 34 and a low-pass filter 35.

A necessary condition for the operation of the demodulator of the QPSK signal is the presence of a reference voltage having a constant initial phase and a frequency equal to the frequency of the QPSK signal. In the proposed demodulator, the reference voltage is extracted directly from the PSK signal.

Currently, a number of original devices have been developed (Pistolkors A.A., Siforov V.I., Travina S.A., Kostas D.F.) circuits in which the reference voltage required for synchronous detection of the received PSK signal is extracted from the received FMN signal itself. However, the phenomenon of “reverse operation” is inherent in these devices, which significantly reduces the reliability of isolating the modulating code from the received PSK signal.

The proposed demodulator of FMN signals is free from the phenomenon of "reverse work".

The total voltage U _Σ ' ₁ (t) from the output of the adder 50 through the public keys 54 and 30 is simultaneously supplied to the first inputs of the multipliers 32 and 33. The reference voltage is applied to the second input of the multiplier 33 from the output of the narrow-band filter 34 (Fig. 4, d)

U ₀ (t) = υ ₀ · Cos (ω t + φ _{straight pr1),} 0≤t≤T _C.

As a result of multiplication, the resulting voltage is formed

U ₄ (t) = υ _n · Cosφ _to (t) + υ _n · Cos [2ω _pr t + φк (t) + 2φ _pr ],

where υ _n = 1/2 · υ _Σ · υ ₀ ;

Analog modulating code (figure 4, e)

U _H (t) = υ _n · Cosφ _k (t),

allocated by the low-pass filter 35 and registered by the registration unit 36.

At the same time, the voltage U _H (t) is supplied to the second input of the multiplier 32, at the output of which a harmonic oscillation is formed

U _O (t) = υ ₄ · Cos (ω _pr t + φ _pr ) + υ ₄ · Cos [ω _pr t + 2φ _to (t) + φ _pr1 ] =

= 2υ ₄ · Cos (ω _np t + φ _CR1 ) = υ ₀ · Cos (ω _pr t + φ _CR1 ),

where υ ₄ = 1/2 · υ _Σ · υ _n .

This oscillation is distinguished by a narrow-band filter 34, used as a reference voltage, and fed to the second input of the multiplier 33.

The delay time τ _{3 of} the delay line 29 is selected so that it is possible to register an analog of the modulating code. After this time, the voltage from the output of the delay line 29 is supplied to the control input of the threshold unit 28 and resets its contents to zero. In this case, the key 30 is closed, and the search unit 20 is transferred to the search mode.

When the next QPSK signal is detected, the device operates in a similar way.

The device detects all the most common types of theft, detects such theft as connecting the load between the input to the building and the meter, detects a malfunction of the devices at the consumer, takes into account all the energy consumed regardless of the type of theft.

The operation of the device described above corresponds to the case of receiving a useful QPSK signal through the main channel at a frequency ω _s (Fig. 5).

If a false signal (interference)

U ₃ (t) = υ ₃ Cos (ω ₃ t + φ ₃ ), 0≤t≤T ₃ ,

taken through the mirror channel at a frequency of ω ₃ , then the amplifiers 22 and 48 are allocated the third and fourth intermediate frequency voltage

U _PR3 (t) = υ _pr3 · Cos (ω _pr t + πγt ² + φ _pr3 ),

U _PR4 (t) = υ _pr3 · Cos (ω _pr t + πγt ² + φ _pr3 + 90 °], 0≤t≤T ₃ ,

where υ _pr3 = 1/2 · υ ₃ · υ _g ;

ω _CR = ω _g -ω ₃ - intermediate frequency;

ω _pr4 = ω _g -ω ₃ .

The fourth voltage U _PR4 (t) from the output of the intermediate frequency amplifier 48 is supplied to the input of the phase shifter 49 by 90 °, at the output of which a voltage is generated

U _PR ' ₄ (t) = υ _pr3 · Cos (ω _pr t + πγt ² + φ _pr3 + 90 ° + 90 °) =

= -υ _pr3 · Cos (ω _pr t + πγt ² + φ _pr3 ), 0≤t≤T ₃ .

Voltages U _PR3 (t) and U _PR ' ₄ (t), supplied to the two inputs of the adder 50, are compensated at its output.

Therefore, a false signal (interference) received through the mirror channel at a frequency of ω ₃ is suppressed by an “outer ring” consisting of mixers 19 and 47, a local oscillator 21, amplifiers 22 and 48 of an intermediate frequency, phase shifters 46 and 49 by 90 °, an adder 50 and implements phase compensation method.

If a false signal (interference)

U _К1 (t) = υ _к1 · Cos (ω _к1 t + φ _к1 ), 0≤t≤T _К1 ,

is received on the first combinational channel at a frequency ω _k1 , then the amplifiers 22 and 48 are allocated the fifth and sixth voltage of an intermediate frequency:

U _PR5 (t) = υ _pr5 Cos (ω _pr t + πγt ² + φ ₅ ),

U _PR6 (t) = υ _pr5 · Cos (ω _pr t + πγt ² + φ _pr5 + 90 °), 0≤t≤T _K1 ,

where υ _np5 = 1/2 · υ _k1 · υ _g ;

ω _ave = 2ω _r -ω _k1 - intermediate frequency;

φ _pr5 = φ _g -φ _k1 .

The voltage U _PR5 (t) from the output of the intermediate frequency amplifier 48 is supplied to the input of the phase shifter 49 by 90 °, at the output of which a voltage is generated

U _PR'6 (t) = υ _pr5 · Cos (ω _pr t + πγt ² + φ _pr5 + 90 ° + 90 °) =

= -υ _pr5 Cos (ω _pr t + πγt ² + φ _pr5 ).

Voltages U _PR5 (t) and U _PR ' ₆ (t), supplied to the two inputs of the adder 50, are compensated at its output.

Therefore, a false signal (interference) received through the first combinational channel at a frequency ω _k1 is suppressed by the “outer ring” that implements the phase compensation method.

If a false signal (interference)

U _K2 (t) = υ _k2 · Cos (ω _k2 t + φ _k2 ), 0≤t≤T _K2 ,

taken through the second combinational channel at a frequency ω _k2 , then the amplifiers 22 and 48 are allocated the seventh and eighth voltage of the intermediate frequency:

U _PR7 (t) = υ _pr7 · Cos (ω _pr t-πγt ² + φ _pr7 ),

U _PR8 (t) = υ _pr7 · Cos (ω _pr t + πγt ² + φ _pr7 -90 °), 0≤t≤T _K2 ,

where υ _pr7 = 1/2 · υ _k2 · υ _g ;

_straight ω = ω _k2 -2ω _g - intermediate frequency;

φ _pr7 = φ _k2 -φ _g .

The voltage U _PR8 (t) from the output of the intermediate frequency amplifier 48 is supplied to the input of the phase shifter 49 by 90 °, at the output of which a voltage is generated

U _PR ' ₈ (t) = υ _pr7 · Cos (ω _pr t-πγt ² + φ _pr8 -90 ° + 90 °) =

= υ _pr7 · Cos (ω _pr t-πγt ² + φ _pr8 ).

Voltages U _PR7 (t) and U _PR ' ₈ (t) are supplied to two inputs of the adder 50, at the output of which the total voltage is formed

U _Σ2 (t) = υ _Σ2 Cos (ω _pr t-πγt ² + φ _pr8 ), 0≤t≤T _K2 ,

where υ _Σ2 = 2υ _pr7 .

This voltage is supplied to the second input of the multiplier 51, the first input of which receives the received false signal (interference) U _K2 (t). At the output of the multiplier 51, a voltage is generated

U ₅ (t) = υ ₅ Cos (2ω _g t + φ _g ), 0≤t≤T _K2 ,

where υ ₅ = 1/2 · υ _k2 · υ _Σ2 ,

which does not fall into the passband of the narrow-band filter 52. The key 54 does not open and the false signal (interference) received on the second combination channel at a frequency ω _{k2 is} suppressed by the “inner ring” consisting of a multiplier 51, a narrow-band filter 52, an amplitude detector 53, key 54 and implements the method of narrow-band filtering.

If a false signal (interference) is received through the direct channel at an intermediate frequency ω _pr

U _n (t) = υ _n · Cos (ω _ave t + φ _n), 0≤t≤T _P,

then it is distinguished by a narrow-band filter 37, the tuning frequency ω _{n2 of} which is chosen equal to the intermediate frequency ω _pr (ω _n2 = ω _pr ), phase inverted by 180 ° phase inverter 38

U _n '(t) = - υ _n · Cos (ω _ave t + φ _n), 0≤t≤T _n.

The voltage U _P (t) and U _P '(t) supplied to the two inputs of the adder 39, at its output are compensated.

Therefore, a false signal (interference) received through the direct passage channel at an intermediate frequency ω _pr is suppressed by a filter plug, consisting of a narrow-band filter 37, a phase inverter 38, an adder 39 and implementing a phase compensation method.

If two or more false signals (interference) fall into the frequency band Δω _p1 located “to the left” of the passband Δω _{p of the} receiver, then they are separated by a band-pass filter 40, 180 ° phase inverted in the phase inverter 41 and fed to the second input of the adder 42 where they are compensated (Fig.6). In this case, the tuning frequency ω _n3 and the passband Δω _{p1 of the} bandpass filter 40 are selected as follows:

_H3 ω = ω ₁ + ω _2/2; Δω _n1 = ω ₁ -ω ₂ ,

where ω ₁ , ω ₂ are the boundary frequencies that determine the frequency band Δω _p1 , located "to the left" of the passband Δω _{p of the} receiver, if two or more false signals (interference) get into it, leads to the formation of intermodulation interference.

Consequently, false signals (interference) received in the frequency band Δω _p1 are suppressed by a filter plug consisting of a bandpass filter 40, a phase inverter 41, an adder 42 and implementing a phase compensation method.

If two or more false signals (interference) fall into the frequency band Δω _p2 located “to the right” of the passband Δω _{p of the} receiver, then they are separated by a band-pass filter 43, phase inverted by 180 ° in the phase inverter 44 and fed to the second input of the adder 45 where they are compensated (Fig.7). The tuning frequency ω _n4 and the passband Δω _{p2 of the} bandpass filter 43 are selected as follows:

_H4 ω ₃ = ω + ω _4/2; Δω _n2 = ω ₄ -ω ₃ ,

where ω ₃ , ω ₄ are the boundary frequencies that determine the frequency band Δω _p2 located “to the right” of the passband Δω _{p of the} receiver, if two or more false signals (interference) get into it, leads to the formation of intermodulation interference.

Consequently, false signals (interference) received in the frequency band Δω _p2 are suppressed by a filter plug consisting of a bandpass filter 43, a phase inverter 44, an adder 45 and implementing a phase compensation method.

To increase the reliability and reliability of remote measurement of consumed electricity and to detect the presence, type and time of theft, a radio channel and complex signals with phase shift keying are used. Each consumer has its own code and operates at a certain frequency.

At the same time, complex signals with phase manipulation have high energetic and structural secrecy.

The energy secrecy of these signals is due to their high compressibility in time or in the spectrum with optimal processing, which reduces the instantaneous radiated power. As a result, a complex signal at the receiving point may be masked by noise and interference. Moreover, the energy of a complex signal with phase shift keying is by no means small, it is simply distributed over the time-frequency domain so that at each point in this region the signal power is less than the power of noise and interference.

The structural secrecy of complex signals with phase shift keying is due to a wide variety of their forms and significant ranges of parameter changes, which makes it difficult to optimize or at least quasi-optimal processing of complex signals of an a priori unknown structure.

Thus, the proposed method and device in comparison with the basic objects provide increased noise immunity and reliability of the transmission of information about the consumed electricity, the availability, type and time of its theft. This is achieved by suppressing false signals (interference) received through the direct channel, mirror, Raman and intermodulation channels.

Table A method of measuring electric energy in two-wire networks with theft protection and a device for its implementation No. p / p Designation Type of connected load Power ratio Working sensor one R _n Normal connection P ₁ = P ₂ = P ₃ Any 2 R _n1 Hidden connection to the counter P ₃ > P ₁ = P ₂ 3 3 R _n2 Concealed connection to meter or earth leak P ₃ > P ₁ = P ₂ 3 four R _n3 Phase reversal or earth leakage P ₁ <P ₂ = P ₃ 2 or 3 5 R _n4 Grounding or earth leakage P ₂ <P ₁ = P ₃ 1 or 3 6 Bypass, open, short circuit of the supply wires of the external current sensor P ₃ <P ₁ = P ₂ 1 or 2 Notes: R _n - standard load connection. Turn on the load for theft: R _n1 - hidden connection to the counter; R _n2 - hidden connection to the meter, with grounding of the load; R _n3 - load connection with grounding and polarity reversal; R _n4 - connection by grounding the load;

Claims (2)

1. The method of measuring electric energy in two-wire networks with theft protection, which consists in generating signals of instantaneous power currents of phase and zero wires, mathematical processing of these signals and comparing the processing results, which are used to judge the fact of theft and the amount of energy consumed, signal of the instantaneous current power of the phase wire, measured before entering the structure, compare the values of the received signals, select the largest of them and use to calculate the consumed electric energy, and by the ratio of the values of all the instantaneous power values obtained, they judge the type of theft, form a modulating code containing the consumer code, the amount of electricity consumed, the type and time of theft, manipulate the phase harmonic oscillation in accordance with the modulating code, amplify the generated complex signal in power with phase shift keying, radiate it on the air, at the control point, search for signals in a given frequency range by tuning the local oscillator frequency, convert it in frequency at the signal being sampled using the local oscillator frequency, the first voltage of the intermediate frequency is isolated, its phase is doubled, the spectrum width of the received signal is measured at the fundamental and doubled intermediate frequencies, they are compared with each other and, if there is a significant difference, the fact of detecting a complex signal with phase shift keying is established, and the complex a signal with phase shift keying at an intermediate frequency with a reference voltage of an intermediate frequency, a low-frequency voltage proportional to the incriminating code, register it and simultaneously multiply with the received complex signal with phase shift keying at the intermediate frequency, emit harmonic oscillation and use it as a reference voltage, characterized in that the useful signal received on the main channel is converted in frequency using the local oscillator voltage, phase shifted by 90 °, a second intermediate frequency voltage is isolated, phase shifted by 90 °, summed with the first intermediate frequency voltage, the first sum obtained The voltage is multiplied with the received useful signal, the harmonic voltage is isolated at the local oscillator frequency, it is detected and the detected voltage is used to allow further processing of the received useful signal, the false signal (interference) received through the mirror channel is converted in frequency using the local oscillator voltage and the local oscillator voltage , phase-shifted by 90 °, emit the third and fourth intermediate frequency voltage, respectively, the fourth intermediate voltage often you are phase shifted by 90 ° and summed with the third voltage of the intermediate frequency, compensating for it, the false signal (interference) received through the first combination channel is converted in frequency using the voltage of the local oscillator and the voltage of the local oscillator phase-shifted by 90 °, the fifth and the sixth voltage of the intermediate frequency, respectively, the sixth voltage of the intermediate frequency is phase shifted by 90 ° and summed with the fifth voltage of the intermediate frequency, compensating for it, a false signal (interference) received by the second the bination channel, the frequency is converted using the local oscillator voltage and the local oscillator voltage 90 °, the seventh and eighth intermediate frequency voltages are isolated, the eighth voltage is phase shifted 90 ° and summed with the seventh intermediate frequency voltage, the resulting second total voltage is multiplied with the received false signal (interference), they form a harmonic voltage at the second harmonic of the local oscillator frequency, which does not fall into the passband of the narrow-band filter, suppressing The most false signal (interference) received via the second combinational channel, a false signal (interference) received through the direct channel is isolated, it is inverted 180 ° in phase and summed with the received false signal (interference), two or more are isolated false signals (interference) falling into the frequency band Δω _p1 , located "to the left" of the passband Δω _{p of the} receiver and falling into which can lead to the formation of intermodulation interference, invert them in phase by 180 ° and summarize with the received false signals and (by interference), compensating for them, isolate two or more false signals (interference) falling into the frequency band Δω _p2 located “to the right” of the passband Δω _{p of the} receiver and falling into which can lead to the formation of intermodulation interference, invert them in phase 180 ° and summed with the received false signals (interference), compensating for them. 2. A device for measuring electrical energy in two-wire networks with anti-theft protection, comprising a zero wire current sensor, the output of which is connected to the first input of the first multiplier, a phase wire current sensor, the output of which is connected to the first input of the second multiplier, an additional phase wire current sensor, included in the phase wire before entering the two-wire supply network into the building, an additional multiplier and a control point, while the second inputs of the first and second multipliers are connected to the output of the sensor and the load voltage, the outputs of the multipliers are connected to the corresponding inputs of the mathematical processing unit of the sensor signals, the first output of which is connected to the indicator of the measurement result, the output of the additional phase wire current sensor is connected to the first input of the additional multiplier, the second input of which is connected to the output of the load voltage sensor, the output of the additional the multiplier is connected to the corresponding input of the block of mathematical processing of sensor signals, the second output of which is connected to the input at the additional information transmission unit, the control point consists of a series-connected receiving antenna and a high-frequency amplifier, series-connected search block, a local oscillator, a first mixer and a first intermediate frequency amplifier, a phase doubler in series, a second spectrum analyzer, a comparison unit, the second input of which is connected with the output of the first spectrum analyzer, a threshold block, the second input of which through the delay line is connected to its output, the first key, the first multiplier, in the second input of which is connected to the output of the low-pass filter, the first narrow-band filter, the second multiplier, the second input of which is connected to the output of the first key, the low-pass filter and the registration unit, characterized in that it is equipped with a second and third narrow-band filters, three phase inverters, four adders , two 90 ° phase shifters, a second mixer, a second intermediate frequency amplifier, a third multiplier, an amplitude detector and a second key, and to the output of the high-frequency amplifier the second narrow-band filter, the first phase inverter, the first adder, the second input of which is connected to the output of the high-frequency amplifier, the first band-pass filter, the second phase inverter, the second adder, the second input of which is connected to the output of the first adder, the second band-pass filter, the third phase inverter, and the third adder are connected , the second input of which is connected to the output of the second adder, the second mixer, the second input of which is connected through the first phase shifter 90 ° to the second output of the local oscillator, the second amplifier is an intermediate hour from the second phase shifter 90 °, the fourth adder, the second input of which is connected to the output of the first intermediate frequency amplifier, the third multiplier, the second input of which is connected to the output of the third adder, the third narrow-band filter, the amplitude detector and the second switch, the second input of which is connected to the output the fourth adder, and the output is connected to the inputs of the first spectrum analyzer and phase doubler and to the second input of the first key, the second input of the first mixer is connected to the output of the third adder.

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