RU2436202C1 - Wideband 180-degree microwave phase changer - Google Patents

Abstract

FIELD: electricity.

SUBSTANCE: phase changer comprises two 3-decibel quadrature directed couplers at joined lines (QDC), two matched loads, four attenuators, two field microwave transistors. Field transistors are used in amplification mode. The inlet of the first QDC is the device inlet. The uncoupled outlet of the first QDC is connected to the first matched load. The outlet with phase shift of 0° of the first QDC is connected to the outlet with phase shift of 0° of the second QDC via the first attenuator, the first field transistor and the third attenuator. The outlet with phase shift of -90° of the first QDC is connected to the outlet with phase shift of -90° of the second QDC via the second attenuator, the second field transistor and the fourth attenuator. The uncoupled outlet of the second QDC is connected to the second matched load. The inlet of the second QDC is used as the device outlet. Supply is sent only to one of two field transistors, thus the path of signal passage from the inlet to the outlet is changed.

EFFECT: reduced maximum possible phase error.

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Description

The invention relates to the field of radio engineering, and more specifically to microwave phase shifters, introducing a constant phase shift in a wide frequency band, and can be used in microwave transmitting and receiving devices.

Known broadband pass-through microwave phase shifter in Schiffman sections [Coats R.P. An Octave-Band Switched-Line Microstrip 3-b Diode Phase Shifter // IEEE Transactions On Microwave Theory And Techniques, Vol.21, No.7, July 1973]. The Schiffman section is a reactive all-through quadrupole based on connected lines [Schiffman B.M. A new class of broad-band microwave 90-degree phase shifters // IRE Transactions on Microwave Theory and Techniques, Vol.6, No4, April 1958]. This phase shifter has the following disadvantages. First, for the implementation of a single-stage phase shifter with a phase shift of 180 °, the highest value of the coupling coefficient between the lines is required [Ultra-wideband microwave devices / ed. A.P. Krenitsky and V.P. Meshchanov. - M .: Radio and communications, 2001. - p. 215, 218]. As a result, the design of the phase shifter on some types of lines, in particular on the microstrip, is difficult to implement. Secondly, when the 180 ° phase-shifting section is implemented as two 90 ° phase-shifting sections connected in cascade, the topology of the phase shifter is complicated and its dimensions increase. Thirdly, to switch between the states of the phase shifter, you need two switches containing a total of at least four active elements, which complicates the design of the product and reduces the reliability of the device.

A known broadband pass-through microwave phase shifter, the action of which is based on a change in the direction of excitation of the slit line segment [Patent for invention No. SU 1542363 A1, class. H01P 1/185. Discrete phase shifter. Date of publication 1995.03.10, author V. Sledkov]. The disadvantage of this phase shifter is the need for manufacturing and fixing in the case of a double-sided board.

A known broadband pass-through microwave phase shifter, the action of which is based on switching the conductors of the transmission line [United States Patent No. 5,680,079, 180-degree phase shifter, Inami et al., October 21, 1997]. The disadvantage of this phase shifter is the impossibility of implementation in a purely microstrip design.

Known broadband reflective microwave phase shifter, adopted as a prototype, which consists of a quadrature 3-decibel directional coupler (CCW), two outputs of which are loaded on semiconductor diodes [Miyaguchi K., et al. An ultra wide band band reflection type 180 deg phase shifter with series and parallel LC circuits. // 2001 MTT-S International Microwave Symposium Digest, vol. 1, p. 237-240]. To realize a phase shift of 180 ° in a wide frequency band, the microwave impedance of the diodes is changed, applying a different control voltage so that at the outputs of the CCW alternately there is an idling mode and a short circuit mode.

This phase shifter has the following disadvantage: with imperfect matching of the KNO arms, even in the case of diodes identical in parameters, equal in fission power, phase difference between the KNO outputs 90 °, there is an additional source of phase error that cannot be compensated [Garver R.V. Broad-Band Diode Phase Shifters // IEEE Transactions On Microwave Theory And Techniques, Vol.4, No.5, May 1972, p.317]. With a small mismatch of the arm of the CCW, a standing wave arises between the CCW and the diode. This standing wave leads to the appearance of a phase error, which can reach Δφ = ± 4arcsin | Г |, where Г is the reflection coefficient from the coupler inputs. For example, with | Г | = 0.032 (power reflection coefficient - 30 dB), the maximum phase error is ± 7.33 °.

The aim of the invention is to reduce the maximum possible phase error of a broadband 180-degree microwave phase shifter.

To achieve this goal, a broadband phase shifter is proposed, comprising a 3-decibel quadrature directional coupler on coupled lines (CCF), the input of which is the input of the phase shifter. According to the invention, a second 3-decibel CCW, two matched loads, four attenuators, two microwave field-effect transistors are introduced into it. Field effect transistors are used in amplifier mode. Each KNO has an input, an output with a phase shift of 0 °, an output with a phase shift of -90 °, an isolated output. The input of the first CCW is the input of the device. The decoupled output of the first CCW is connected to the first matched load. The output with a phase shift of 0 ° of the first KNO is connected through the first attenuator to the gate of the first field-effect transistor. The output with a phase shift of -90 ° of the first KNO is connected through the second attenuator to the gate of the second field-effect transistor. The drain of the first field-effect transistor is connected through the third attenuator to the output with a phase shift of 0 ° of the second CCW. The drain of the second field-effect transistor is connected through the fourth attenuator to the output with a phase shift of -90 ° of the second CCW. The decoupled output of the second CCW is connected to the second matched load. The input of the second CCW is used as the output of the device.

The goal is achieved by converting a reflective phase shifter into a loop-through so that imperfect matching does not lead to a phase error.

The combination of distinctive features and properties of the proposed phase shifter from the scientific and technical literature are unknown, therefore, it meets the criteria of novelty and inventive step.

Functional diagram of the device shown in figure 1. The topology of the board made the layout of the phase shifter shown in figure 2.

The phase shifter contains KNO 1, the input of which is the input of the device. The isolated output of the KNO 1 is connected to the matched load 2. The output with a phase shift of 0 ° KNO 1 is connected through the attenuator 3 to the gate of the field effect transistor 4. The output with a phase shift of -90 ° KNO 1 is connected through the attenuator 5 to the gate of the field effect transistor 6. The drain of the field effect transistor 4 is connected through an attenuator 7 to the output with a phase shift of 0 ° KNO 8. The drain of the field-effect transistor 6 is connected through an attenuator 9 to the output with a phase shift of -90 ° KNO 8. The isolated output of the KNO 8 is connected to the matched load 10. The input of the KNO 8 is used as the output devices.

Attenuator 3 is identical in properties to attenuator 5, field-effect transistor 4 is identical in properties to field-effect transistor 6, attenuator 7 is identical in properties to attenuator 9, KNO 1 is identical in properties to KNO 8.

The principle of operation of the phase shifter is based on the properties of 3-decibel quadrature directional couplers on connected lines. It is known that for couplers on coupled lines, the phase difference between the outputs is 90 ° and does not depend on the frequency [Bova N.T., Efremov Yu.G., Konin V.V. and other microwave electronic devices. - Kiev: Technique, 1984. - p.28]. The field effect transistor, which is not powered, provides a signal isolation of about 20 dB.

In the first working state of the phase shifter, the supply voltage is applied to the field-effect transistor 4, the supply voltage is not supplied to the field-effect transistor 6, the signal from the input of the device passes to the output of the device along the path input KNO 1 - output KNO 1 with a phase shift of 0 ° - attenuator 3 - field transistor 4 - attenuator 7 - KNO 8 output with a phase shift of 0 ° - KNO 8 input.

In the second operating state of the phase shifter, the supply voltage is applied to the field effect transistor 6, the supply voltage is not applied to the field effect transistor 4, the signal from the input of the device passes to the output of the device along the path input KNO 1 - output KNO 1 with a phase shift of -90 ° - attenuator 5 - field transistor 6 - attenuator 9 - output KNO 8 with a phase shift of -90 ° - input KNO 8.

Thus, the signal passing through the phase shifter in the first operational state has a phase shift of 180 ° with respect to the signal passing through the phase shifter in the second operational state. The properties of CCW 1.8 ensure the constancy of the indicated phase shift in a wide band.

Consistent loads 2, 10 and attenuators 3, 5, 7, 9 serve to match the device and minimize the maximum possible phase error.

In the first working state of the phase shifter, a supply voltage is not supplied to the field effect transistor 6, therefore, the input of the attenuator 5 and the output of the attenuator 9 are mismatched and decoupled from each other. The maximum reflection coefficient from the input of the attenuator 5 G ₁ modulo does not exceed the squared modulus of the attenuator 5. The maximum reflection coefficient from the output of the attenuator 9 G ₂ modulo does not exceed the squared modulus of the attenuator 9.

Thus, the isolated output of the KNO 1 is loaded on a matched load, the output of the KNO 1 with a phase shift of -90 ° is loaded on the load with a reflection coefficient of G ₁ . The isolated output of KNO 8 is loaded to a matched load, the output of KNO 8 with a phase shift of -90 ° is loaded to a load with a reflection coefficient of G ₂ . Under these conditions, the transmission coefficient from the input of the KNO 1 to the output of the KNO 1 with a phase shift of 0 ° is equal to:

,

,

Where

S _ij are the coefficients of scattering matrices KNO 1 and KNO 8;

S ₁₂ - transmission coefficient from input to output with a phase shift of 0 °;

S ₁₃ - transmission coefficient from input to output with a phase shift of -90 °;

S ₂₃ - transmission coefficient from the output with a phase shift of 0 ° to the output with a phase shift of -90 ° (determines the isolation between the outputs);

S ₃₃ - reflection coefficient from the output with a phase shift of -90 °.

The transmission coefficient from the input of the KNO 8 to the output of the KNO 8 with a phase shift of 0 ° is equal to:

.

.

In the second operating state of the phase shifter, the supply voltage is not supplied to the field effect transistor 4, therefore, the input of the attenuator 3 and the output of the attenuator 7 are mismatched and decoupled from each other. The reflection coefficient from the input of the attenuator 3 is equal to G ₁ , the reflection coefficient from the output of the attenuator 7 is equal to G ₂ . Thus, the isolated output of the KNO 1 is loaded on a matched load, the output of the KNO 1 with a phase shift of 0 ° is loaded on the load with a reflection coefficient of G ₁ . The isolated output of KNO 8 is loaded to a matched load, the output of KNO 8 with a phase shift of 0 ° is loaded to a load with a reflection coefficient of G ₂ . Under these conditions, the transfer coefficient from the input of the KNO 1 to the output of the KNO 1 with a phase shift of -90 °:

where S ₂₂ is the reflection coefficient from the output with a phase shift of 0 °.

The transmission coefficient from the input of the KNO 8 to the output of the KNO 8 with a phase shift of -90 ° is equal to:

Phase Shift:

φ = arg (K ₃ K ₄ ) -arg (K ₃ K ₄ )

The values of E ₁ and E _{2 are} determined by the reflection coefficient at the inputs of the attenuators 3 and 5 G ₁ , the reflection coefficient at the outputs of the attenuators 7 and 9 G ₂ :

as | Г ₁ | → 0 and | Г ₂ | → 0 E ₁ → 0, Е ₂ → 0.

Also, the values of E ₁ and E _{2 are} determined by the isolation of the CCW:

as | S ₂₃ | → 0 E ₁ → 0, E ₂ → 0.

вследствие свойств КНО.

due to the properties of the CCW.

The maximum phase error occurs at the least favorable ratio of the complex quantities E ₁ and E ₂ .

Thus, by decreasing | G ₁ |, | G ₂ |, | S ₂₃ |, it is possible to reduce the maximum phase error.

When the reflection coefficient for power from the outputs of the coupler is -30 dB, isolation -30 dB, equal to the division of power between the outputs, attenuation introduced by attenuators 6 dB:

max | G ₁ | = max | G ₂ | = 0.5

| S ₂₂ | = | S ₃₃ | = | S ₂₃ | = 0,032

| S ₁₂ | = | S ₁₃ | = 0.707

max | E ₁ | = max | E ₂ | = 0,0326

Δφ = ± 3.74 °

Thus, the invention allows to reduce the maximum phase error, compared with the prototype.

At the filing date, a prototype phase shifter was developed and manufactured with the following parameters: operating frequency range of 6-14 GHz, phase shift between two states of 180 ± 3 °, VSWR of the input and output in two states of no more than 1.4, the introduced signal attenuation - 2 ± 1 dB, the difference between the attenuation of the signal in two states is not more than 2 dB.

Claims (1)

A 180-degree broadband microwave phase shifter containing a first 3-decibel quadrature directional coupler on connected lines, the input of which is a phase shifter input, characterized in that a second 3-decibel quadrature directional coupler on connected lines is inserted into it, the first and second matched loads, the first , the second, third and fourth attenuators, the first and second field-effect transistors that are used in the amplifying mode, and the decoupled output of the first directional coupler is connected to the first matched load, the output with a phase shift of 0 ° of the first directional coupler is connected through the first attenuator to the gate of the first field effect transistor, the output with a phase shift of -90 ° of the first directional coupler is connected through the second attenuator to the gate of the second field effect transistor, the drain of the first field effect transistor is connected through the third attenuator to the output with a phase shift of 0 ° of the second directional coupler, the drain of the second field-effect transistor is connected through the fourth attenuator to the output with a phase shift of -90 ° W of the opposite directional coupler, the decoupled output of the second directional coupler is connected to the second matched load, the input of the second directional coupler is the output of the device.

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