RU2276463C2 - Spatial-temporal code with non-zero complex weighing for transmission by multiple antennas - Google Patents

Abstract

FIELD: communication systems with distributed transmission, in particular, method and device for non-zero complex weighing and space-time encoding of signals for transmission by multiple antennas.

SUBSTANCE: method and device provide for expansion of space-time block code N×N' to space-time block code M×M', where M>N, with utilization of leap-like alternation of symbols phase in space-time block code N×N', to make it possible to transfer space-time block code from distributed antennas in amount, exceeding N'.

EFFECT: distribution of transmission from more than two antennas.

2 cl, 16 dwg

Description

Technical field

This invention relates to a method and apparatus for achieving transmit diversity in communication systems, in particular to a method and apparatus for non-zero complex weighting and space-time coding of signals for transmission by multiple antennas.

BACKGROUND OF THE INVENTION

With the development of communication systems in the construction of communication systems are becoming increasingly important requirements for equipment and performance. Promising wireless systems, i.e. third and fourth generation systems, in comparison with the first-generation analog systems and second-generation digital systems currently in use, are designed to provide, in addition to high-quality voice services, high-quality and high-speed data transmission services. Along with the performance requirements for system services, there are restrictions on the design of equipment, which will greatly affect the design of mobile terminals. Third and fourth generation wireless mobile terminals should be more compact, lighter, more economical units, capable of also providing the advanced voice and data services provided in these promising wireless systems.

Time-varying multipath fading occurs in wireless systems when the transmitted signal propagates to the receiver along multiple paths, causing fading due to constructive and destructive summation of the signals in the receiver. Several methods are known for overcoming the effects of multipath fading, for example, time interleaving with error correction coding, implementing frequency diversity using spread spectrum methods or transmitter power control methods. However, each of these approaches has disadvantages with respect to use in third and fourth generation wireless systems. Time interleaving can lead to undesirable delays, spreading techniques may require a large bandwidth to overcome the large coherence bandwidth, and higher power transmitter may be required to apply power control methods than would be desirable for advanced receiver-to-transmitter feedback methods , which increases the complexity of the mobile terminal. All these shortcomings make it difficult to achieve the desired characteristics of the third and fourth generation mobile terminals.

Another method of overcoming the effects of multipath fading in wireless systems is antenna diversity. With diversity reception, two or more physically separated antennas are used to receive the transmitted signal, and then the transmitted signal is processed by combining and switching to generate the received signal. The disadvantage of diversity reception is that the necessary physical separation between the antennas can make diversity reception practically inapplicable in the forward link in new wireless systems where a small size mobile terminal is required. A second method for implementing antenna diversity is diversity transmission. When transmitting diversity, the signal is transmitted from two or more antennas, and then it is processed in the receiver using the maximum likelihood sequence estimation unit (OPDM), receivers with minimum mean square error (ISCED), receivers of maximum posterior probability or their approximations. Diversity transmission is more applicable for the forward link in wireless systems since it is easier to implement multiple antennas in a base station than in a mobile terminal.

The diversity transmission for the case of two antennas is well understood. Alamouti has proposed a diversity transmission method for two antennas that provides second-order diversity for signals with complex values. S. Alamouti, "A Simple Transmit Diversity Technique for Wireless Communications," IEEE Journal on Selected Areas of Communications, pp. 1451-1458, October 1998. The Alamouti method provides for the simultaneous transmission of two signals from two antennas during a symbol period. During one symbol period, a signal denoted by S0 is transmitted from the first antenna, and a signal denoted by S1 is transmitted from the second antenna. During the next symbol period, the signal -S1 * is transmitted from the first antenna, and the signal S0 * is transmitted from the second antenna, where * denotes the complex conjugation operator. A similar diversity transmission system is also possible in the code area. For example, two copies of the same character can be transmitted in parallel using two orthogonal Walsh codes. Similar methods can also be used to construct a spatial frequency coding method.

The Alamouti method cannot be directly extended to the case of more than two antennas. Tarok et al. Proposed a method for using space-time (PV) block coding at 1/2 and 3/4 speeds for transmission from three and four antennas using complex signal diagrams. V. Tarokh, H. Jafarkhani and A. Calderbank, "Space-Time Block Codes from Orthogonal Designs", IEEE Transactions on Information Theory, pp. 1456-1467, July 1999. The disadvantage of this method is the loss of transmission speed and that the multilevel character of symbols subjected to PV encoding places an increased requirement on the maximum / average ratio for the transmitted signal and imposes strict requirements on the design of a linear power amplifier. Additional methods mitigating these problems are proposed in O. Tirkkonen and A. Hottinen, "Complex space-time block codes for four Tx antennas", Proc. Globecom 2000, November 2000, San Francisco, USA. Other proposed methods include an orthogonally diversity transmission method (ORP) + a space-time transmission diversity (PVR) scheme for four antennas at a speed of 1. L. Jalloul, K. Rohani, K. Kuchi and J. Chen, "Performance Analysis of CDMA Transmit Diversity Methods "Proceedings of IEEE Vehicular Technology Conference, Fall 1999; and M. Harrison, K. Kuchi, "Open and Closed Loop Transmit Diversity at High Data Rates on 2 and 4 Elements", Motorola Contribution to 3GPP-C30-19990817-017. This method requires an external code and provides second-order diversity by virtue of the PVRP block (Alamouti block) and a second-order interleaving gain from using the PPR block. The performance of this method depends on the power of the external code. Since this method requires external code, it is not applicable for non-encoded systems. For the case of a convolutional code with a speed of 1/3, the performance of the ODP + PVRP method and the Tarok method of PV block coding at a speed of 3/4 are approximately the same. Another method for speed 1 is proposed in O. Tirkkonen, A. Boariu and A. Hottinen, "Minimal non-orthogonality rate 1 space-time block code for 3+ Tx antennas", Proc. ISSSTA 2000, September, 2000. The method proposed in this publication allows for high performance, but requires a sophisticated receiver.

Thus, it is desirable to have a method and apparatus that would provide the advantage of diversity transmission from more than two antennas and, at the same time, would not increase the complexity of the system design.

SUMMARY OF THE INVENTION

The present invention provides a method and apparatus for non-zero complex weighting and space-time coding of signals for transmission by multiple antennas. The method and apparatus provide an extension of the space-time block code N × N ′, where N is the number of transmission paths and N ′ is the number of output symbols per transmission path to a space-time block code M × M ′, where M> N generated from using repetition and non-zero complex weighting of characters in the space-time block code N × N ′, so that it is possible to transmit the space-time block code along M diversity transmission paths. Diversity paths may include separate antennas or beams. The time length of the larger code M ′ may be equal to the time length of the source code N ′. In accordance with the method and device, the input symbol stream is converted to generate a conversion result containing a space-time block code. Then N output streams of the space-time block code, each of which consists of N ′ output symbols, are repeated, and at least one of the repeated streams is subjected to nonzero complex time-weighting to generate M streams from N ′ output symbols for transmission along M paths diversity transmission. Nonzero complex weighing may include a phase shift.

According to one embodiment, N is at least 2 and M is at least 3. Then, at least two of the N streams of N ′ output symbols corresponding to the N source streams of N ′ output symbols are transmitted from the first at least one antennas, and at least one of the MN streams of N ′ symbols subjected to non-zero complex weighting is transmitted from the second at least one antenna. The first at least one antenna and the second at least one antenna may comprise any of M antennas.

According to another embodiment, the method and apparatus can be implemented in a transmitter having common or dedicated pilot channels that provide an effective channel estimate of the coefficients needed to decode the space-time code. In this embodiment, either common or dedicated pilot channels, or both, can be implemented in the transmitter. According to one alternative to this embodiment, training symbols are transmitted over the N diversity transmission paths, which makes it possible to estimate the N independent diversity transmission paths. For this, the code sequence of the dedicated pilot channel can be multiplexed into each of N streams of N ′ output symbols of the original space-time block code to generate N streams from N ′ output symbols and the sequence of the pilot channel. Then, non-zero complex weighting can be applied to generate M phase-shifted streams from N ′ output symbols and a pilot channel sequence. Then, at least two of the N source streams of N ′ output symbols and the pilot channel sequence are transmitted from the first at least one antenna, and at least one of the MN complexly weighted streams of N ′ output symbols and the pilot channel sequence are transmitted from the second at least one antenna. Another way to provide an estimate of N channels is that common pilot channels are transmitted so that N common pilot channels are transmitted from each of the first at least one antenna, and MN complexly weighted copies of some of the N common pilot channels are transmitted from each from the second at least one antenna. The complex weights used for common channels in each of the second at least one antenna are identical to those used to construct M-N additional complex-weighted streams from N ′ output symbols from the original N flows from N ′ output symbols. In these embodiments, the receiver may or may not be aware of the method used to extend the space-time block code N × N ′ to the space-time block code M × N ′ and the temporal weighting sequences used.

According to other embodiments, where N is at least 2 and M is at least 3, pilot channels can be used to provide an estimate of at least N + 1 diversity transmission paths. Then, at least one of N streams of N ′ output symbols corresponding to N source streams of N ′ output symbols is transmitted from the first at least one antenna, and at least one of MN complexly weighted streams of N ′ symbols is transmitted from the second at least one antenna. Different common pilot channels are transmitted from each of the first at least one antenna and the second at least one antenna. In these embodiments, the receiver needs to at least partially know the method used to extend the space-time block code N × N ′ to the space-time block code M × N ′ and the temporal weighting sequences used.

In various embodiments, complex weighting can be applied by applying a periodic or random complex weighting template to each of the symbol streams subjected to complex weighting. You can also pre-set the relationship between the complex weights of the symbol streams transmitted on different antennas.

Brief Description of the Drawings

Fig. 1a is a block diagram of a transmitter according to an embodiment of the invention;

Fig. 1b is a block diagram of fragments of an STRP transmitter with a common pilot channel according to an embodiment of the invention;

FIG. 2 is a block diagram of fragments of a PVRP transmitter with a common pilot channel in accordance with another embodiment of the invention; FIG.

Figure 3 is a block diagram of fragments of a transmitter with a dedicated pilot channel, corresponding to another variant embodiment of the invention;

FIG. 4 is a block diagram of fragments of an embodiment of a receiver for use with the transmitter shown in FIG. 1;

FIG. 5 is a block diagram of fragments of an embodiment of a receiver for use with the transmitter shown in FIG. 2 or the transmitter shown in FIG. 3;

FIG. 6 is an embodiment of a retraction of the PVRP demodulator 508 shown in FIG. 5;

7 is a block diagram of fragments of a TAC transmitter according to an embodiment of the invention;

Fig. 8 is a block diagram of fragments of an ODP transmitter according to an embodiment of the invention;

FIG. 9 is a block diagram of fragments of an embodiment of a receiver for use with the transmitter shown in FIG. 7;

FIG. 10 is a block diagram of fragments of an embodiment of a receiver for use with the transmitter shown in FIG. 8;

11 is a block diagram of fragments of a transmitter of a long block code block corresponding to a variant embodiment of the invention;

12 is a block diagram of fragments of a PVRP transmitter with a common / dedicated pilot channel, according to another embodiment of the invention;

FIG. 13 is a block diagram of fragments of a receiver for use with the transmitter shown in FIG. 12;

Fig - block diagram of fragments of the receiver for use in controlling the power of the transmitter shown in Fig;

15 is a diagram defining a phase shift pattern that can be used in various embodiments of the invention.

DETAILED DESCRIPTION OF THE INVENTION

FIG. 1 a shows a block diagram of a transmitter 150 according to an embodiment of the invention. The transmitter 150 has an input 152 for receiving an input symbol stream, a block code processor 154 for converting an input symbol stream to generate a transform result represented by an orthogonal space-time block code, and for outputting 2 symbol streams of a transform result, a non-zero complex weighting unit 156 for non-zero complex weighting of the first of two streams of characters, block 158 non-zero complex weighting for non-zero complex weighting of the second of two x symbol streams, RF transmitter 160 for transmitting a first symbol stream with Ant.1, RF transmitter 162 for transmitting a non-zero complex weighted symbol stream with Ant.2, RF transmitter 164 for transmitting a second symbol stream with Ant.3 and an RF transmitter 166 for transmitting a phase-shifted second symbol stream from Ant.4. Antennas Ant.1-Ant.4 can be polarized relative to each other to provide improved diversity reception. For example, Ant.1 or Ant.2 can be vertically polarized relative to the horizontal polarization of Ant.3 or Ant.4, respectively. An embodiment of the transmitter 150 shown in FIG. 1 a may be implemented in various forms suitable for different technologies and systems in order to extend the 2 × N ′ block code for transmission over 4 transmission diversity paths. At transmitter 150, each of the 4 transmission diversity paths includes a separate antenna, Ant. 1-Ant. 4. Transmit diversity can be used in various systems, for example, code division multiple access (CDMA) systems, time division multiple access (TDMA) systems, or any other type of digital communication system. As an alternative to the embodiment of FIG. 1 a, non-zero complex weighting can be performed on only some of the transmission paths in order to create relative phase shifts between the transmissions from Ant.1 and Ant.2 or Ant.3 and Ant.4. For example, non-zero complex weighting can also be applied to the input signals of the RF transmitter 160 and 164, creating a version of the non-zero complex weighting of each of the symbol streams, but maintaining a relative phase shift between the transmitted signals. As an alternative to transmitter 150, less than 4 antennas can be used to implement 4 diversity paths. For example, the signals arriving at the RF transmitters 164 or 166 can be combined with each other and transmitted from one antenna. Other alternatives are possible, according to which less than 4 explode paths are used, for example, non-zero complex weighting and transmission along two explode paths of only one of the 2 data streams. As an alternative to the embodiment shown in FIG. 1 a, a non-zero complex weighting operation can be performed at the output of the RF transmitter units 160, 162, 164, 166, i.e. non-zero complex weighting can be implemented as a continuous phase change after modulation and filtering in the base band of symbols subjected to space-time encoding.

Various alternatives of non-zero complex weighting are possible for these transmissions with Ant.2 and Ant.4. For example, for Ant.4, you can apply the phase pattern W ₁ (t) = exp (j * pi * phase_in_ degrees / 180) used for Ant.2, and the phase pattern -W ₁ (t), phase shifted 180 degrees relative to W ₁ (t). Examples of the 4FMn diagram are the phase shift pattern in degrees {0, 90, 180, 270} for Ant. 2 and {180, 270, 0, 90} for Ant. 4. Other examples of patterns are {0, 45, 90, 135, 180, 225, 270, 315} for 8 FPS and {0, 22.5, 45, 67.5, ... 337.5} for 16 FPS. 15 is a diagram defining another phase shift pattern that can be used in various embodiments of the invention. This sequence of shifts in degrees {0, 135, 270, 45, 180, 315, 90, 225} can be transmitted from antenna 2 using the shift pattern in degrees {180, 315, 90, 225, 0, 135 in antenna 4 , 270, 45}. The phase shift can be periodic or random. A periodic phase shift refers to a given phase pattern, for example, to a periodically repeating complex weighting factor W ₁ (t). Complex weights can be set so that the sequence of complex weights defines the maximum length path so that successive samples of the effective channel are as independent as possible. In this way, the need for interleaving can be eliminated and thus a low transmission delay can be achieved. A pseudo-random phase shift can be a sequence of randomly selected phases from the PPS diagram. Alternatively, another non-zero complex weighting scheme in which the phase difference between successive phase states is as small as possible has an advantage in estimating channel coefficients or metrics related to power control based on a channel subjected to non-zero complex weighting. In this case, phase states can still cover 360 degrees over one coding block. As with conventional systems, channel interleaving can be used in embodiments. You can also jointly implement a sequence of non-zero complex weighing and an interleaver so that the symbols at the output of the interleaver are as independent as possible. In addition, by changing the relative phase between antennas 1 and 2 and antennas 3 and 4, respectively, the method can be implemented so as to have a phase shift or change in all antenna elements, but to maintain relative phase shifts between antennas 1 and 2 and antennas 3 and 4 For example, when changing the phase, in order to realize an effective change corresponding to 100 Hz, you can use 50 Hz phase change in antenna 1 and -50 Hz phase change in antenna 2. Similarly for antennas 3 and 4.

The phase shift can change every T seconds. The choice of T depends on the total duration of the data symbols and the method used to estimate the channel coefficients. You can maintain a constant phase over the total duration of the data symbols within at least one space-time coding unit, and, to ensure the correct channel estimation, you can use the appropriate dedicated or common pilot sequence / training sequence. The pilot sequence may be a Walsh code used in CDMA systems, or a sequence of training symbols with good correlation properties, which is used for channel estimation in TDMA. Pilot symbols can use the same nonzero complex weights as the data in the space-time block. Alternatively, pilot channels may be transmitted without phase change. In this case, the effective channel for the data can be derived from a previously known phase hopping pattern together with the channel estimate obtained from the channel without phase change. In cases where non-zero complex weighting is applied to the common pilot channels, the same or different phase patterns can be applied to the data channels and the common pilot channels. Channel estimation using pilot or training sequences without phase change (transmitted on shared or dedicated channels) provides higher channel estimates because the channel is more stationary.

FIG. 1 b shows a block diagram of fragments of a space-time transmit diversity (SSR) transmitter 100 with a common pilot channel according to an embodiment of the invention. The transmitter 100 can operate as an extension of the 4-antenna transmit diversity to version 99 of the standard of the third generation system of broadband CDMA (WDMKR). The transmitter 100 comprises an input 126, a block code processor 124, traffic channel symbol stream processing taps inputs 102a-102d, antenna gain units 104a-104d, phase shifters 106a and 106b, phase shifter inputs 112a and 112b, code multipliers 108a-108d, inputs 114a-114d pilot processing taps, antenna gain units 116a-116d, code multipliers 118a-118d, high-frequency transmitter 128, including high-frequency transmitters 128a-128d and antennas Ant.1-Ant.4.

1b, data to be transmitted, including channel-coded and interleaved input symbol stream X (t) containing S1S2 characters, is input 126. Block code processor 124 converts every two received S1S2 characters to generate a result transforms containing the orthogonal space-time block code 2 × 2. According to an embodiment, the block code processor 124 may perform Alamouti transform to generate the block code in the form of the following matrix:

(1)

(one)

Then, the matrix is divided into 4 streams of 2 symbols, each of the streams being fed to one of the inputs 102a-102d of the tap for processing the stream of symbols of the traffic channel. According to FIG. 1, the stream S1S2 enters 102a, S1S2 enters 102b, -S2 * S1 * enters 102c and -S2 * S1 * enters 102d. Non-zero complex weighing is performed by the antenna gain units 104a-104d and the phase shifters 106a and 106b. The antenna gain for each processing tap is controlled by the antenna gain blocks 104a-104d. After adjusting the antenna gain, the phase shifters 106a and 106b phase shift the stream S1S2 output from the antenna gain unit 104b and the stream -S2 * S1 output from the antenna gain unit 104d. The phase shifter control units 112a and 112b can control the phase shifters 106a and 106b to cause a phase shift using a continuous or discrete phase change pattern. Then, the CDMA scrambling code is supplied to code multipliers 108a-108d to generate an S1S2 stream to an RF transmitter 128a for transmission from Ant.1, S1S2 (exp (jΦk1)) to an RF transmitter 128b for transmission from Ant.2, -S1 * S2 * - to the RF transmitter 128c for transmission from Ant.3 and -S2 * S1 (exp (jΦk2)) - to the RF transmitter 128d for transmission from Ant.4. High-frequency transmitters can carry out the formation of modulating pulses in the base band, modulation and conversion to the carrier frequency. In some implementations, an abrupt or continuous phase change after the steps of generating pulses in the baseband and modulation can be optionally applied.

Sequences X1-X4 of the common pilot channel are fed to the inputs 114a-114d of the pilot processing processing tap. Then, the pilot sequences are separately processed by the antenna gain units 116a-116d and the code multipliers 118a-118d. The encoded output signals of the code multipliers 118a-118d are supplied to the RF transmitters 128a-128d, respectively, of the RF transmitter 128.

Then, the pilot sequence X1 is transmitted from Ant.1, the pilot sequence X2 is transmitted from Ant.2, the pilot sequence X3 is transmitted from Ant.3 and the pilot sequence X4 is transmitted from Ant.4.

Figure 4 shows a block diagram of fragments of the receiver for use with the transmitter 100 shown in fig.1b. Figure 4 shows the signal processing for one receiving tap of the receiver. Received pilot sequences X1-X4 transmitted from the transmitter 100 are received and fed to the channel estimate processing taps 402a-402d, respectively. The channel estimation block 404 performs the channel estimation function, for example, the function of a moving average low-pass filter, for each channel 1 - channel 4. The channel 1 - channel 4 estimates are output from outputs 406a-406d to adder 410a, phase shifter 408a, adder 410b and phase shifter 408b. The phase shifter 408a receives the input signal from the phase shifter control unit 414a and reports the channel 2 estimate the same phase shift used in the traffic channel symbols S1S2 transmitted from Ant.2 of the transmitter 100. The phase shifter 408b receives the input signal from the phase shifter control unit 414b and reports the channel estimate 4, the same phase shift used in the -S2 * S1 * symbols of the traffic channel transmitted from Ant.4 of transmitter 100. Adder 410a combines the phase-shifted version of the estimate for channel 2 with the estimate for channel 1, and the adder 410b combines with the phase-wise version of the estimate for channel 4 with the estimate for channel 3. The combined estimate for channels 1 and 2 (412a) and the combined estimate for channels 3 and 4 (412b) are fed to the STPD demodulator 418, which processes the received traffic signals from input 416 s using channel estimates. A demodulated signal is processed in a tap combiner, a deinterleaver, and a channel decoder 420 to generate received S1S2 symbols.

In an alternate embodiment of the common pilot channel for 4 antenna diversity, the common pilot channels are out of phase in the same manner as the traffic channels before transmission. Figure 2 shows a block diagram of fragments of a transmitter 200 STRP with a common pilot channel, corresponding to another variant embodiment of the invention. The transmitter 200 comprises an input 226, a block code processor 224, entrances of the channel processing of the traffic channel symbol stream, antenna gain blocks 204a-204d, phase shifters 206a and 206b, phase shifter inputs 212a and 212b, code multipliers 208a-208d, code multiplier input 210 , pilot sequence processing taps, inputs 214a-214d, antenna gain units 216a-216d, phase shifters 218a-218b, phase shifter control blocks 224a and 224b, code multipliers 220a-220d, code multiplier input 222, RF transmitter 228, containing RF transmitters 228a-228d, and antennas Ant. 1-Ant. 4.

Processing and transmission of the traffic channel in the transmitter 200 is carried out in the same manner as in the transmitter 100 shown in FIG. However, the transmitter 200 uses common phase-shifted pilot channels. The sequence P1 of the common pilot channel is fed to the inputs 214a and 214b of the pilot processing processing tap, and the sequence P2 of the common pilot channel is fed to the inputs 214c and 214d of the pilot processing tap. Then, the pilot sequences are processed separately in the antenna gain units 216a-216d. The pilot sequence P1 output from the antenna gain unit 216a is supplied to a code multiplier 220a. The pilot sequence P2 output from the antenna gain unit 216c is supplied to a code multiplier 220c. The pilot sequence P1 output from the antenna gain unit 216b is supplied to the phase shifter 218a. The pilot sequence P2 output from the antenna gain unit 216d is supplied to the phase shifter 218b. The phase shifter 218a, 218b performs a phase shift under the control of the phase shifter control unit 224a, 224b, respectively. The phase shift can be the same continuous or discrete phase hopping pattern that is used for traffic channels. The phase-shifted pilot sequence P1 output from the phase shifter 218a is supplied to the code multiplier 220b, and the phase-shifted pilot sequence P2 output from the phase shifter 218b is fed to the code multiplier 220d. The encoded pilot sequence P1, output from the code multiplier 220a, enters the RF transmitter 228a for transmission from Ant.1. The coded phase-shifted pilot sequence P1 output from the code multiplier 220b is supplied to the RF transmitter 228b for transmission from Ant.2, the encoded pilot sequence P2 output from the code multiplier 220b is supplied to the RF transmitter 228c for transmission from Ant .3, and the encoded phase-shifted pilot sequence P2 output from the code multiplier 220d is fed to the RF transmitter 228d for transmission from Ant.4.

Phase shifters 218a and 218b may carry out a phase shift in accordance with various alternatives, for example, described for phase shifting carried out according to the embodiment of FIG. 1.

FIG. 5 is a block diagram of fragments of an embodiment of a receiver 500 for use with the transmitter shown in FIG. 2. The receiver 500 comprises an input 502a of the channel estimation processing of channel 1 and channel 2 and an input 502b of the channel estimation processing of channel 3 and channel 4, a channel estimation unit 504, a TDDC demodulator 508, a traffic signal input 510 and a tap combiner, a deinterleaver, and a channel decoder 512.

The received pilot sequence P1 (ch1 + ch2φ), received in channels 1 and 2 with Ant. 1 and Ant. 2, respectively, of transmitter 200, is input 502a. The received pilot sequence P2 (ch3 + ch4φ), received in channels 3 and 4 with Ant.3 and Ant.4, respectively, of the transmitter 200, is input 502b. The channel estimation unit 504 performs channel estimation using, for example, the function of a moving average low-pass filter and provides a combined estimate for channels 1 and 2 (channel estimate 1.2) and a combined estimate for channels 3 and 4 (channel estimate 3 ,four). The channel estimates are fed to the PVRP demodulator 508, which processes the received traffic signals from input 510 using channel estimates. The demodulated signal is processed in block 512 of a tap combiner, a deinterleaver, and a channel decoder to generate received S1S2 symbols. FIG. 6 shows an embodiment of the retraction of the PVRP demodulator 508 shown in FIG. 5, which uses channel estimate 1.2 and channel estimate 3.4 to demodulate received traffic signals.

According to another embodiment of the 4-antenna diversity, dedicated transmitter channels can be implemented in the WDMCR transmitter 150 shown in FIG. 1. Figure 3 shows a block diagram of fragments of a transmitter 300 STRP with a dedicated pilot channel, corresponding to another variant embodiment of the invention. The transmitter 300 includes an input 318, a block code processor 316, channel symbol stream processing taps inputs 302a-302d, antenna gain units 304a-304d, phase shifters 306a and 306b, phase shifter inputs 312a and 312b, code multipliers 308a-308d, code multiplier input 310 and Antennas Ant.1-Ant.4.

The transmitter 300 shown in FIG. 3 is an implementation using dedicated pilot channels transmitted by embedding pilot sequences in a traffic channel symbol stream. The input 318 and the block code processor 316 operate in the same manner as the input 126 and the block code processor 124 shown in FIG. In the transmitter 300, when the symbols S1S2 are input to the tap inputs 302a and 302b for processing the symbol stream, the pilot channel sequence U1 is fed to the inputs 302a and 302b and is multiplexed between the symbol sets S1S2. In addition, -S2 * S1 * is supplied to the tap-off inputs 302c and 302d for processing the symbol stream, and the pilot channel sequence U2 is fed to the inputs 302c and 302d and is multiplexed between the -S2 * S1 * character sets. Another possibility is to set 4 different dedicated pilot sequences, one for each transmit antenna.

The multiplexed symbol streams at inputs 302a-302d are provided to antenna gain units 304a-304d, respectively. In the antenna gain blocks 304a-304d, a channel gain is applied. A stream comprising S1S2 and a pilot sequence U1 is output from the antenna gain unit 304a to a code multiplier 308a. The stream containing S1S2 and the pilot sequence U1 is output from the antenna gain unit 304b to the phase shifter 306a, where it receives a phase shift in accordance with the input signal from the phase shifter control unit 312a, and then it is supplied to the code multiplier 308b. A stream containing -S2 * S1 * and a pilot sequence U2 is output from the antenna gain unit 304c to a code multiplier 308c, and the same stream, -S2 * S1 * and a pilot sequence U2 is output from an antenna gain unit 304d to a phase shifter 306d, where it receives a phase shift in accordance with the input signal from the phase shifter control unit 312b, and then enters the code multiplier 308d. Code multipliers 308a-308d multiply the corresponding stream by a scrambling code. The code-multiplied stream S1S2 and the pilot sequence U1 is supplied to the RF transmitter 314a for transmission from Ant.1. The code-phase-shifted stream S1S2 and the pilot sequence U1 are fed to the RF transmitter 314b for transmission from Ant.2. The code-multiplied stream -S2 * S1 * and the pilot sequences U2 is supplied to the RF transmitter 314c for transmission from Ant.3, and the code-multiplied phase-shifted stream -S2 * S1 * and the pilot sequences U2 is fed to the RF transmitter 314d for transmission from Ant. 4. RF transmitters can perform pulse shaping in the base band, modulate and convert to the carrier frequency. In some implementations, non-zero weighting may optionally be applied after pulse formation in the baseband and modulation.

The receiver shown in FIG. 5 can be modified for use with the transmitter 300 shown in FIG. 3. In this case, the receiver 500 will function similarly, but the inputs 502a and 502b will input U1 (Ch1 + Ch2φ) and U2 (Ch3 + Ch4φ), respectively, to the channel estimator 504c. According to yet another embodiment of 4-antenna diversity, it is possible to implement a combined embodiment of dedicated pilot channels and common pilot channels. On Fig shows a block diagram of fragments of the transmitter 1200 PVRP with a dedicated / common pilot channel, corresponding to another variant embodiment of the invention.

Transmitter 1200 operates in essentially the same way as transmitter 300 shown in FIG. 3, except that common pilot channels are added to Ant.1 and Ant.2. Sequences P1 and P2 of the common pilot channel are fed to the inputs 1218a and 1218b of the allocation of the processing of the pilot sequence, respectively. The pilot sequences are separately processed by antenna gain units 1220a and 1220b and code multipliers 1222a and 1222b. The encoded output signals of the code multipliers 1222a and 1222b are supplied to the RF transmitters 1214a and 1214c, respectively, of the RF transmitter 1214. The RF transmitters can generate pulses in the base band, modulate and convert to the carrier frequency. In some implementations, non-zero weighting can optionally be applied after the formation of modulating pulses and modulation.

The transmitter 1200 shown in FIG. 12 provides common pilot channels without phase change to Ant.1 and Ant.3 and dedicated pilot channels to Ant.1, Ant.2, Ant.3 and Ant.4. Pilot sequences can be multiplexed in one channel slot, for example, in an embodiment where the transmission frame contains 15 channel slots. For common and dedicated control channels, you can set different antenna gains. Antenna gains can also vary over time.

FIG. 13 is a block diagram of fragments of a receiver 1300 for use with the transmitter shown in FIG. 12. The receiver 1300 includes a tap for processing channel 1 and channel 2 having inputs 1302a and 1302b, a tap for processing channel 3 and channel 4 having inputs 1302c and 1302d, phase shifter input 1304, channel estimation unit 1306, PVRD demodulator 1310, traffic signal input 1312 and inverted interleaver and decoder 1314.

The received pilot sequences P1, U1, P2 and U2 are received at the inputs 1302a, 1302b, 1302c and 1302d, respectively, of the receiver 1300. The channel estimator 1306 estimates the channel using, for example, the function of the moving average low-pass filter, and provides a combined estimate for channels 1 and 2 (channel estimate 1.2) 1308a and the combined estimate for channels 3 and 4 (channel estimate 3.4) 1308b. Channel estimates are fed to the PVRP demodulator 1310, which processes the received traffic signals from input 1312 using channel estimates. The demodulated signal is processed in block 1314 of a combiner of taps, a deinterleaver, and a channel decoder to generate received symbols S1, S2.

To control the power, you can use pre-known information about the abrupt phase change. 14 shows fragments of a receiver for evaluating a power control according to an embodiment of the invention. The receiver 1400 comprises a channel estimation unit 1402, channel estimation taps inputs 1404a-1404d, phase shifter inputs 1408a and 1408b, channel estimation phase shifters 1406a, channel estimation outputs 1410a and 1410b, squaring units 1412a and 1412b, and power control processor 1414.

Channel estimator 1402 calculates channel coefficients from common or dedicated channels, for example, from transmitter 1200, for all four antennas during a given channel interval “t”. This may be channel prediction for the channel interval t + 1; alternatively, in channels with slow fading, you can use the channel estimate for the channel interval t. These channel coefficients are denoted by channel sc # 1 (t), channel sc # 2 (t), channel sc # 3 (t) and channel sc # 4 (t) at inputs 1404a-1404d, respectively. For multiple taps, for example, canal sc # 1 (t) is a vector channel estimate corresponding to all taps from Ant.1.

Based on the previously known information about the phase jump at the inputs of the phase shifter and channel estimation information for the current channel interval “t”, the channel coefficients for the channel interval “t + 1” are estimated as follows:

canal sc # 12 (t + 1) = canal sc # 1 (t) + canal sc # 2 (t) e ^{φ12 (t + 1)}

canal sc # 34 (t + 1) = canal sc # 3 (t) + canal sc # 4 (t) e ^{φ34 (t + 1)} , (2)

where φ12, φ34 are known in advance.

Estimation of the received signal power for the channel interval (t + 1) can be made on the basis of channel ots # 12 (t + 1) and channel ots # 34 (t + 1):

Процессор 1414 генерирует команду управления мощностью с использованием канальной оценки принимаемой мощности.

The processor 1414 generates a power control command using a channel estimate of the received power.

The method and apparatus of the invention can also be implemented with diversity in the area of Walsh codes. 7 depicts a block diagram of fragments of a transmitter 700 space-time expansion (TAC), corresponding to a variant embodiment of the invention.

Transmitter 700 is an embodiment of the TAC transmitter 150 of FIG. 1 a, in which the space-time block processor performs the conversion in the Walsh code region. The used matrix of the block TAC code can be represented as:

(3)

(3)

матрицы.Similarly to the embodiment shown in figa, each row of the matrix and its phase-shifted version are transmitted from separate antennas Ant.1-Ant.4. The characters S1 and S2 in each line are transmitted simultaneously for two character periods, and not sequentially. The data symbols are transmitted to the transmitter 700 from the input 718 of the channel encoder 720. The channel encoder 720 encodes, perforates (punctures), interleaves and formats the input data symbols and once again displays the output symbol S1 of the encoder as even data and once again the output symbol S2 of the encoder as odd data. The even data is sent for processing to symbol repetition blocks 702a, b, e, f, Walsh function blocks 704b and 704d, Walsh multipliers 706a, b, e, f, adders 708a-708d and complex adders 710a and 710b. The odd data arrives for processing on the symbol repetition blocks 702c, d, g, h, Walsh function blocks 704b and 704d, Walsh multipliers 706c, d, g, h, adders 708a-708d and complex adders 710a and 710b. The output of complex adder 710b is a string

matrices.

поступает на комплексный умножитель 712а для генерации

и

поступает на комплексный умножитель 712b для генерации

Затем

поступает на ВЧ-передатчик 714а для передачи с Ант.1,

поступает на ВЧ-передатчик 714b для передачи с Ант.2,

поступает на ВЧ-передатчик 714c для передачи с Ант.3, и

поступает на ВЧ-передатчик 714d для передачи с Ант.4.Then

enters the complex multiplier 712a to generate

and

arrives at complex multiplier 712b to generate

Then

arrives at the RF transmitter 714a for transmission from Ant.1,

enters the RF transmitter 714b for transmission from Ant.2,

arrives at the RF transmitter 714c for transmission from Ant.3, and

arrives at the RF transmitter 714d for transmission from Ant.4.

FIG. 9 shows a block diagram of fragments of an embodiment of a receiver 900 for use with a transmitter 700 shown in FIG. 7. Receiver 900 comprises an input 912, Walsh function blocks 902b and 902d, Walsh multipliers 902a and 902c, channel multipliers 904a-904d, complex adders 906a and 906b, multiplexer 908 and output 910. Received input signals are received at input 912 and processed by a TAC demodulator. The pilot channel transmission and channel estimation procedures may be the same as explained for the case of the STP. Blocks 904c and 904b may be the same as 412a, 412b described in FIG. 4 for the case of a common pilot channel without phase change. In the case of common pilot channels or dedicated phase-hopped pilot channels, the transmission of channel estimates can be obtained from the channel estimate unit 504 described with reference to FIG. 5. These channel estimates arrive at the TAC demodulators in FIG. 9 as h1 and h2. h1 corresponds to the combined channel estimate from Ant.1, Ant.2 and h2 corresponds to the combined channel estimate from Ant.3, Ant.4. After TAC demodulation using 902a, b, c, d and 904a, b, c, d and 906a, b, output 908 produces a TAC demodulated signal to be transmitted to block 512 combiner, deinterleaver, and channel decoder specified figure 5.

The proposed invention can also be implemented in an embodiment of the orthogonally diversity transmission (PPR) of the invention. FIG. 8 is a block diagram of fragments of an ODP transmitter 800 according to an embodiment of the invention. The transmitter 800 comprises an input 822, a channel encoder 820, symbol repetition units 802a-802d, Walsh function blocks 804a and 804b, Walsh multipliers 806a-806d, complex adders 808a-808b, complex multipliers 810a and 810b, and RF transmitters 812a-812d. The transmitter is an orthogonal transmit diversity (POD) embodiment of the transmitter 150 shown in FIG. 1 a, in which a space-time block processor transforms into a Walsh code space. The used matrix of the OPC block code can be represented as:

(4)

(four)

, и результат на выходе комплексного сумматора 808b представляет собой

.

поступает на комплексный умножитель 818а для генерации

и

поступает на комплексный умножитель 818b для генерации

Затем

поступает на ВЧ-передатчик 812а для передачи с Ант.1,

поступает на ВЧ-передатчик 812b для передачи с Ант.2,

поступает на ВЧ-передатчик 812c для передачи с Ант.3, и

поступает на ВЧ-передатчик 714d для передачи с Ант.4.As in the embodiment shown in figa, each row of the matrix and its phase-shifted version are transmitted from separate antennas Ant.1-Ant.4. The data symbols are transmitted to the transmitter 800 from the input 822 of the channel encoder 820. The channel encoder 820 encodes, perforates (tingles), interleaves and formats the input data symbols and, once again, outputs the encoder output symbol S1 as even data and, once again, the encoder output symbol S2 as odd data. The even data is supplied for processing to symbol repeat units 802a and 802b, Walsh function block 804a, Walsh multipliers 806a and 806b complex adder 808a. The odd data arrives for processing on the symbol repeat units 802c and 802d, the Walsh function block 804b, the Walsh multipliers 806c and 806d, and the complex adder 808b. The result at the output of complex adder 808a is

, and the result at the output of complex adder 808b is

.

arrives at complex multiplier 818a to generate

and

arrives at complex multiplier 818b to generate

Then

arrives at the RF transmitter 812a for transmission from Ant.1,

arrives at the RF transmitter 812b for transmission from Ant.2,

arrives at the RF transmitter 812c for transmission from Ant.3, and

arrives at the RF transmitter 714d for transmission from Ant.4.

FIG. 10 is a block diagram of fragments of an embodiment of a receiver 1000 for use with a transmitter 800 shown in FIG. The receiver 1000 comprises an input 1010, Walsh function blocks 1002a and 1002b, Walsh multipliers 1010a and 1010b, multipliers 1004a and 1004b, a multiplexer 1006 and an output 1008. The received input signals are received at input 1010 and are processed by the OPC demodulator 1000 based on the known channel coefficients h1 * and h2 *. The channel coefficients h1 and h2 for this PPR block are obtained in the same manner as explained in FIG. 4 and FIG. 5. The ODP demodulator 1000 is implemented using 1010, 1010a, b and 1012a, b and 1004a, b and 1006. The ODP-demodulated output 1008 is supplied to the tap combiner unit 512, the deinterleaver and the channel decoder indicated in FIG. 5.

The embodiment of FIG. 1 may also be implemented in a TDMA transmitter for operation in an EDGE (Electronic Data Acquisition) system. Figure 11 shows a block diagram of fragments of a transmitter of a long block code block according to an embodiment of the invention. The transmitter 1100 comprises an input 1118, 1120, tap-off inputs 1116a-1116d for processing a symbol stream, time reversal blocks 1102 and 1104, complex conjugation blocks 1106a and 1106b, a multiplier 1108, phase multipliers 1110a and 1110b, phase multiplier control blocks 1112a and 1112b and antennas Ant.1, Ant.2, Ant.3 and Ant.4. The channel encoder 1120 encodes, perforates, interleaves, and formats the character stream received at input 1118. The channel encoder 1120 also splits the input character stream into even and odd data streams. The even-numbered data stream enters the tap input 1116a and the RF transmitter 1122a for transmission from Ant.1 during the first half of the data packet, and the odd-numbered data stream goes to the tap input 1116c and the high-frequency transmitter 1122c for transmission from Ant. 2 during the first half data packet. During the second half of the data packet, the even data stream enters the tap input 1116b, undergoes time conversion in the time reversing unit 1102, complex conjugation in the complex conjugation block 1106a, and enters the RF transmitter 1122c for transmission from Ant.3. The odd data stream is fed to the tap input 1116d, subjected to time reversal in the time reversing block 1104, complex conjugation in the complex conjugation block 1106b, multiplied by -1 in the multiplier 1108, and fed to the RF transmitter 1122d for transmission from Ant.4 for the second half the data packet. The training sequence SEQ1 is entered in the middle of the packet transmitted from Ant.1, and the training sequence SEQ2 is entered in the middle of the packet transmitted from Ant.1. Phase multipliers 1112a and 1112b phase out the input signals of the RF transmitters 1122b and 1122d using multiplier units 1110a and 1110b, respectively. The output signal of the phase multiplier 1112a is supplied to the RF transmitter 1122b for transmission from Ant.2, and the output signal of the phase multiplier 1112b is supplied to the RF transmitter 1122d for transmission from Ant.4. High-frequency transmitters can carry out the formation of pulses of the base band, modulation and conversion to the carrier frequency. In some implementations, phase multiplication can optionally be applied after the steps of generating baseband pulses and modulation.

The phase shift used in phase multipliers 1122a and 1122b remains constant over the length of the packet, but the phase changes from packet to packet. The phase may change periodically or randomly based on the MPSK diagram, as explained above. According to a preferred embodiment, the phase shift in Ant. 4 remains equal to the phase shift in Ant. 2 with a 180 degree shift or multiplication by -1. Phase multiplication can be performed before or after the formation of the pulses of the base band. As an alternative to the embodiment of FIG. 11, transmission from Ant. 1 and Ant. 3 may be alternated.

In EDGE, a slightly modified transmitter shown in FIG. 3 can also be used. With respect to EDGE, the spatio-temporal block indicated by 316 is not used character-by-character, but block by block. The length of the block can be chosen equal to the first half of the packet. In EDGE, the length of the first half and second half of the packets is 58 characters. In this case, S1 and S2 denote symbol blocks, and () * denotes the time reversal of the symbol block and the complex conjugation operation. S1 * denotes a time reversed and complex conjugate block of characters S1. -S2 * denotes time-reversed, complex conjugate and multiplied by -1 block of S2 characters. Two training sequences can be selected as pilot sequences U1 and U2, for example, well-known CAZAC sequences. Extension codes 308a, b, c, d are not used in EGDE. The phase multiplication blocks 306a and 306b are stored.

A receiver designed for a 2-antenna space-time block code can be used as a receiver for the embodiments shown in FIG. 1 or FIG. 2.

From the above descriptions and embodiments, those skilled in the art can understand that, although the method and apparatus of the present invention are illustrated and described with respect to specific embodiments, the described embodiments are subject to numerous modifications and replacements, and numerous other embodiments of the invention can be implemented. without changing the nature and scope of the invention defined by the claims.

Claims (20)

1. The method of transmitting a signal from multiple antennas, namely, that receive a stream of characters in the transmitter, converting the input symbol stream to generate a conversion result containing an orthogonal space-time block code N × N ', and generating N first signals, where N is the number of transmission paths, N' is the number of output symbols per transmission path, carry out non-zero complex time-weighting of at least one of the N first signals of the conversion result to generate at least one second signal, each of the at least one second signal being phase shifted relative to one of the N first signals, from which it is formed, wherein the non-zero complex weighting includes a phase shift of at least one of the N first signals by means of at least the first predetermined sequence of step change in zy, and the weighting coefficients of the jerky change for a given sequence of jerky phase changes are obtained from the phase shift keying diagram (QPS), which has 8 states, and the given sequence of jerky phase changes in degrees is (0, 135, 270, 45, 180, 315, 90 225), and each of the N first signals of the conversion result from the first at least one antenna and essentially each of the at least one second signal from the second at least one antenna are transmitted substantially simultaneously; N utye first signals and said at least one second signal together form M signals, where M is greater than N. 2. The method according to claim 1, characterized in that at least one second signal that is phase shifted relative to one of the N first signals from which it was generated at the stage of non-zero complex weighing, contains the first of the second signal and, according to at least the second of the second signal, and at least the first predetermined phase jump sequence includes the first predetermined phase jump sequence in degrees (0, 135, 270, 45, 180, 315, 90, 225) and, at least the second preset is followed the abruptness of the phase jump, and the second predetermined sequence of the phase jump in degrees is (180, 315.90, 225, 0, 135, 270, 45). 3. The method according to claim 2, characterized in that the first of the second signal is phase shifted by means of the first predetermined phase jump sequence, and the second of the second signal is phase shifted by the second predetermined phase jump sequence. 4. The method according to claim 3, characterized in that the phase shift jumps of the first predetermined phase jump sequence and the second predetermined phase jump sequence, whereby the first and second signals of said second signals are respectively phase shifted at the non-zero complex weighting step, followed by consecutive selected intervals. 5. The method according to claim 4, characterized in that the successive selected intervals, followed by jumps in phase shifts of the first and second predetermined sequences of abrupt phase changes, phase shifting the first and second of the second signals, contain periodic intervals. 6. The method according to claim 3, characterized in that the jumps in phase shifts of the first predetermined sequence of abrupt phase change and the second predetermined sequence of abrupt phase change, by which the first and second signals from the second signals are respectively phase shifted at the stage of non-zero complex weighing, follow at random intervals. 7. The method according to claim 1, characterized in that jumps in phase shifts of a given sequence of abrupt phase change, by means of which at least one of the second signals is phase shifted at the stage of non-zero complex weighing, are followed at successive selected intervals. 8. The method according to claim 7, characterized in that the successive selected intervals, followed by jumps in phase shifts of a given sequence of phase-hopping phase shifting, at least one second signal in phase, comprise periodic intervals. 9. The method according to claim 1, characterized in that jumps in phase shifts of a given sequence of abrupt phase change, by means of which at least one second signal is phase shifted at the stage of non-zero complex weighing, are followed at random intervals. 10. A device for transmitting a signal containing a processor for converting an input symbol stream to generate a transform result containing an orthogonal space-time block code N × N ′ and generating N first signals, where N is the number of transmission paths, N ′ is the number of output symbols per transmission path, at least one weighing device for non-zero complex time-weighting of at least one of the N first signals of the conversion result to generate at least one second signal, each of at least one second weighted signal being shifted in phase relative to one of the N first signals from which it is generated, wherein the non-zero complex weighting includes a phase shift of at least one of the N first signals in accordance with at least the first task the sequence of the phase jump, and the weighting coefficients of the phase jump for the phase jump sequence are obtained from the phase shift keying diagram (PSK) having 8 states, and the given sequence of the phase jump in degrees is (0, 135, 270, 45, 180 , 315, 90, 225), and a transmitter for transmitting substantially simultaneously each of the N first signals of the conversion result from the first at least one antenna and each of the at least one second signal from the second at least one antenna, said N first signals and said at least one second signal together form M signals, where M is greater than N. 11. The device according to claim 10, characterized in that at least one second signal, phase-shifted relative to one of the N first signals, from which it was generated by at least one weighing device, comprises the first of the second a signal and at least a second of a second signal, wherein at least a first predetermined phase jump sequence includes a first predetermined phase jump sequence in degrees (0, 135, 270, 45, 180, 315, 90, 225) and at least a second adannuyu hopping sequence phase change, wherein the second predetermined hopping sequence in degrees phase change is (180, 315, 90, 225, 0, 135, 270, 45). 12. The device according to claim 11, characterized in that the first of the second signal is phase shifted by means of a first predetermined phase jump sequence, and the second of the second signal is phase shifted by a second predetermined phase jump sequence. 13. The device according to p. 12, characterized in that the jumps of the phase shift of the first predetermined sequence of abrupt phase change and the second predetermined sequence of abrupt phase change, using which the first and second of the second signals are respectively phase shifted by at least one device weighings are followed at successive intervals selected. 14. The device according to item 13, wherein the successive selected intervals, followed by jumps in phase shifts of the first and second predetermined sequences of abrupt phase changes, phase-shifting the first and second of the second signals, contain periodic intervals. 15. The device according to p. 12, characterized in that the jumps of phase shifts of the first predetermined sequence of abrupt phase change and the second predetermined sequence of abrupt phase change, using which the first and second of the second signals are respectively phase shifted by at least one device weighing, followed at random intervals. 16. The device according to claim 11, characterized in that at least one weighing device comprises a first weighing device for non-zero complex weighing of the first of the first of N first signals generated by the processor, and at least a second weighing device for non-zero complex weighting of the second of the first of the N first signals generated by the processor. 17. The device according to clause 16, wherein the first weighing device phase outsets the first of the first signal through the first predetermined phase jump sequence, and the second weighing device phase outs the second of the first signal through the second predetermined phase jump sequence. 18. The device according to claim 10, characterized in that jumps in phase shifts of a given sequence of abrupt phase change, using which at least one second signal is phase shifted by means of at least one weighing device, are followed at successive intervals . 19. The device according to p. 18, characterized in that the successive selected intervals at which the phase jumps of a given sequence of a phase-wise phase-shifting at least one second signal are repeated in sequence contain periodic intervals. 20. The device according to claim 10, characterized in that jumps in phase shifts of a given sequence of abrupt phase change, with which at least one second of the second signal is phase shifted by means of at least one weighing device, followed by random at intervals.

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