MXPA03008030A

MXPA03008030A - Non-zero complex weighted space-time code for multiple antenna transmission.

Info

Publication number: MXPA03008030A
Application number: MXPA03008030A
Authority: MX
Inventors: Kuchi Kiran
Original assignee: Nokia Corp
Priority date: 2001-03-28
Filing date: 2002-03-26
Publication date: 2005-04-29
Also published as: WO2002080375A2; BR0208208A; CN100536450C; AU2002251395A1; CA2440033C; WO2002080375A3; EP1512256A2; RU2003131399A; CA2440033A1; CN1611047A; EP1512256A4; JP2005503045A; RU2276463C2

Abstract

The present invention presents a method and apparatus for phase hopping and space-time coding signals for transmission on multiple antennas (160,162,164,166). The method and apparatus provides expansion of a N x N space time block code to a M x M space time block code, where Mgt;N, by using phase hopping on the symbols within the N x N space time block code to allow transmission of the space time block code on a number of diversity antennas greater than N . A result of M antenna diversity may be achieved for M transmit antennas.

Description

SPACE CODE-COMPLEX WEIGHT NOT NULL FOR TRANSMISSION IN MULTIPLE ANTENNAS ' FIELD OF THE INVENTION The present invention relates to a method and apparatus for achieving transmission diversity in telecommunications systems and, more particularly, to a method and apparatus for space-time encoding signals of non-zero complex weighting for transmission in multiple antennas BACKGROUND OF THE INVENTION As wireless communication systems evolve, the design of wireless systems has become increasingly demanding in relation to equipment and performance requirements. Future wireless systems, which will be third and fourth generation systems compared to first generation analog and second generation digital systems currently in use, will be required to provide high-speed, high-speed data services in addition to services of high quality voice. There will be equipment design constraints along with the performance requirements of the system service, which will strongly impact the design of mobile terminals. The third and fourth generation wireless mobile terminals will need to be smaller, lighter, more energy efficient units that are also more capable of providing sophisticated voice and data services of these future wireless systems. The multi-trajectory attenuation of variation in time is an effect in wireless systems whereby a transmitted signal propagates along multiple trajectories to a receiver causing attenuation of the received signal due to the constructive and destructive sum of the signals in the receptor. Several methods for overcoming the effects of multipath attenuation are known, such as time intercalation with error correction code, implementation of frequency diversity using spread spectrum techniques, or transmitter power control techniques. However, each of these techniques has disadvantages in relation to use for third and fourth generation wireless systems. Time interleaving can introduce unnecessary delay, spread spectrum techniques may require a large allocation of bandwidth to overcome a high coherence bandwidth, and power control techniques may require more transmitter power than desirable for receiver-to-transmitter feedback techniques that increase the complexity of the mobile terminal. All these disadvantages have a negative impact to achieve the desired characteristics for third and fourth generation mobile terminals.

Antenna diversity is another technique to overcome the effects of multiple trajectory attenuation in wireless systems. In diversity reception, two or more physically separate antennas are used to receive a transmitted signal, which is then processed by combination and switching to generate a received signal. A disadvantage of the reception of diversity is that the required physical separation between antennas can make the reception of diversity impractical to be used in the future link in the new wireless systems where a small size of mobile terminal is desired. A second technique for implementing antenna diversity is transmission diversity. In transmission diversity, a signal is transmitted from two antennas and then processed in the receiver using, for example, a maximum probability sequence estimator (MLSE), average least squares error receivers (MMSE). , by its acronym in English), Maximum receivers a posteriori, and their approximations. The transmission diversity has a more practical application for the future link in wireless systems because it is easier to implement multiple antennas in the base station than in the mobile terminal. The transmission diversity for the case of two antennas is well studied. Alamouti has proposed a transmission diversity method for two antennas that offer second order diversity for complex value signals. S. Alamouti, nA Simple Transmit Diversity 'Technique for Wireless Communications, "IEEE Journal on Selected Areas of Communications, pages 1451-1458, October 1998. The Alamouti method comprises transmission simultaneously of two signals of two antennas during a period of symbols During a period of symbols, the signal transmitted from a first antenna is denoted by SO and the signal transmitted from the second antenna is denoted by YES During the next period of symbols, the signal -SI * is transmitted from the first antenna and the signal SO * is transmitted from the second antenna, where * is the conjugate complex operator.A similar system of transmission diversity can also be made in the code domain. For example, two copies of the same symbol can be transmitted in parallel using orthogonal Walsh codes. build a space-frequency coding method. The extension of the Alamouti method to more than two antennas is not direct. Tarokh and colleagues have proposed a method that uses Space Block codes Speed Time = 1/2, and 3/4 on three of four antennas that employ complex signal constellations. V. Tarokh, H.

Jafarkhani, and A. Calderbank, "Space-Time Block Codes from Orthogonal Designs (Space-Time Block Codes from Orthogonal Designs)", IEEE Transactions on Information Theory, pages 1456-1467, July 1999. This method has the disadvantage in a loss in transmission speed and in the fact that the multi-level nature of the ST-coded ols increases the maximum to average ratio requirement of the transmitted signal and imposes constraints on the design of the power amplifier linear. Additional techniques mitigating these problems have been proposed in 0. Tirkkonen and A. Hottinen, uComplex space-time block codes for four Tx antennas (complex space-time block codes for four Tx antennas), "Memories of the Globecom 2000, November 2000, San Francisco, United States of America Other proposed methods include a four orthogonal transmission diversity antenna (OTD) method + spacetime diversity scheme (STTD). English), speed = 1. L. Jalloul, K. Rohani, K. Kuchi, and J. Chen, "Performance Analysis of CDMA Transmit Diversity Methods", Memoirs of the IEEE Vehicular Tecnology Conference, Autumn 1999, and M. Harrison, K. Kuchi, "Open and Closed Loop Transmit Diversity at High Data Rates on 2 and 4 Elemente (Diversity of Open Circuit Transmission and Cerra to Altzas Speeds in 2 and 4 Elements), "Contribution from Motorola to 3GPP-C30-19990817-017. This method requires an external code and offers second order diversity to the block of STTD (Block of Alamouti) and a gain of intercalation of second order of the use of the block OTD. The performance of this method depends on the strength of the external code. Since this method requires an external code, it is not applicable to non-encoded systems. For the case of the convolutional velocity code = 1/3, the performance of the methods, OTD + STTD method and the method of space block codes Velocity time = 3/4 of Tarokh, are approximately equal. Another method of speed 1 is proposed by O. Tirkkonen, A. Boariu, and A. Hottinen, "Minimal non-orthogonality rate 1 space-time block code for 3+ Tx antennas (speed-space block code 1) non-orthogonal minimum for more than 3 Tx antennas), "in reports of ISSSTA 2000, September 2000. The method proposed in this publication achieves high performance but requires a complex receiver. It would therefore be advantageous to have a method and apparatus that will provide the advantage of transmission diversity in more than two antennas and at the same time without increasing the complexity of the system design.

BRIEF DESCRIPTION OF THE INVENTION The present invention presents a method and apparatus for coding signals of non-zero complex weighting and of space-time for transmission in multiple antennas. The method and apparatus provides expansion of a space-time block code N x N ', where N is the number of transmission paths and N' is the number of output ols per transmission path, to a block code of space time M x M ', where M > N, generated by using complex repetition weighting and not null of the ols in the space time block code N x N ', to allow the transmission of the space-time block code in an M number of diversity transmission paths. The diversity transmission paths may comprise separate antennas or beams. The temporal length of the largest M 'code can be equal to the temporal length of the original code, N'. In the method and apparatus, a transform is effected in a stream of input ols, to generate a transform result comprising a space-time block code. The N output currents of the space-time block code, each consisting of N 'output symbols, are subsequently repeated and at least one of the repeated currents is weighted by non-zero complexes over time to generate M currents of N' output symbols for transmission in diversity transmission paths M. Complex non-zero weighting can include phase changes. In one embodiment, N is at least 2 and M is at least 3. At least two of N currents of N 'output symbols, corresponding to the original N currents of N' output symbols, are then transmitted each in a first at least one antenna and at least one of the current MN weighted by non-zero complexes of N 'symbols is transmitted in one of a second at least one antenna. The first at least one antenna and the second at least one antenna can comprise any of M antennas. In another embodiment, the method and apparatus can be implemented in a transmitter having common or dedicated pilot channels that allow efficient channel estimation or the coefficients that are required to decode the spacetime code. In this mode the common and dedicated pilot channels can be implemented alone or both together in the transmitter. In an alternative modality of this modality, training symbols are transmitted in N transmission diversity trajectories, making it possible to estimate the N independent diversity transmission trajectories. For this, a dedicated pilot channel code sequence can be multiplexed in each of the N currents of N 'output symbols of the original space-time block code, to generate N currents of N' output symbols and channel sequence pilot. Then, the repetition and the non-zero complex weighting can be applied to generate M changed phase currents of N 'symbols and sequences of pilot channels. At least two of N original streams of N 'output symbols and sequences of pilot channels are then transmitted on one of the first at least one antenna and at least one of the MN complex weighing currents of N' output symbols and sequences of Pilot channels are transmitted in one of the second at least one antenna. Another way to allow the estimation of N channels is to transmit common pilot channels in such a way that N common pilot channels are transmitted in each of the first at least one antenna, and MN complex weighting copies of some of the N common pilot channels are transmitted in each of the second at least one antenna. The complex weighting factors used for the common channels in each of the second at least one antenna are the same as those used to construct the additional complex weighting MNs of N 'output symbols of the original N currents of N' symbols of exit. In these embodiments, the receiver may or may not know the method used to expand the space-time block code of N x N 'to a space-time block code of M x N', and the temporary weighting sequences employed .

In other embodiments, where N is at least 2 and M can be at least 3, the pilot channels can be arranged to allow the estimation of at least N + l diversity transmission paths. At least one of the N currents of N 'output symbols, corresponding to the original N currents of N' output symbols, are each transmitted on a first at least one antenna and at least one of the MN complex weighing currents of N 'symbols is transmitted each in one of a second at least one antenna. Different common pilot channels are transmitted in each of the first at least one antenna and in at least one of the second at least one antenna. In these embodiments, the receiver needs at least partial knowledge of the method used to expand the time-space block code of N x N 'to a space-time block code of M x N', and the time-weighting sequences employed. The complex weighting in the various modalities can be applied by applying a periodic or random complex weighting pattern to each of the symbol currents that are weighted by complexes. The relationship between the complex weighting of the symbol streams transmitted on the various antennas may also be predefined. BRIEF DESCRIPTION OF THE FIGURES The figure shows a block diagram of a transmitter according to an embodiment of the invention.

Figure Ib shows a block diagram of portions of a STTD transmitter of common pilot channels according to an embodiment of the invention; Figure 2 shows a block diagram of portions of a STTD transmitter of common pilot channels according to another embodiment of the invention; Figure 3 shows a block diagram of portions of a dedicated pilot channel STTD transmitter according to a further embodiment of the invention; Figure 4 shows a block diagram of a mode of a receiver for use with the transmitter of Figure 1; Figure 5 shows a block diagram of portions of a mode of a receiver for use with the transmitter of Figure 2 or with the transmitter of Figure 3; Figure 6 shows the rake receiver finger mode of the STTD demodulator 508 of Figure 5; Figure 7 shows a block diagram of portions of an STS transmitter according to an embodiment of the invention; Figure 8 shows a block diagram of portions of an OTD transmitter according to an embodiment of the invention; Figure 9 shows a block diagram of portions of a mode of a receiver for use with the transmitter of Figure 7; Fig. 10 shows a block diagram of portions of a receiver mode for use with the transmitter of Fig. 8; Figure 11 shows a block diagram of portions of an ST block code transmitter according to an embodiment of the invention; Figure 12 shows a block diagram of portions of a common / dedicated pilot channel STTD transmitter according to another embodiment of the invention; Figure 13 shows a block diagram of portions of a receiver for use with the transmitter of Figure 12; and Fig. 14 shows a block diagram of portions of a receiver for use in the transmitter power control of Fig. 12. Fig. 15 shows a constellation defining a phase change pattern that can be used in various modes of the invention. DETAILED DESCRIPTION OF THE INVENTION Referring now to FIG. 1, a block diagram of a transmitter 150 according to an embodiment of the invention is shown. The transmitter 150 includes an input 152 for receiving an input symbol stream, a block code processor 154 for effecting a transform in the stream of input symbols to generate a transform result which can be represented by a block code of orthogonal time space and extracting 2 streams of symbols from the result of the transform, a non-zero or non-zero weighting factor 156 for performing the non-zero complex weighting of a first of two symbol streams, a non-zero complex weigher 158 for the non-zero complex weighting of the second of the two symbol streams, an RF transmitter 160 for transmitting the first symbol stream in Ant.l, an RF transmitter 162 for transmitting the complex non-zero weighted current of symbols in Ant. 2, an RF transmitter 164 for transmitting the second stream of symbols in Ant. 3, and an RF transmitter 166 for transmitting the second stream of phase change symbols in Ant. 4. Antennas Ant. 1 - Ant. 4 they may be polarized one relative to the other to provide improved diversity reception. For example, Ant. 1 or Ant. 2 may be vertically polarized in relation to a horizontal polarization of Ant. 3 or Ant. 4, respectively. The mode of the transmitter 150 of the figure may be implemented in various ways suitable for different technologies and systems to expand a block code of 2 x N 'for transmission through 4 transmission diversity paths. In the transmitter 150, each of the transmission diversity paths includes a separate antenna, Ant. 1 - Ant. 4. This may include multiple code division (CDMA) access systems, time division multiple access (TDMA), or any other type of digital communications systems whose transmission diversity can be introduced. In an alternative to the embodiment of FIG. 1a, the non-zero complex weighting can be performed completely on the selected transmission paths to create relative phase shifts between the transmissions in Ant. 1 and Ant. 2 or in Ant. 3 and Ant. 4. For example, the non-zero complex weighting could also be applied before the inputs to the RF transmitters 160 and 164, creating a non-zero complex weighted version of each of the symbol currents , but maintaining a relative phase change between the transmitted signals. An alternative mode of the transmitter 150 can be implemented using less than 4 antennas, to implement the 4 diversity paths. As an example, the input signals to the RF transmitters 164 or 166 can be connected to each other and transmitted in a single antenna. Other alternatives are also possible in which less than 4 diversity trajectories are used, for example, only one of the 2 data streams can be weighted by non-zero complexes and transmitted in two diversity paths. In an alternative embodiment of FIG. 1a, the non-zero complex weighting can be performed after the RF transmitter blocks 160, 162, 164, 166, ie, the non-zero complex weighting could be implemented as a continuous phase sweep after the modulation, and baseband filtering of space-coded symbols. The complex weighting is not null for these transmissions in Ant. 2 and Ant. 4 can be made according to several alternatives. For example, a phase pattern Wi (t) = exp (j * pi * phase_in_gradients / 180) used in Ant. 2 can be applied and the phase pattern -Wi (t), which is 180, can be used in Ant. degrees out of phase with W ^ t). Examples of this would be a phase pattern of changes in degrees of. { 0, 90, 180, 270.}. in Ant. 2 and. { 180, 270, 0, 90.}. in Ant. 4 for the 4PSK constellation. Other example patterns are. { O, 45, 90, 135, 180, 225, 270, 315.}. for 8PSK and. { 0, 22.5, 45, 67.5, 337.5} for 16PSK. Figure 15 shows a constellation defining another phase change pattern that can be used in various embodiments of the invention. This sequence of changes in degrees of. { 0, 135, 270, 45, 180, 315, 90, 225.}. can be transmitted on antenna 2 while using the pattern of changes in degrees of. { 180, 315, 90, 225, 0, 135, 270, 45.}. in the antenna. The phase change can be periodic or random. The periodic phase change refers to a predefined phase pattern for example, the complex weighting l (t) repeated periodically. The complex weighting can be defined in such a way that the sequence of the complex weights defines a path. of maximum length, to make the successive samples of the effective channel as independent as possible. This can do the redundant interleaving and thus allow a low delay transmission. The pseudo-random phase change used may be a sequence of random phase selections of an MPSK constellation. Alternatively, another non-zero complex weighting scheme in which the phase difference between successive phase states is as small as possible is advantageous when estimating channel coefficients or the metric related to the power control of a channel with complex weighting not null In this case, the phase states can still cover 360 degrees for the duration of an encoder block. The intercalation of channels can be used in the modalities as in the conventional systems. It is also possible to implement the non-zero complex weighting sequence and the interleaver together, so that the symbols at the output of the interleaver are as independent as possible. In addition, by changing the relative phase between antennas 1 and 2, and 3 and 4, respectively, the method can be implemented in such a way that there is a change or phase sweep in all the antenna elements, but it is maintained, the change of relative phase between antennas 1 and 2, and 3 and 4. As an example, with the phase sweep, you can have a phase sweep of 50 Hz on antenna 1 and a phase sweep of -50 Hz on the antenna 2, in order to implement an effective sweep of 100 Hz. Similarly for antennas 3 and 4. Phase rotation can change every T seconds. The choice of T depends on the total time duration of the data symbols and the method used to estimate the channel coefficients. The phase may remain constant for the total duration of time occupied by the data symbols in at least one space-time coding block and the dedicated or common training sequence / pilot sequence may be used to allow an appropriate channel estimation. The pilot sequence could be a Walsh code, such as that used in CDMA systems, or a sequence of training symbols with good correlation properties used for the estimation of channels in the TDMA .. Pilot symbols can apply the same complex weighting complexes not null as data in the space-time block. Alternatively, pilots can be transmitted without variation by jumps. In the case of the effective channel for the data, a jump pattern known a priori can be derived together and the channel estimate can be obtained from a channel without variation by jumps. In cases where a non-zero complex weighting is applied to common pilots, the same or a different phase pattern can be applied to both data and common pilots. The estimation of channels that uses pilot sequences without variation of jumps or of training (already transmitted in common or dedicated channels) provides better estimates of channels when the channel is more stationary. Referring now to Figure Ib, there is a block diagram of portions of a common pilot channel transmitter 100 of space-time transmission (STTD) transmitter 100 according to one embodiment of the invention. Transmitter 100 can operate as a 4-antenna transmission diversity extension to release 99 of the third-generation broadband CDMA (CDMA) system standard. The transmitter 100 comprises an input 126, a block code processor 124, traffic stream symbol stream derivation inputs 102a-102d, antenna gain blocks 104a-104d, phase changers 106e and 106b, phase exchanger inputs 112a and 112b, code multipliers 108a-108d, pilot sequence processing derivation inputs 114a-114d, antenna gain blocks 116a-116d, code multipliers 118a-118d, an RF transmitter 128 , including RF transmitters 128a -128d, and antennas Ant. 1 - Ant. 4. In Fig. Ib, the data to be transmitted including a coded channel and the interleaved stream of input symbols X (t) comprising the symbols S1S2 are received at the input 126. The block code processor 124 performs a transform in every two received symbols S1S2 to generate a transform result comprising an orthogonal space-time block code of 2 x 2. In the mode, the block code processor 124 can perform an Alamouti transform to generate the block code in the form represented by the following matrix: S1 S2 The matrix is then divided into four streams of two symbols with each of the currents being input to one of the traffic stream derivation inputs of traffic channel symbols 102a-102d. As shown in Fig. 1, current S1S2 is input to 102a, S1S2 is input to 102b, -S2 * S1 * is input to 102c, and for -S2 * S1 * is input to 102d. The non-zero complex weighting is performed by means of the antenna gain blocks 104a-I04d and the phase changers 106a and 106b. The antenna gain for each of the processing branches is adjusted in the gain blocks 104a-104d. After the antenna gain is adjusted, the phase changers 106a and 106b apply a phase change to the output of the current S1S2 of the antenna gain block 104b and the output of the current -S2 * S1 * of the block of antenna gain 104d. The phase change control blocks 112a and 112b can control the phase changers 106a and 106b by causing a change using a continuous or discrete phase shift variation pattern. A CDMA mixing code is then input to the code multipliers 108a-I08d to generate the current S1S2 to the RF transmitter 128a for transmission in Ant. 1, S1S2 (ep (j »kl)) to the RF transmitter 128b for transmission in Ant. 2, -S1 * S2 * to the RF transmitter 128c for transmission in Ant. 3 and -S2 * S1 * (exp (j <.}. > k2)) to the RF transmitter 128d for transmission in Ant 4. The RF transmitters can perform baseband pulse shaping, modulation, and bearer conversion In some implementations you can choose to apply the variation by phase jumps or scanning after the steps of shaping and modulating baseband pulses The common pilot channel sequences XI-X4 are input to the pilot sequence processing derivation inputs 114a-114d. then separately by means of antenna gain blocks 116a-116d, and code multipliers 118a-118d.The encoded outputs of the code multipliers 118a-118d are then input to the transmitters 128a-128d, respectively, of the t RF transmitter 130.

The pilot sequence XI is then introduced in Ant. 1, the pilot sequence X2 is transmitted in Ant. 2 ', the pilot sequence X3 is transmitted in Ant. 3, and the pilot sequence X4 is transmitted in Ant. 4. Referring now to Figure 4, there is a block diagram of portions of a receiver for use with the transmitter 100 of Figure Ib. Figure 4 shows the signal processing for a rake finger receiving section of a receiver. The received pilot sequences XI-X4 transmitted from the transmitter 100 are received and input to the channel estimation processing branch 402a-402d, respectively. The channel estimator 404 then executes a channel estimation function, for example an average low pass filter movement function, for each of channel 1 - channel 4. The estimates of channel 1 - channel 4 are then extracted from the outputs 406a-406d to adder 410a, phase changer 408a, adder 410b and phase changer 408b. The phase changer 408a receives an input from the phase change control block 414a and changes the estimate for channel 2 by the same phase change used in the transmission of the traffic channel symbols S1S2 of Ant. 2 in the transmitter 100. Phase changer 408b receives an input from the control block of phase changer 414b, changes the estimate for channel 4 by the same phase change used in the traffic channel symbols -S2 * S1 * transmitted from Ant. 4 at transmitter 100. The changed phase version of the estimate for channel 2 is combined with the estimate for channel 1 by adder 410a, and the changed phase version of the estimate for the channel is combined with the estimate for channel 3 in adder 410b. The combined estimate for channels 1 and 2 (412a) and the combined estimate for channels 3 and 4 (412b) are then input to the STTD 418 demodulator, which processes the traffic signals received from input 416 using the channel estimates. The demodulated signal is then processed in a rake combiner, deinterleaver and channel 'decoder 420 to generate the received symbols S1S2. In an alternative mode of common pilot channels for a variety of 4 antennas, the phase of common pilot channels is changed in the same way as for traffic channels before transmission. Referring now to Figure 2, there is a block diagram of portions of a STTD transmitter 200 of common pilot channels according to another embodiment of the invention. The transmitter 200 comprises an input 226, a block code processor 224, traffic channel symbol stream derivation inputs 202a-202d, antenna gain blocks 204a-204d, phase changers 206a and 206b, inputs of phase changers 212a and 212b, code multipliers 208a-208d, input of code multipliers 210, pilot sequence processing derivation inputs 214a-214d, antenna gain blocks 216a-216d, phase changers 218a and 218b, phase change control blocks 224a and 224b, code multipliers 220a-220d, code multipliers input 222, an RF transmitter 228, including RF transmitters 228a-228d, and antennas Ant. 1 and Ant. 4. The processing of traffic channels and transmission in the transmitter 200 is performed in the same way as that used for the processing of traffic channels in the transmitter 100 of FIG. 1. However, the transmitter 200 uses common pilot channels, which change phase. The sequence of common pilot channels Pl is input to the pilot sequence processing derivation inputs 214a and 214b and the sequence of common pilot channels P2 is input to the pilot sequence processing derivation inputs 214c and 214d. Then the pilot sequences are processed separately by means of the antenna gain blocks 216a-216d. The output of pilot sequences Pl of the antenna gain block 216a is input to the code multiplier 220a. The output of pilot sequences P2 of the antenna gain block 216c is input to the code multiplier 220c. The output of pilot sequences Pl of the antenna gain block 216b is input to the phase changer 218a. The output of * iloto sequences P2 of the antenna gain block 216d is input to the phase changer 218b. The phase changer 218a and 218b apply a phase change under the control of the phase changer control block 224a and 224b, respectively. The phase change can be the same continuous or discrete skip variation pattern used for the traffic channels. The output of phase changed pilot sequences Pl of the phase changer 218a is then input to the code multiplier 220b and the output of changed phase pilot sequences P2 of the phase changer 218b is then input to the code multiplier 22Od. The output of coded pilot sequences Pl of the code multiplier 220a is then input to the transmitter of F 228a for transmission in the Ant. 1. The output of co-changed phase-shifted pilot sequences Pl of the code multiplier 220b is input to the RF transmitter 228b for transmission in Ant. 2, the output of coded pilot sequences P2 of the code multiplier 220c is input to the RF transmitter 228c for transmission in Ant. 3, and the sequence output 'coded phase-shifted pilot P2 of the code multiplier 22Od. the RF transmitter 228d is introduced for transmission in Ant. 4. The phase change made by the phase changers 218a and 218b may be in accordance with several alternatives, for example, as described for the phase change made in the modality of FIG. 1. Referring now to FIG. 5, there is a block diagram of portions of a mode of a receiver 500 for use with the transmitter. e Figure 2. The receiver 500 comprises a derivation input 502a of estimation processing of channel 1 and channel 2 and a derivation input 502b of estimation processing of channel 3 and channel 4, a channel estimator 504, a STTD demodulator 508, a traffic signal input 510, and a rake receiver combiner, a deinterleaver and a channel decoder 512. The received pilot sequence Pl (chl + ch.20) received on channels 1 and 2 of Ant. 1 and Ant. 2, respectively, of the transmitter 200 is input to the input 502a. The received pilot sequence P2 (ch3 + ch40) received on channels 3 and 4 of Ant. 3 and Ant. 4, respectively, of the transmitter 200 is input to the input 502b. The channel estimator 504 performs an estimation of the channel using, for example, a moving average filter function, and combined estimates of outputs for channels 1 and 2 ("chest 1.2") (506a), and an estimate combined for channels 3 and 4 ("chest 3,4") (506b). The channel estimates are then input to a STTD demodulator 508, which processes the traffic signals received from the input 510 using the channel estimates. The demodulated signal is then processed in a combined rake, de-interleaver and channel decoder 512 to generate the received symbols S1S2. Figure 6 shows a rake receiver finger mode of the STTD demodulator 508 of Figure 5 which uses chestl, 2, and chest3,4 to demodulate the received traffic signals. In another modality for the diversity of 4 antennas, dedicated pilot channels can be implemented in a WCDMA version. of the transmitter 150 of Figure 1. Referring now to Figure 3, a block diagram of portions of a STTD transmitter 300 of dedicated pilot channels is shown, according to a further embodiment of the invention the transmitter 300 comprises an input 318, a block code processor 316, channel symbol stream processing derivation inputs 302a -302d, antenna gain blocks 304a-304d, phase changers 306a and 306b, phase change inputs 312a and 312b, code multipliers 308a-308d, code multiplier input 310, and antennas Ant. 1-Ant. 4. Transmitter 300 of FIG. 3 is an implementation using dedicated pilot channels that are transmitted by means of pilot sequences integrated in the stream of traffic channel symbols. The input 318 and the block code processor 316 operate in the same manner as the input 126 and the block code processor 124 of FIG. 1. In the transmitter 300, when the symbols S1S2 are input to the processing derivation inputs of symbol streams 302a and 302b, the sequence of pilot channels Ul is input to the multiplexed inputs 302a and 302b between the symbol sets S1S2. Also, -S2 * S1 is input to the symbol stream processing derivation inputs 302c and 302d, and the sequence of pilot channels U2 is input to the inputs 302c and 302d, and multiplexed between the symbol sets of -S2. * S1 *. Another possibility is to define 4 different dedicated pilot sequences, one for each transmission antenna. The multiplexed symbol streams at the inputs 302a-302d are then input to the antenna gain blocks 304a-304d, respectively. The channel gain is applied in the gain blocks 304a-304d. The stream comprising S1S2 and the pilot sequence Ul leaves the antenna gain block 304a towards the code multiplier 308a. The stream comprising S1S2 and the pilot sequence Ul leaves the antenna gain block 304b to the phase changer 306a, where it changes phase according to the input of the changer control block phase 312a and subsequently enters the code multiplier 308b . The stream comprising -S2 * S1 * and the pilot sequence U2 leaves the antenna gain block 304c to the code multiplier 308c, and the same current, -S2 * S1 * and the pilot sequence, leaves the gain block * of 304d antenna to the 306b phase changer, where it changes phase according to the input of the phase changer control block 312b and subsequently enters the code multiplier 308d. The code multipliers 308a-308d multiply the appropriate current by means of a combination code. The multiplied stream of codes S1S2 and the pilot sequence Ul is subsequently input to the RF transmitter 314a for transmission in Ant. 1. The changed phase current multiplied by codes S1S2 and the pilot sequence Ul is input to the RF transmitter 314b for transmission in Ant. 2. The current multiplied by codes -S2 * S1 * and the pilot sequence U2 is input to the transmitter of RF 314c for transmission in Ant. 3, and the changed phase current multiplied by codes -S2 * S1 * and the pilot sequence U2 is input to the RF transmitter 314d for transmission in Ant. 4. The RF transmitter 314a - 314d performs the modulation and conversions of the carrier before transmitting the currents in Ant. 1 - Ant. 4. RF transmitters can perform baseband pulsing, modulation, and bearer conversion. In some implementations it is possible to choose to apply the non-zero weighting after the conformation of baseband pulses and modulation.

The receiver of Figure 5 can be modified for use with transmitter 300 of Figure 3. In this case, receiver 500 could work similarly but inputs 502a and 502b could enter Ul (Chl + Ch20) and U2 (Ch3 + Ch40) , respectively, to the channel estimator 504c. In another modality for the diversity of 4 antennas, dedicated pilot channels and common pilot channels can be implemented in a combined mode. Referring now to Figure 12, a block diagram of portions of a dedicated / common pilot channel STID transmitter 1200 according to another embodiment of the invention is shown. The transmitter 1200 operates essentially in the same way as the transmitter 300 of Figure 3 with the exception that the common pilot channels are aggregated in Ant. 1 and Ant. 3. The sequences of common pilot channels Pl and P2 are input to the pilot sequence processing derivation inputs 1218a and 1218b, respectively. Then, the pilot sequences are processed separately through the antenna gain blocks 1220a and 1220b, and the code multipliers 1222a and 1222b. The encoded outputs of the code multipliers 1222a and 1222b are then input to the RF transmitters 1214a and 1214c, respectively, of the RF transmitter 1214. The RF transmitters can perform baseband pulses, modulation, and bearer conversions. . In some implementations it is possible to select to apply the non-zero weighting after the conformation of baseband pulses and modulation. The transmitter 1200 of Figure 12 provides common pilot channels without variations by Ant jumps. 1 and Ant. 3 and dedicated pilot channels in Ant. 1, Ant. 2, Ant. 3, and Ant. 4. Pilot sequences may be multiplexed in a range, for example in a mode in which there are 15 slots in a transmission frame. Antenna gains can be set differently for common and dedicated control channels. The antenna gains can also be of variable time. Referring now to Figure 13, there is a block diagram of portions of a receiver 1300 for use with the transmitter of Figure 12. The receiver 1300 comprises the channel 1 and channel 2 processing leads with inputs 1302a and 1302b, and channel 3 and channel 4 processing leads with inputs 1302c and 1302d; a phase changer input 1304, a channel estimator 1306, a STTD demodulator, a traffic signal input 312, and a deinterleaver and decoder 1314. The received pilot sequences Pl, Ul, P2, and U2 are input to the inputs 1302a, 1302b, 1302c, and 1302d, respectively, of the receiver 1300. The channel estimator 1306 performs channel estimation using, for example, a low pass filter having an average function, and outputs a combined estimate for the channels 1 and '2 (chest 1,2) 1308a, and a combined estimate for channels 3 and 4 (chest 3,4) 1308b. The channel estimates are then input to the STTD demodulator 1310, which processes the traffic signals received from the input 1312 using the channel estimates. The demodulated signal is then processed in the rake receiver, combiner, deinterleaver, and channel decoder 1314 to generate the received symbols SI, S2. An earlier knowledge of the variation by phase jumps can be used for power control purposes. Referring now to Figure 14, portions of a receiver for estimating power control are shown, according to one embodiment of the invention. The receiver 1400 includes a channel estimator 1402, channel estimate derivation inputs 1404a-1404d, phase changer inputs 1408a and 1408b, phase changers 1406a and 1406b, channel estimate outputs 1410a and 1410b, quadrature blocks 1412a and 1412b, and a control processor 1414. The channel estimator 1402 calculates the channel coefficients of the common or dedicated channels of the channel., for example, the example transmitter 1200, for all four antennas during a given interval "t". This can be a channel prediction for the interval t + 1, alternatively the channel estimate for the interval t can be used in slow attenuation channels. These channel coefficients are denoted by chanest # l (t), chanest # 2 (t), chañest # 3 (t), and chanest # 4 (t) at inputs 1404a-1404d, respectively. For multiple rake receiver fingers, for example chanest # l (t) is a vector channel estimate that corresponds to the rake receiver fingers of Ant. 1. Using the prior knowledge of variation by phase jumps at the inputs of the phase shifter 1408a and 1408b and knowledge of the channel estimate for the current interval "t", the channel coefficients for the interval "t + 1" are estimated: chanest # 12 (t + 1) = chañest # l (t) + chanest # 2 (t) ef12 (t + 1) ahanest # 34 (t + 1) = chanest # 3 (t) + chanest # 4 (t) e * 3 (t + 1) where f12, 034 are known as priority. The estimation of the received signal power for the interval (t + 1) can be made based on chanest # 12 (t + 1) and chanest # 12 (t + 1); recharged_power (t + 1) = \\ chanest # 12 (t + 1) \\ 2 + IIchanest # 34 (t + 1) || 2 A power control command is generated by the processor 1414 using the received-power estimation. The method and apparatus of the invention can also be implemented with diversity in the Walsh code domain. Referring now to Figure 7, there is a block diagram of portions of a space time disperser transmitter (STS) 700 according to one embodiment of the invention; The transmitter 700 is an STS mode of the transmitter 150 of the figure in which the space-time block processor performs the transform in the Walsh code domain. The matrix of the STS block code can be represented as: S \ WX-S2 * W2 where Wx = [Wx Wx] W2 = [WX Wx] (3) IS2WX + S \ * W2 \ As has been done for the embodiment of FIG. 1a, each row of the matrix and its changed phase version are each transmitted in separate antennas Ant. 1-7Ant. 4. The symbols SI and S2 in each row are transmitted simultaneously through two periods of symbols, instead of sequentially. The data symbols are input to the transmitter 700 at the input 718 of the channel encoder 720. The channel encoder 720 encodes, punches, interleaves, and formats the input data symbols and extracts any other output symbol from the SI encoder as data. pairs and any other data symbol of encoder S2 as odd data. The even data is then processed through the symbol repetition blocks 702a, b, e, f, Walsh function blocks 704b and 704d, Walsh multipliers 706a, b, e, f, summers 708a -708d and complex add-ons 710a and 710b. The odd data is processed by the symbol repetition blocks 702c, d, g, h, the Walsh function blocks 704b and 704d, the Walsh multipliers 706c, d, g, h, the summers 708a -708d, and the add-ons complexes 710a and 710b. The result at the output of the complex adder 710a is the array row ÍSIPF, -52 * W2 and the result at the output of the complex adder 710b is the array row S2Wl-Sl * W2. Subsequently Slfj - S2 * W2 is introduced to the complex multiplier 712a to generate. { Sl ^ - S2 * W2} e and S2W¡ - Sl * W2 is introduced to complex multiplier 712b to generate. { S2W -51 * 2] eJ * 2.

Then, SiW-y-S2 * W2 is introduced to the RF transmitter 714a for transmission in Ant. 1,. { SIW ^ - S2 * W2} eg is introduced to the transmitter RF 714b for transmission in Ant. 2, S2W - Sl * W2 is introduced to the RF transmitter 714c for transmission in Ant. 3, and. { Sl ^ - Sl * W2je-7 * 2 is input to the RF transmitter 714d for transmission in Ant. 4. Referring now to Fig. 9, a block diagram of portions of a mode of a receiver 900 for use with the transmitter 700 of Figure 7. The transmitter 700 comprises the input 912, the Walsh function blocks 902b and 902d, the multipliers 902a and 902c, the channel multipliers 904a-904d, the complex add-ons 906a and 906b, the multiplexer (Mux). ) 908, and output 910. A received input signal is received 'at input 912, and processed by the STS demodulator. The procedures for transmission of pilot channels and channel estimation can be the same as explained in the case of STTD. The channel estimates 904c and 904b can be the same as for 412a, 412b of FIG. 4 for the case of the common pilot channel without variation by jumps. For the case of common pilots of variations by jumps or of dedicated pilot transmission the channel estimates can be obtained from the channel estimation block 504 of figure 5. These channel estimates are input to the STS demodulator in figure 9 as hl and h2. hl corresponds to the combined channel estimate of Ant. 1, Ant. 2 and h2 correspond to the Ant channel estimate. 3, Ant. . After the demodulation of STS using 902a, b, c, d and 904a, b, c, d, and 906a, b, the output of 908 is the STS demodulated signal that will be sent to the rake combiner, deinterleaver, and decoder block of channels 512 of Figure 5. The proposed invention may also be implemented in an orthogonal transmission diversity (OTD) embodiment of the invention. Referring now to Figure 8, a block diagram of portions of an OTD transmitter 800 according to one embodiment of the invention is shown. The transmitter 800 comprises an input 822, a channel encoder 820, symbol repetition blocks 802a-802d. * Walsh function blocks 804a and 804b, Walsh multipliers 806a-806d, complex add-ons 808a-808b, complex multipliers 810a and 810b, RF transmitters 812a-812d. The transmitter is an orthogonal transmission diversity mode (OTD) of the transmitter 150 of the figure in which the space-time block processor performs the transform in the Walsh code domain. The OTD block code matrix used can be represented as: As was done for the embodiment of FIG. 1a, each row of the array and its phase change version is each transmitted on separate antennas Ant. 1 - Ant 4. The data symbols are input to the transmitter 800 at the input 822 of the channel decoder 820. The channel decoder 820 decodes, punches, interleaves, and formats the input data symbols and extracts any other symbol YES output of the encoder as even data and any other symbol S2 of the decoder output as odd data. The even data is then processed through the repetition blocks of. symbols 802a and 802b, the Walsh function block 804a, the Walsh multipliers 806a and 806b, and the complex add-ons 808a. The odd data is processed through the symbol repetition blocks 802c and 802d, the Walsh function block 804b, the Walsh multipliers 806c and 806d, and the corrugator 808b. The result of the output of the complex adder 808a is SW ^ and the result at the output of the corrugator adder 808b is S2W1 - S \ Wl is then introduced to the complex multiplier 818a to generate. { SlWt je-7 * 1 and S2W2 is introduced to complex multiplier 818b to generate ^ S2W2 ^ eJ '* 2 | SIW and is subsequently introduced to the RF transmitter 812a for transmission in Ant. 1, . { 1? > 1]] ß? 1 is introduced to the RF transmitter 812b for transmission in Ant. 2, S2Wi is introduced into the RF transmitter 812c for transmission in Ant. 3, and. { S2W2} eJ * 2 is introduced to the RF transmitter 812d for transmission in Ant. Four . Referring now to Figure 10, a block diagram of portions of a mode of a receiver 1000 is shown for use with the transmitter 800 of Figure 8. The transmitter 800 comprises an input 1010, Walsh function blocks 1002a and 1002b, Walsh multipliers 1010a and 1010b, multipliers 1004a and 1004b, multiplexers 1006 and an output 1008. A received input signal received at input 912 is demodulated using an OTD 1000 demodulator using knowledge of channel coefficients hl * and h2 *. The channel coefficients hl and h2 for this OTD block are derived in the same manner as explained in figure 4 and figure 5. The OID 1000 demodulator is implemented using '1010, 1010a, b and 1012a, and 1004a, b and 1006 The demodulated output of OTD 1008 is sent to the rake combiner, deinterleaver, and channel decoder block 512 of Fig. 5. The embodiment of Fig. 1 can also be implemented in a TDMA transmitter to operate in an EDGE system. Referring now to Figure 11, a block diagram of portions of a long block code transmitter ST according to one embodiment of the invention is shown. The transmitter 1100 comprises the inputs 1118, 1120, symbol stream processing derivation inputs 1116a-1115d, time reversal blocks 1102 and 1104, complex conjugate blocks 1106a and 1106b, multipliers 1108, phase multipliers 1110a and 1110b, blocks controllers phase multipliers 1112a and 1112b, and antennas Ant. 1, Ant. 2, Ant. 3, and Ant. 4. The channel encoder 1120 encodes, punches, interleaves and formats a stream of symbols at the input 1118. The channel decoder 1120 also divides the stream of input symbols into odd and even data streams. The even data stream is input to the bypass input 1116a and the RF transmitter 1122a for transmission in Ant. 1 during the first half of a data burst transmission and the odd data stream is input to the bypass input 1116c and to RF transmitter 1112c for transmission in Ant. 2 during the first half of the data burst transmission. During the second half of a burst transmission, the even data stream is input to the bypass input 1116b, the time is received in the time reversal block 1102, it is conjugated by complexes in the complex conjugate block 1106a and is sent to the RF transmitter 1122c for transmission in Ant. 3. The odd data stream is input to the bypass input 1116d, the time is inverted in the time reversal block 1104, complexed in the complex conjugating block 1106b, multiplied by a negative in the multiplier 1108 and sent to the RF transmitter 1122d for transmission in Ant. 4 during the second half of the data burst transmission. A training sequence SEQ1 is integrated in the middle of the burst transmission in Ant. 1 and a training sequence SEQ2 is integrated in the middle of a burst transmission in Ant. 2. Phase multipliers 1112a and 1112b, change the phase of the inputs to RF transmitters 1122b and 1122d, using multiplier blocks 1110a and 1110b, respectively. The output of the phase multiplier 1112a is then input to the RF transmitter 1122b for transmission in Ant. 2 and the output of the phase multiplier 1112b is input to the RF transmitter 1122d for transmission in Ant. 4. RF transmitters can effect conformation of baseband pulses, modulation, and carrier conversion. In some implementations, it may be selected to apply phase multiplication after the baseband and modulation pulse shaping steps. The phase rotation applied in the phase multipliers 1122a and 1122b is kept constant during the burst transmission period, changing the phase based on a burst transmission by burst transmission. The phase may be selected periodically or randomly from an MPSK constellation as explained above. In a preferred embodiment the phase rotation in the Ant. 4 remains the same as the phase rotation in Ant. 2 with a change of 180 degrees or multiplied by -1. The phase multiplication can be carried out before or after the formation of baseband pulses. In an alternative embodiment of Figure 11 the transmission in Ant. 1 and Ant. 3 can be exchanged. The transmitter shown in Figure 3 can also be applied to EDGE with some modification. The space-time code described in 316 is applied in block form instead of the symbol form for an EDGE application. The duration of the block can be selected as a first half of the burst transmission. In EDGE the length of the first half and the second half of the burst transmissions are equal to 58 symbols. In this case SI and S2 denote a block of symbols and () * denotes the time reversal of a block of symbols and a complex conjugation operation. YES * denotes that the time of the SI symbol block is inverted and is conjugated by complexes. -S2 * denotes that the time of the symbol block S2 is inverted, is conjugated by complexes and is multiplied by -1.0. The Ul and U2 pilot sequences can be selected as two training sequences such as the well-known CAZAC sequences. Scatter codes 308a, b, c, d, will not be applied in EDGE. Phase multiplication blocks 306a and 106b are retained. A receiver designed for a 2-antenna space-time block code can be used for the embodiments of Figure 1 or Figure 2. From the above description and embodiments, someone skilled in the art will realize that, although the method and The apparatus of the present invention has been illustrated and described in relation to particular modalities thereof, it will be understood that numerous modifications and substitutions of the described modalities can be made, and that numerous other embodiments of the invention can be implemented without departing from the spirit and scope of the invention. the invention as defined in the following claims. It is noted that in relation to this date, the best method known to the applicant to carry out the aforementioned invention, is that which is clear from the present description of the invention.

Claims

CLAIMS Having described the invention as "above, the content of the following claims is claimed as property: 1.
A method for transmitting a signal of a plurality of antennas, characterized in that the method comprises the steps of: receiving a stream of symbols in a transmitter, performing a transform on that stream of input symbols to generate a transform result, that transform result comprises an orthogonal space-time block code N x? ' , and generates N 'first signals, complex weighting not null, over time, at least one of the N' first signals of that transform result generates at least one second signal, each of the at least one second signal changes from phase in relation to one of the N 'first signals from which it was generated, and transmission, substantially simultaneously, of each of the N' first signals of that result of being transformed into one of a first at least one antenna and, each of the at least one second signal in one of a second at least one antenna, the method according to claim 1, characterized in that the stream of input symbols comprises the symbols SI, S2 and the block code. of space time comprises a space-time block code of 2 x 2, and the N 'signals comprise the currents of (SI, S2) transmitted in ti and t2, respectively, and (-S2 *, SI *) transmitted in you and t2, respectively 3.
The m The method according to claim 1, characterized in that the input symbol stream comprises the symbols SI, S2, and the space-time block comprises a space-time block code of 2 x 2, and the N 'signals comprise the currents. of (SI, -S2 *) transmitted in you and t2, respectively, and (S2, SI *) transmitted in you and t2, respectively.
The method according to claim 1, characterized in that the first at least one antenna and the second at least one antenna comprises a first plurality of N 'antennas and a second plurality of N' antennas, respectively, the symbol current of The input comprises a stream of traffic channel symbols and in that the method additionally comprises the steps of: transmitting each 2N 'common pilot channel signals in an antenna separated from that first plurality of N' antennas or an antenna separate from that second plurality of N 'antennas.
5. The method according to claim 1, characterized in that the input symbol stream comprises a traffic channel stream and the method further comprises the step of: receiving N 'signals from common pilot channels in that transmitter; complex weighting not null, over time, of each of the N 'signals of common pilot channels to generate N' signals from common pilot channels with non-zero complex weighting; transmission, substantially simultaneously, of each of the N 'signals of common pilot channels in one of the first at least one antenna, and each of the N' signals of common pilot channels of complex non-zero weighting in one of the second at least one antenna.
6. The method according to claim 1, characterized in that the input symbol stream includes a traffic channel stream, and the method further comprises the steps of: inserting each of the N 'pilot signals after which each of the N 'first signals of the result of the transform results in the generation of N' first signals including an inserted pilot signal; wherein the non-zero complex weighting stage comprises a complex non-zero weighting, over time, of each of the first N 'signals including a pilot signal inserted to generate N' second signals including an inserted pilot signal; and where the transmission stage comprises the transmission, substantially simultaneously, of each of the N 'first signals including pilot signals inserted into one of a first at least one antenna, and each of the second signals includes a pilot signal inserted into one of a second at least one antenna The method according to claim 1, characterized in that the non-zero complex weighting stage comprises a phase change of at least one of the first N 'signals using a continuous analog phase sweep. The method according to claim 1, characterized in that the non-zero complex weighting stage comprises the phase change of a predetermined variation sequence by at least one of the first N 'signals. The method according to claim 8, characterized in that the variation weights per hop for the predetermined variation sequence are derived from a constellation PSK having Z states in which all the states are sampled with the same frequency in a transmission box. The method according to claim 8, characterized in that the hop variation weights for the predetermined hopping sequences are derived from a PSK constellation having Z states. The method according to claim 1, characterized in that the time slot block code comprises a STS block code of 2 x 2 and the N 'first signals comprise the currents of (S1W1 - S2 * 2) transmitted in you. and (S2W1 + S1 * W2) transmitted in you, where W1 and W2 are each a series concatenation of at least two Walsh codes. The method according to claim 1, characterized in that the time slot block code comprises a STS block code of 2 x 2 and the N 'first signals comprise the currents of (S1W1 + S2W2) transmitted in ti and ( -S2 * W1 + S1 * W2) transmitted in you, where W1 and W2 each are a series concatenation of at least two Walsh codes. 13. An apparatus for transmitting a signal, characterized by the transmitter comprising: a stream of input symbols; a processor for effecting a transform in the stream of input symbols to generate a transform result, the transform result comprises an orthogonal space-time block code 'x N', and for generating N 'first signals. at least one weigher for, non-zero complex weighting, over time, at least one of the first N 'signals of the transform result to generate at least one second signal, each of the at least one of the second signals weighted with phase change in relation to one of the N 'first signals from which it was generated, Y; a transmitter for transmitting, substantially simultaneously, each of the first N 'signals of the transformed result into one of a first at least one antenna, and each of the N' second signals into one of a second at least one antenna . The apparatus according to claim 13, characterized in that the input symbol stream comprises the symbols SI, S2 and the space-time block code comprises a space-time block code of 2 x 2, and the N ' first signals comprise the currents of (SI, S2) transmitted in ti and t2, respectively, and (-S2 *, SI *) transmitted in ti and t2, respectively. The apparatus according to claim 13, characterized in that the input symbol stream comprises the symbols SI, S2 and the space-time block comprises a time-space block code of 2 x 2 and the first N 'signals comprise the currents of (SI, -S2 *) transmitted in ti and t2, respectively, and (S2, SI *) transmitted in ti and t2, respectively. The method according to claim 13, characterized in that the first at least one antenna and the second at least one antenna comprise a first plurality of N 'antennas and a second plurality of N' antennas, respectively, the symbol current of The input comprises a stream of traffic channel symbols and wherein the transmitter additionally comprises: at least one input for the reception of N 'signals from common pilot channels in the transmitter; a weighting, the non-zero complex weighting for complex weighting not null, over time, each of the N 'common pilot channel signals to generate common pilot channel signals with non-zero complex weighting; and wherein the transmitter additionally transmits each of the N 'signals of common pilot channels in a separate antenna of one of the first at least one antenna and each of the N' signals of common pilot channels with complex weighting not null in one antenna separated from one of the second at least one antenna. The apparatus according to claim 13, characterized in that the input symbol stream includes a stream of traffic channels and wherein the apparatus additionally comprises; a multiplexer for inserting each of the N 'pilot signals after one of the N' first 'signals of the transform results in the generation of N' first signals including a pilot signal inserted; and at least one weighting for the non-zero complex weighting, over time, each of the N 'signals includes a pilot signal inserted to generate N' second signals including an inserted pilot signal; and wherein the transmitter transmits, substantially simultaneously, each of the first N 'signals including a pilot signal inserted into one of a first at least one antenna, and each of the N' second signals includes a pilot signal inserted in one of a second at least one antenna. The apparatus according to claim 13, characterized in that the at least one weight changes the phase of at least one of the first N 'signals using a continuous analog phase sweep. 19. The apparatus according to claim 13, characterized in that the at least one phase weight changes the phase of at least one of the first N 'signals using a predetermined jump variation sequence. The apparatus according to claim 19, characterized in that the hopping variation weights for the predetermined hopping sequence are derived from a PSK constellation by the random permutation of Z possible states for successive intervals of the transmission frame. The apparatus according to claim 13, characterized in that the space-time block code comprises an STS block code of 2 x 2 and the first N 'signals comprise the currents of (S1W1-S2 * 2) transmitted in you, and (S2W1 + S1 * 2) transmitted in you, where W1 and W2 are each, a serial concatenation of at least two Walsh codes. 22. The apparatus according to claim 13, characterized in that the space-time block code comprises a STS block code of 2 x 2 and the N 'first signals comprise the currents' of (S1W1 + S2W2) transmitted in you. and (-S2 * 1 + S1 * W2) transmitted in you, and where W1 and W2 are each, a serial concatenation of at least two Walsh codes. 23. A method for transmitting a signal of a plurality of antennas, characterized in that the method comprises the steps of: receiving a stream of symbols in a transmitter; performing a transform on the stream of input symbols to generate a transform result, the transform result comprises an orthogonal time-space block code N x N ', and the generation of N' first signals; complex weighting not null, through the time, of at least one of the N 'first signals of the transformed result to generate at least one second signal, each of the at least one second signal changes phase in relation to one of the N 'first signals from which it was generated, and wherein that non-zero complex weighting comprises phase change of at least one of the first N' signals by a predetermined variation sequence by jumps, wherein the variation weight by jumps for the sequence of variations by predetermined jumps it is derived from a constellation PSK that has 8 states, and where the sequence of variations per jump is degrees is (0, 135, 270, 45, 180, 315, 90, 225); and transmission, substantially simultaneously, of each of the first N 'signals of the transformed result into one of a first at least one antenna and, each of the at least one second signals into one of a second at least one antenna . 24. An apparatus for transmitting a signal, characterized in that the transmitter comprises: a stream of input symbols; a processor for effecting a transform in the stream of input symbols to generate a transform result, the transform result comprises an orthogonal space-time block code 'x?' , and generates N 'first signals; at least one weighting for, non-zero complex weighting, through the time of, at least one of the first N 'signals of the transformed result to generate at least one second signal, each of the at least one second weighted signal changes from phase in relation to one of the first N 'signals from which it was generated, and wherein the non-zero complex weight comprises a phase change of at least one of the first N' signals by means of a predetermined jump variation sequence. , wherein the variation weight per hop for the predetermined hop variation sequence is derived from a PSK constellation having 8 states, and wherein the predetermined hop variation sequence in degrees is (0, 135, 270, 45, 180, 315, 90, 225); and a transmitter for transmitting, substantially simultaneously, each of the first signals of the transformed result into one of a first at least one antenna, and each of the N 'second signals into one of a second at least one antenna.